JP2000102283A - Single-phase brushless motor and pole-place discrimination circuit thereof - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a single-phase brushless motor, in which the place of the magnetic poles of a rotor magnet can be discriminated, and a pole-place discrimination circuit thereof. SOLUTION: The total of the sectional areas of interpole yokes 58 is formed approximately as one tenth of the sectional area of a main leg section 57. Accordingly, magnetic flux is partially saturated by the interpole yokes 58, even by a slight armature current. Since the armature current increases suddenly when magnetic flux is saturated, the polarity of a rotor magnet 52 is discriminated on the basis of the sudden increased timing. For example, when the rotor magnet 52 is stopped at a place (a), magnetic flux becomes saturated at an early time, because magnetic flux 52a generated by the rotor magnet 52 and magnetic flux 54a generated through energizing are directed in the same direction, and the sudden increased timing of the armature current arrives at an early time. Since magnetic flux 52b generated by the rotor magnet 52 and magnetic flux 54b generated by electric conduction are directed in the opposite directions under the state of (b), however, the time for the saturation of magnetic flux to occur is delayed, and the sudden increased timing of the armature current comes at a late time.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a single-phase brushless motor and a magnetic pole position determining circuit of the single-phase brushless motor, and more particularly, to a method of rotating a rotor magnet without rotating the rotor magnet when the motor is stopped. The present invention relates to a single-phase brushless motor capable of determining a magnetic pole position and a magnetic pole position determining circuit of the single-phase brushless motor.

[0002]

2. Description of the Related Art In order to accurately start a single-phase brushless motor (skeleton motor), the direction of the first wave of energization to an armature winding is set according to the magnetic pole position of a rotor magnet when the motor is stopped. There is a need to. Tokuhei 7-63
Japanese Patent Publication No. 232 discloses a sensorless drive circuit for driving a single-phase brushless motor sensorless without using a position detecting element such as a Hall element. In this sensorless drive circuit, before starting the motor, a predetermined current is supplied to the armature winding to move the magnetic pole of the rotor magnet to a predetermined position, and then the motor is started.

[0003]

However, when the rotor magnet is moved, the rotor magnet vibrates, and the vibration does not easily stop. The time until such vibration stops varies depending on the inertia of the motor, etc.
It takes tens of seconds for long ones. In addition, if the motor is started while the rotor magnet is vibrating, the motor may rotate reversely depending on the vibration condition. Therefore, in the drive circuit described in Japanese Patent Publication No. 7-63232, the motor must be started after the vibration of the rotor magnet stops, and the motor cannot be started quickly. was there. Furthermore, Japanese Patent Publication 7-6323
In the drive circuit described in Japanese Patent Laid-Open Publication No. 2002-207, the rotor magnet is moved or not moved at the time of starting depending on the stop position of the rotor magnet. There was a problem that the amount of rotation could not be grasped.

SUMMARY OF THE INVENTION The present invention has been made to solve the above-mentioned problems, and a single-phase brushless motor and a single-phase brushless motor capable of determining the magnetic pole position of a rotor magnet without rotating the rotor magnet. It is an object of the present invention to provide a magnetic pole position determining circuit for a phase brushless motor.

[0005]

Means for Solving the Problems To achieve this object, a single-phase brushless motor according to claim 1 comprises a stator core on which an armature winding is wound, and a rotor magnet, and the rotation The stator magnet is configured to stop at a predetermined angle from the lock position, and the stator core includes a pair of main legs, and an inter-pole yoke connecting the pair of main legs, The cross-sectional area of the pole yoke is formed to be approximately one-tenth or less of the cross-sectional area of the main leg.

When power is supplied to the armature winding of a single-phase brushless motor, magnetic flux links (occurs) to the stator core. The magnetic flux loops in the direction of one main leg, the yoke between poles, the other main leg, and the one main leg, or in the opposite direction, depending on the direction of conduction. The magnetic flux of the stator core increases with an increase in the armature current flowing through the armature winding. When the magnetic flux of the stator core is saturated due to the increase, the self-inductance value of the armature winding sharply decreases, and the armature current sharply increases.

[0007] Incidentally, a magnetic flux is also applied to the stator core by a magnetic field generated by a rotor magnet. For this reason,
When the magnetic flux provided by the rotor magnet and the magnetic flux generated by energizing the armature winding are in the same direction (addition direction), compared to when both magnetic fluxes are in the opposite direction (subtraction direction). However, the magnetic flux of the stator core saturates quickly, and the armature current rapidly increases at an early stage. Therefore, the polarity of the rotor magnet at the time of stop can be determined based on the timing at which the armature current suddenly increases.

According to the single-phase brushless motor of the first aspect, the stator core is provided with the pole yoke having a smaller cross-sectional area than the main leg, so that the magnetic flux of the stator core is reduced by the pole yoke. Saturates first. Moreover, since the cross-sectional area of the pole yoke is formed to be about one-tenth or less of the cross-sectional area of the main leg, the pole yoke is fixed by a small armature current without rotating the rotor magnet. The magnetic flux of the armature core is partially saturated, and the stator is operated as a saturable reactor, so that a sudden increase in the armature current can be realized. Further, since the rotor magnet is configured to stop at a predetermined declination from the lock position by the cogging torque, the magnetic pole position of the rotor magnet at the time of stopping the motor is determined based on the sudden increase timing of the armature current. can do.

According to a second aspect of the present invention, there is provided a single-phase brushless motor.
2. The single-phase brushless motor according to claim 1, wherein the stator core includes the pair of main legs and a pair of interpole yokes connecting the pair of main legs at two different positions. 3. The pair of main legs and a pair of pole yokes constitute a closed slot structure that encloses the rotor magnet, and the total cross-sectional area of the pair of pole yokes is one of the cross sections of the main leg. It is formed to be approximately one-tenth or less of the area.

[0010] The single-phase brushless motor according to claim 3 is
3. The single-phase brassless motor according to claim 1, wherein the pole yoke is formed of ferrite having a rectangular magnetizing characteristic, and a cross-sectional area of the pole yoke formed of the ferrite is the main leg. 4. Is formed to approximately four-tenths or less of the cross-sectional area of. Ferrite having a rectangular magnetizing characteristic has a lower magnetic flux density than silicon silicon steel sheet and a sharp change in magnetic permeability.Therefore, by forming the pole yoke with the ferrite, the cross-sectional area of the pole yoke is reduced by the cross-sectional area of the main leg. Even if it is formed about 4/10 or less, the magnetic flux of the stator core can be abruptly partially saturated by the pole yoke without rotating the rotor magnet.

[0011] The single-phase brushless motor according to claim 4 is
4. The single-phase brushless motor according to claim 1, wherein a notch formed in the stator core for stopping the rotor magnet at a predetermined angle from a lock position by cogging torque is provided. 5. The inter-electrode yoke is formed adjacent to the notch. As described above, by forming the pole yoke adjacent to the cutout, the single-phase brushless motor can be simplified and easily manufactured.

A magnetic pole position discriminating circuit for a single-phase brushless motor according to a fifth aspect is used for the single-phase brushless motor according to any one of the first to fourth aspects, and inspects the armature winding. A test voltage energizing circuit for energizing a voltage, a current detecting circuit for detecting a current flowing to the armature winding by applying the test voltage by the test voltage energizing circuit, and an increase in the armature current detected by the current detecting circuit A polarity discriminating circuit for discriminating a magnetic pole position when the rotor magnet is stopped based on a speed.

According to the magnetic pole position discriminating circuit of the single-phase brushless motor according to the present invention, the inspection voltage is applied to the armature winding by the inspection voltage applying circuit, and the armature current flowing to the armature winding by the application of the inspection voltage. Is detected by the current detection circuit. A pole formed in the stator core of the single-phase brassless motor according to any one of claims 1 to 4 to approximately 1/10 or less of the cross-sectional area of the main leg (approximately 4/10 or less in the case of ferrite). Since the intermediate yoke is provided, the magnetic flux generated in the stator core by the application of the inspection voltage to the armature winding is partially saturated by the yoke between the poles. As a result, the self-inductance value of the armature winding sharply decreases, and the armature current sharply increases. Such a sudden increase in the armature current is detected by the current detection circuit, and the polarity discrimination circuit determines the magnetic pole position of the rotor magnet when the motor is stopped by the sudden increase timing, that is, the current increase speed. It should be noted that the rate of increase of the armature current (rapid increase timing, partial saturation timing of magnetic flux) can be detected, for example, based on whether or not the armature current has reached a predetermined value a predetermined time after the start of application of the inspection voltage. .

A magnetic pole position determining circuit for a single-phase brushless motor according to a sixth aspect of the present invention is the magnetic pole position determining circuit for a single-phase brushless motor according to the fifth aspect, wherein the inspection voltage energizing circuit includes a single-phase brushless motor as an inspection voltage. And outputs an alternating voltage having a frequency higher than the frequency at the time of driving. Therefore, the motor can be kept stopped by the cogging torque, and the inspection voltage can be supplied while the motor is stopped without rotating the rotor magnet.

[0015]

Hereinafter, preferred embodiments of the present invention will be described with reference to the accompanying drawings. The driving principle of the brushless motor in this embodiment has already been described in Japanese Patent Application No. 7-207665.
The description is omitted.

FIG. 1 is a sectional view of a single-phase brushless motor according to this embodiment. This single-phase brushless motor 51 uses a rotor magnet 52 as a rotor, an armature winding 53 as a stator, and a stator core 54 around which the armature winding 53 is wound. , A skeleton type single-phase brushless motor.

Notches 56 are formed near the magnetic poles 55 of the stator core 54 at two positions symmetrical with respect to the rotor magnet 52. Due to the notch 56, two positions for maintaining a predetermined declination θ from a lock position where the magnetic flux axis X of the rotor magnet 52 and the magnetic flux axis Y generated in the stator core 54 by energizing the armature winding 53 coincide with each other. In addition, the rotor magnet 52 is configured to be stopped by the cogging torque (see FIGS. 1A and 1B).

The pair of main legs 57 of the stator core 54 are connected by a pair of interpole yokes 58 formed between the pair of notches 56 and the magnetic poles 55.
4 has a closed slot structure including the rotor magnet 52 therein. The sum of the cross-sectional areas of the pair of pole yokes 58 is
The main leg 57 is formed so as to have a sectional area of approximately one-tenth or less of a minimum portion. For this reason, the magnetic flux generated in the stator core 54 by the energization of the armature winding 53
Saturates first. In addition, as described above, the total cross-sectional area of the pole yoke 58 is approximately one-tenth or less of the cross-sectional area of the minimum portion of the main leg 57, so that the rotor magnet 52 does not rotate slightly. The magnetic flux of the stator core 54 is partially saturated in the yoke 58 between the poles by the armature current.

As will be described later, in the present embodiment, the polarity of the rotor magnet 52 at the time of stoppage is determined by utilizing the partial saturation of the magnetic flux in the gap yoke 58. Then, the energizing direction of the energizing first wave for driving the brushless motor 51 is determined based on the determination result.

FIG. 2 shows the single-phase brushless motor 5 described above.
1 is a circuit diagram of one sensorless drive circuit 1. FIG. With this motor drive circuit 1, the single-phase brushless motor 51 can be driven sensorlessly without using a position detecting element such as a Hall element.

The motor drive circuit 1 includes an auxiliary power supply circuit 2, an inverter circuit 3, a current detection circuit 4, a sampling circuit 5, an amplification circuit 6, a start compensation circuit 7, a priority circuit 8, a distribution circuit 9, a zero reset circuit 10, and a check. Wave oscillation circuit 2
1, comparison circuit 23, conduction direction setting circuit 24, switching circuit 2
5, monostable multivibrators MM1 and MM2, comparator CP1 and the like.

The auxiliary power supply circuit 2 is a circuit for generating and outputting stable 10 volt and 5 volt voltages from a 30 volt DC power supply 50. 1 generated by the auxiliary power supply circuit 2
A voltage of 0 volt is supplied to each circuit as a driving voltage.
The 5 volt voltage is supplied to the check wave oscillation circuit 21 and the comparison circuit 23.

The inverter circuit 3 includes a brushless motor 5
This is a circuit for energizing one armature winding 53 with a DC voltage of 30 volts as an alternating voltage. Inverter circuit 3
The source terminals of two P-MOS field-effect transistors Qu and Qv as upper-arm transistors are connected to the plus-side input terminal P of the DC power supply 50, and the lower-side input terminal N of the DC power supply 50 is connected to the lower arm. Source terminals of two N-MOS field effect transistors Qx and Qy as transistors are connected. These four field effect transistors Qu, Qv, Qx, Qy form two inverter arms corresponding to the armature winding 53.

Each field effect transistor Qu, Qv, Q
x and Qy are resistors Ru1 and Rv each having a gate terminal of 1 kΩ.
1, Rx1 and Ry1, each output u,
v, x, and y, respectively, and are configured to be turned on and off according to the output of the distribution circuit 9. Further, between the gate and the source of each of the field effect transistors Qu, Qv, Qx, Qy, resistors Ru2, Rv2, Rx2, Ry of 10 kΩ for protection and prevention of floating of the gate voltage.
2 are connected respectively. Further, a freewheel is provided between the source and the drain of each of the field effect transistors Qu, Qv, Qx, and Qy to recirculate the current caused by the back electromotive force generated in the armature winding 53 of the brushless motor 51 when the transistor is off. Diodes Du, Dv, Dx, D
y are connected in anti-parallel.

The current detection circuit 4 includes a brushless motor 51
Is a circuit for converting a current (armature current) flowing through the armature winding 53 into a voltage, and a 1Ω shunt resistor Rs inserted between the ground-side input terminal N of the DC power supply 50 and the inverter circuit 3. It consists of. As shown in FIG. 3A, when an alternating voltage is applied to the armature winding 53 of the brushless motor 51, an armature current shown in FIG. 3B flows. This armature current flows through the shunt resistor Rs as shown in FIG. 3 (c), and is converted into a voltage and output as shown in FIG. 3 (d).

The sampling circuit 5 stores the instantaneous value of the output voltage of the current detection circuit 4 and stores the instantaneous value in the amplification circuit 6
This is a circuit for outputting to The sampling circuit 5
An analog switch AS1, a capacitor C1, and a resistor R
2 is provided. One channel terminal of the analog switch AS1 is connected to the output terminal of the current detection circuit 4, and the other channel terminal is connected to one end of which has one end grounded.
It is connected to a μF capacitor C1 and a 2MΩ resistor R2. The gate of the analog switch AS1 is connected to the output terminal Q of the monostable multivibrator MM1,
It is turned on while the sampling command 66 (FIG. 3 (h)) is being output.

The capacitor C1 is connected to the analog switch AS
1 is connected to the output terminal of the current detection circuit 4 during ON, and stores the output voltage. Only the analog switch AS1, the resistor R2, and the non-inverting input terminal of the operational amplifier OP2 are connected to the non-ground terminal of the capacitor C1, and the resistance value of the resistor R2 is as large as 2MΩ.
The voltage value of the capacitor C1 is held for a predetermined time even after the analog switch AS1 is turned off. Therefore, the voltage value (instantaneous output) of the current detection circuit 4 immediately before the analog switch AS1 is turned off is stored in the capacitor C1.
FIG. 3E shows an output voltage waveform of the sampling circuit 5.

By the way, the commutation command 65 (FIG. 3 (g)) which triggers the commutation operation is, as described later, the output voltage of the current detection circuit 4 of the sampling circuit 5 where the output voltage is amplified by the amplification circuit 6. It is output when it becomes higher than the output voltage. Therefore, if a large voltage value is held in the capacitor C1 of the sampling circuit 5 for some reason,
The output voltage of the current detection circuit 4 cannot be higher than the amplified output voltage of the sampling circuit 5, the commutation command 65 cannot be generated, the commutation operation cannot be performed, and the brushless motor 51 stops. .

However, since the resistor R2 is connected in parallel to the capacitor C1 of the sampling circuit 5, the electric charge accumulated in the capacitor C1 is gradually discharged, though slightly, by the resistor R2. As a result, the voltage value of the capacitor C1 also gradually decreases. Therefore, the resistance R
2 is connected in parallel with the capacitor C1, the commutation command 65 can always be regenerated even if a large voltage value is erroneously held in the capacitor C1, and the brushless motor 51 is stopped. There is no end.

The resistance value of the resistor R2 is determined by the value of the capacitor C1.
And the lower limit of the commutation frequency of the inverter circuit 3 at the time of startup. That is, when the lower limit value of the commutation frequency at the time of starting is about 3 Hz, a time constant slightly larger than twice the 6 Hz cycle is set,
The resistance value of the resistor R2 and the capacitance of the capacitor C1 are determined so as to be in a range of about 0.2 seconds. In this embodiment, since the capacitance of the capacitor C1 is 0.1 μF, the resistance value of the resistor R2 is 2 MΩ.

Incidentally, depending on the type of the operational amplifier OP2 of the amplifier circuit 6, a leakage current (input bias current) may flow from the non-inverting input terminal to the ground. In such a case, the leakage current increases the voltage value of the capacitor C1, that is, the stored current detection circuit 4
(Commutation target voltage) changes in the ascending direction, so that a normal commutation operation cannot be performed. However, by connecting the resistor R2 in parallel with the capacitor C1, such a leakage current can flow through the resistor R2, so that the voltage value of the capacitor C1 can be prevented from increasing, and the voltage value of the capacitor C1 always decreases. Therefore, the output voltage of the current detection circuit 4 can be maintained in the capacitor C1.

The amplifier circuit 6 amplifies the voltage value stored in the sampling circuit 5 and outputs the amplified voltage value to the priority circuit 8. The amplifier circuit 6 includes an operational amplifier OP2 and two resistors R
3 and R4, and a 100 k.OMEGA. Variable resistor VR2 for reducing the output of the non-inverted amplifier to 1 or less. The commutation target voltage of the comparator C
Output to the inverting input terminal of P1.

The operational amplifier OP2 of the non-inverting amplifier has a non-inverting input terminal connected to a capacitor C1 as an output terminal of the sampling circuit 5, and an output terminal of the operational amplifier OP2 having a resistor R3 and a variable resistor having one end grounded. VR2
Is connected. The other end of the resistor R3 is connected to an operational amplifier OP
2 and one end of a resistor R4.
The other end of 4 is grounded.

The resistance values of the two resistors R3 and R4 of the non-inverting amplifier are both the same 100 kΩ. Therefore, the output of the sampling circuit 5 is supplied to the non-inverting amplifiers OP2 and OP2.
Amplified almost twice by R3 and R4. The output of the sampling circuit 5 that has been amplified approximately twice is output to the variable resistor VR2, reduced to one or less, and output to the priority circuit 8. In this embodiment, the output of the sampling circuit 5 that has been amplified approximately twice by the non-inverting amplifiers OP2, R3, and R4 is:
It is reduced to 0.9 times by the variable resistor VR2. Therefore, the output of the sampling circuit 5 as a whole is amplified by a factor of 1.8.

In the case of the single-phase brushless motor 51, the waveform of the armature current is closer to a triangular wave than in a three-phase machine. Therefore, in the case of a three-phase machine, the output voltage of the sampling circuit is amplified by about 1.2 to about 1.4 times, whereas in the case of a single-phase machine, it is desirable to amplify by about 1.5 to about 2.0 times.
For this reason, in the present embodiment, amplification is 1.8 times. FIG. 3F shows an output voltage waveform of the amplifier circuit 6.

It should be noted that the variable resistor VR2
By adjusting the position of the slider, the amplification factor of the entire amplification circuit 6 can be changed, so that the amplification factor can be changed in accordance with the use situation. That is, by adjusting the slider position of the variable resistor VR2, the brushless motor 5
Tuning can be performed so that the motor efficiency is maximized in the one steady operation region.

The starting compensation circuit 7 includes a brushless motor 51
Circuit for outputting the commutation target voltage to the inverting input terminal of the comparator CP1 instead of the amplified output of the sampling circuit 5 so that the brushless motor 51 can generate a sufficient starting torque at the start of It is.

The starting compensation circuit 7 has a resistor R5 of 10 kΩ. One end of the resistor R5 serves as an input terminal S of the starting compensating circuit 7 via the inverter IV1 of the switching circuit 25 and the resistor R32, and an auxiliary power supply. Connected to the 10 volt line of circuit 2. The other end of the resistor R5 is connected to the plus side terminal of a 10 μF electrolytic capacitor C2 and one end of a switch SW1, and the other end of the switch SW1 is
F is connected to the positive terminal of the electrolytic capacitor C3. Since the negative terminal of this electrolytic capacitor C3 is connected to the negative terminal of the other electrolytic capacitor C2, when the switch SW1 is turned on, both electrolytic capacitors C2 and C3 are connected in parallel, and The capacitance increases (to 43 μF). Both electrolytic capacitors C
2 and C3 are both connected to a 100 kΩ resistor R6 whose one end is grounded to the circuit and a cathode of a diode D1 whose anode is grounded. Diode D2
Connected to the anode.

The priority circuit 8 includes an output voltage of the sampling circuit 5 amplified by the amplification circuit 6 and a starting compensation circuit 7
And a circuit for outputting the larger one of the output voltages to the inverting input terminal of the comparator CP1 as a commutation target voltage, and is constituted by a diode D2. The diode D2 has an anode connected to the output terminal of the starting compensation circuit 7, a cathode connected to the output terminal of the amplifier circuit 6, and the inverting input terminal of the comparator CP1.

Referring to FIG. 4, when the single-phase brushless motor 51 is started, the priority circuit 8 outputs a signal from the comparator C
The commutation target voltage 60 output to the inverting input terminal of P1 will be described. The start compensation circuit 7 includes a capacitor C2 (C
3) and a differential circuit composed of a series circuit of resistors R5 and R6. When a voltage of less than 10 volts is applied from the auxiliary power supply circuit 2 to the input terminal S of the start compensation circuit 7 via the inverter IV1, the start compensation circuit 7 To the priority circuit 8 from the voltage value of slightly less than 10 volts, the voltage 61 gradually decreasing with time. On the other hand, the voltage of the current detection circuit 4 sampled by the sampling circuit 5 is amplified by about 1.8 times and output from the amplification circuit 6 (62). The larger of the two voltages 61 and 62 is output from the priority circuit 8 to the inverting input terminal of the comparator CP1 as the commutation target voltage 60. FIG. 4 also shows the output voltage 63 of the current detection circuit 4 to be compared with the commutation target voltage 60.

As described above, by setting the drive voltage of the start compensation circuit 7 higher than the detection voltage of the current detection circuit 4 and stabilizing the voltage, the commutation target voltage 60 is set high when the brushless motor 51 starts. be able to. Therefore, at the time of starting, a sufficient armature current can be supplied to generate a starting torque.

Since the diode D1 is connected in anti-parallel to the resistor R6 of the starting compensation circuit 7, the input terminal S
When the voltage to is 0 V, the capacitor C2 (C3) can be discharged quickly. Therefore, the DC power supply 50
Is turned off, or when the switch SW2 of the switching circuit 25 is switched from the start mode ST to the check mode CK, the voltage of the input terminal S becomes 0 volt (circuit ground), and the capacitor C2 (C3) Is reliably discharged in a short time. Therefore, the start compensation circuit 7 can function normally every time the brushless motor 51 starts to be started.

The switch SW1 of the starting compensation circuit 7 is a switch for setting the rotation direction of the single-phase brushless motor 51. When the switch SW1 is off,
The single-phase brushless motor 51 rotates in the forward direction (FIGS. 4 and 5), and conversely, rotates in the reverse direction when the switch SW1 is turned on (FIGS. 6 and 7). By turning off or on the switch SW1, the capacitances of the capacitors of the differentiating circuit of the starting compensation circuit 7 become 10 μF and 43 μF (= 10 μF).
+33 μF), and the gradient of the commutation target voltages 61 and 61 ′ output from the start compensation circuit 7 changes. Due to the change of the gradient, the timing of the first commutation operation changes, and the single-phase brushless motor 51 rotates forward or backward. This operation will be described later.

The non-inverting input terminal of the comparator CP1 has
The output terminal of the current detection circuit 4 is connected via the zero reset circuit 10, and the output terminal of the priority circuit 8 is connected to the inverting input terminal. The output terminal of the comparator CP1 is
Auxiliary power supply circuit 2 via 10 kΩ pull-up resistor R7
And the binary counter DL of the distribution circuit 9 via the analog switch AS2.
1 and the input terminal A of the monostable multivibrator MM1. Therefore, the comparator CP1
As a result, the magnitude of the output voltage of the current detection circuit 4 is compared with the magnitude of the output voltage of the priority circuit 8, and the comparison result is output to the binary counter DL of the distribution circuit 9 via the analog switch AS2.
1 and output to the monostable multivibrator MM1.
FIG. 3G shows the output voltage waveform of the comparator CP1. The high output of the comparator CP1 becomes the commutation command 65.

The gate terminal of the analog switch AS2 is connected to the switching circuit 25. Only when the switch SW2 of the switching circuit 25 is in the start mode (drive mode) ST, the analog switch AS2 is turned on, the analog switch AS3 is turned off, and the output of the comparator CP1 is output from the binary counter DL1 of the distribution circuit 9. And to the monostable multivibrator MM1.

The monostable multivibrator MM1 receives the rising pulse 65 (commutation command) of the comparator CP1 and changes the sampling command 66 (FIG. 3 (h)) to the gate (and zero reset) of the analog switch AS1 of the sampling circuit 5. Output to the circuit 10). Since the analog switch AS1 is turned on while the sampling command 66 is being output, the instantaneous output of the current detection circuit 4 is held in the capacitor C1 at the timing when the sampling command 66 falls. Therefore, the timing of extracting the instantaneous output by the sampling circuit 5 is determined by the pulse width of the sampling command 66. Therefore, when the initial rising tendency of the shunt current (FIG. 3C) weakens (usually in the range of 1/2 to 2/3 of the commutation period).
Then, the values of the resistor R8 and the capacitor C4 connected to the monostable multivibrator MM1 are determined so that the sampling command 66 ends. In this embodiment, the resistance R8 is 22 kΩ and the capacitor C4 is 470 pF.

The distribution circuit 9 switches the output to the transistors Qu, Qv, Qx, Qy of the inverter circuit 3 every time the commutation command 65 of the comparator CP1 is input,
This is a circuit for causing the single-phase brushless motor 51 to perform a commutation operation, and includes a binary counter DL1 and two inverters Iu and Iv.

The binary counter DL1 comprises a D-latch having a set terminal S and a reset terminal R. Since the inverted output terminal Q and the data input terminal D are connected, each time the input terminal C receives the commutation command 65, the binary counter DL1 inverts the output. The inverted output is
The signal is output to the inverter circuit 3 via the inverters Iu and Iv or directly. Each of the inverters Iu and Iv is composed of an open collector type NPN type digital transistor whose emitter terminal is grounded, and has a high withstand voltage. Each of the inverters Iu and Iv is an N-MOS having a source terminal grounded in place of a digital transistor.
You may make it comprise a field effect transistor. Moreover, you may comprise using a photocoupler etc. as needed.

The zero reset circuit 10 has a sampling command 66 output from the monostable multivibrator MM1.
In response to this, a circuit for resetting the output voltage of the current detection circuit 4 output to the non-inverting input terminal of the comparator CP1 to 0 volt for each commutation operation. The zero reset circuit 10 includes a resistor R11 of 100 kΩ and a resistance of 0.01 μF.
And an inverter IV2 composed of an open collector NPN digital transistor whose emitter terminal is grounded to the circuit.

One end of the resistor R11 is connected to the output terminal of the current detection circuit 4, and the other end of the resistor R11 is connected to one end of a capacitor C7 that is grounded to form an RC low-pass filter. This RC low-pass filter removes electrostatic transfer (induction) noise, electromagnetic noise, and the like. The other end of the resistor R11, that is, the output terminal of the RC low-pass filter is connected to the non-inverting input terminal of the comparator CP1 and the output terminal of the inverter IV2. It is connected to the output terminal Q of the multivibrator MM1.

Therefore, the monostable multivibrator MM1
Outputs a sampling command 66 to the non-inverting input terminal of the comparator CP1 via the inverter IV2. That is, the simulated reset is performed to 0 volt. Since the sampling command 66 is output at the same timing as the commutation command 65, although the pulse width is different, the non-inverting input terminal of the comparator CP1 is reset every time the commutation command 65 is generated.

The check wave oscillating circuit 21 outputs a check wave from 200 Hz to 2 kHz to the distribution circuit 9 in order to determine the polarity of the rotor magnet 52 in a stopped state when the brushless motor 51 starts. .

The check wave oscillating circuit 21 includes a comparator CP2. A non-inverting input terminal of the comparator CP2 is connected to a resistor R13 of 100 kΩ and an anode of a diode D3. The other end of the resistor R13 is connected to the 5-volt line of the auxiliary power supply circuit 2, and the cathode of the diode D3 is connected to the other end of the auxiliary power supply circuit 2.
A 10 kΩ resistor R15 connected to the volt line, an output terminal of the comparator CP2, and a 220 kΩ resistor R14
And connected to. The other end of the resistor R14 is connected to a capacitor C9 of 2200 pF grounded to the circuit and an inverting input terminal of the comparator CP2.

The output terminal of the check wave oscillation circuit 21 is connected to one channel terminal of the analog switch AS3. The other channel terminal of the analog switch AS3 is connected to the input terminal C of the binary counter DL1 of the distribution circuit 9, and its gate terminal is connected to the start terminal ST of the switching circuit 25. Therefore, the switching circuit 25
Is in the check mode CK,
The analog switch AS3 is turned on, and the check wave of the check wave oscillation circuit 21 is output to the binary counter DL1.

The comparison circuit 23 determines the polarity of the rotor magnet 52 that is in a stopped state when the brushless motor 51 is started, so that the armature winding is transmitted through the check wave oscillation circuit 21, the distribution circuit 9 and the inverter circuit 3. This is a circuit for determining whether or not the armature current generated by the inspection voltage applied to 53 is equal to or greater than a predetermined value.

The comparison circuit 23 includes a comparator CP3
The one end of a 1 μF capacitor C10 and one end of a 100 kΩ resistor R16 are connected to the non-inverting input terminal of the comparator CP3. Capacitor C1
The other end of 0 is connected to the output terminal of the current detection circuit 4, and the other end of the resistor R16 is connected to the inverting input terminal of the comparator CP3 and the 5-volt line of the auxiliary power supply circuit 2. Further, a 1 kΩ pull-up resistor R17 is connected to the output terminal of the comparator CP3, and its output terminal is connected to the input terminal D of the conduction direction setting circuit 24 as the output terminal of the comparison circuit 23.

The capacitor C10 and the resistor R16 constitute an AC coupling circuit using the 5 volt output of the auxiliary power supply circuit 2 as a reference voltage. Even when the output terminal voltage of the current detection circuit 4 includes a DC component, From the connection end of C10 and the resistor R16, only the AC component of the output voltage of the current detection circuit 4 appears with the 5-volt output of the auxiliary power supply circuit 2 as a reference potential. Since the inverting input terminal of the comparator CP3 is also connected to the 5-volt output of the auxiliary power supply circuit 5, the comparator CP3 outputs high only while the AC component in the output voltage of the current detection circuit 4 is on the plus side. The row is output while the AC component is on the negative side.

The monostable multivibrator MM2 outputs the output of the comparison circuit 23 to the D-latch DL of the conduction direction setting circuit 24.
2 is a circuit for determining the timing of latching. The input terminal A of the monostable multivibrator MM2 is connected to the output terminal Q of the binary counter DL1 of the distribution circuit 9, and the output terminal Q of the monostable multivibrator MM2.
The bar is connected to the input terminal C of the D latch DL2 of the conduction direction setting circuit 24. Therefore, when the Q output of the binary counter DL1 becomes high, the capacitor C11 and the resistor R1
8 for a predetermined time determined by 8
The Q output of M2 goes low. That is, the Q-bar output of the monostable multivibrator MM2 becomes high after the lapse of a predetermined time, and the output of the comparator CP3 of the comparison circuit 23 becomes the D-latch DL2 at the rising timing.
It is latched in. In this embodiment,
The capacitor C11 is 100 pF and the resistor R18 is 22 kΩ.
Has been.

The energization direction setting circuit 24 is a circuit that sets the energization direction of the first energization start wave when the single-phase brushless motor 51 is started, based on the output of the comparison circuit 23. The switch SW2 of the switching circuit 25 is in the check mode C
Immediately after switching from K to the start mode ST, the operation is performed for a short time.

The energization direction setting circuit 24 includes a D latch D
L2 and two two-input NAND circuits NA1 and NA2. As described above, the input terminal D of the D latch DL2
Is connected to the output terminal of the comparator CP3 of the comparison circuit 23, and the input terminal C is connected to the output terminal Q bar of the monostable multivibrator MM2. Also, this D latch DL
2, the set terminal S and the reset terminal R are both grounded, so that when the signal input to the input terminal C rises, the data input to the input terminal D from the output terminal Q
The inverted data is output from the output terminal Q bar.

The output terminal Q of the D latch DL2 is connected to the input terminal of one NAND circuit NA1, and the output terminal Q is connected to the input terminal of the other NAND circuit NA2. The remaining input terminals of both NAND circuits NA1 and NA2 have 100
kΩ resistor R21 and 0.001 μF capacitor C1
2 are connected. The other end of the resistor R21 is connected to the 10 volt line of the auxiliary power supply circuit 2, and the other end of the capacitor C12 is connected to the start terminal ST of the switching circuit 25. Therefore, the switch SW of the switching circuit 25
2 is set to the check mode CK, the other end of the capacitor C12 is connected to the auxiliary power supply circuit 2 via the resistor R32.
Connected to 0 volt line. When the switch SW2 of the switching circuit 25 is in the start mode ST,
The other end of the capacitor C12 is grounded.

The output terminals of both NAND circuits NA1 and NA2 are
The output terminal of one of the NAND circuits NA1 is connected to the reset terminal R of the binary counter DL1, and the output terminal of the other NAND circuit NA2 is connected to the set terminal S of the binary counter DL1. , Are connected respectively.

When the switch SW2 of the switching circuit 25 is switched from the check mode CK to the start mode ST, the differential circuit constituted by the capacitor C12 and the resistor R21 is connected to the input terminals of the NAND circuits NA1 and NA2.
After a sharp drop from 10 volts to 0 volts, a differential start pulse that sharply increases from 0 volts to 10 volts is input. During a short period of time during which the differentiated start pulse maintains the low level, energization from the energization direction setting circuit 24 to the set and reset terminals S and R of the binary counter DL1 is started based on the value latched by the D latch DL2. A signal for determining the direction of current application of the first wave is output.

The switching circuit 25 has a switch SW2 whose one end is grounded and whose other end can be switched between a start terminal ST and a check terminal CK. Check terminal CK is
One end of a resistor R31 of 10 kΩ and an analog switch AS2
And the start terminal ST
Is one end of a 10 kΩ resistor R32 and an analog switch A
It is connected to the gate terminal of S3. Both resistors R31,
The other end of R32 is connected to the 10 volt line of the auxiliary power supply circuit 2. The input terminal of the inverter IV1 is connected to the start terminal ST.
The output terminal S of V1 is connected to the input terminal S of the starting compensation circuit 7. Therefore, the switch SW2 is in the start mode S
When switched to T, a drive voltage of less than 10 volts is supplied to the start compensation circuit 7, and the start compensation circuit 7 operates.

When the switch SW2 is at the check terminal CK, a low voltage is applied to the gate terminal of the analog switch AS2 to turn it off, and a high voltage is applied to the gate terminal of the analog switch AS3 to turn it on. That is, when the switch SW2 is at the check terminal CK, the output of the check wave oscillation circuit 21 is output to the monostable multivibrator MM1 and the binary counter DL1 of the distribution circuit 9.

On the other hand, the switch SW2 is connected to the start terminal ST
, The high voltage is applied to the gate terminal of the analog switch AS2 to turn on the analog switch AS2.
A low voltage is applied to the gate terminal of S3 to turn off. That is, when the switch SW2 is at the start terminal ST, the output of the comparator CP1 is output to the monostable multivibrator MM1 and the binary counter DL1 of the distribution circuit 9.

Next, the operation of the single-phase brushless motor 51 and the sensorless drive circuit 1 configured as described above will be described. First, the operation of determining the polarity of the rotor magnet 52 when the single-phase brushless motor 51 is stopped will be described with reference to FIGS. FIG. 8 is a diagram showing each current-voltage waveform when the rotor magnet 52 of the single-phase brushless motor 51 is stopped at the position shown in FIG. 10A, and FIG. 52 is a diagram showing each current-voltage waveform when stopped at the position of FIG. 10 (b).

The switching circuit 2 determines the polarity of the rotor magnet 52.
5 is started by switching the switch SW2 of No. 5 to the check mode CK. When the switch SW2 is switched to the check terminal CK, the analog switch AS2 is turned off, the output of the comparator CP1 is cut off, and the analog switch AS3 is turned on. DL1
, A check wave of 200 Hz to 2 kHz is output.
The binary counter DL1 is activated every time the check wave rises.
The output of the distribution circuit 9 is inverted (FIG. 8D, FIG.
(D)), each transistor Qu, Q of the inverter circuit 3
v, Qx, Qy are turned on or off, and a rectangular alternating voltage (inspection voltage) is applied to the armature winding 53 of the brushless motor 51.
Is turned on.

The frequency of the inspection voltage is the same as that of the check wave (200 Hz to 2 k) output from the check wave oscillation circuit 21.
(Hz), which is Ｈｚ of 100 Hz to 1 kHz. The frequency of the inspection voltage is approximately 1.5 times or more as compared with the frequency at the time of driving being approximately 60 Hz or less. Accordingly, a sufficient armature current cannot be obtained by applying the inspection voltage, and the restraining force due to the cogging torque due to the attraction between the rotor magnet 52 and the stator core 54 cannot be exceeded. There is no rotation.

On the other hand, as described above, in the single-phase brushless motor 51 of the present embodiment, the total cross-sectional area of the pole yoke 58 connecting the main legs 57 of the stator core 54 is equal to that of the one main leg 57. It is formed to have a sectional area of about 1/10 or less. Therefore, even with a small armature current caused by the short-time application of the inspection voltage, the magnetic flux generated in the stator core
Partially saturated at 8. When the magnetic flux is saturated in the pole yoke 58 that is a part of the stator core 54, the stator performs a saturable reactor operation, and the inductance value decreases rapidly.
The armature current increases rapidly. Therefore, the polarity of the rotor magnet 52 is determined based on the sudden increase timing of the armature current.

For example, when the rotor magnet 52 is stopped at the position shown in FIG. 10A, the magnetic flux 52a is interlinked (is generated) with the stator core 54 by the rotor magnet 52. . In this state, the inspection voltage 53 is applied to the armature winding 53.
a, the stator core 54 is activated by the current 53a.
Generates a magnetic flux 54a. In the state of FIG. 10A, the magnetic flux 52a generated in the stator core 54 by the rotor magnet 52
And the magnetic flux 54a generated in the stator core 54 by the application of the inspection voltage 53a are in the same direction, that is, the addition direction. With the increase of the armature current by the application of the inspection voltage 53a, the linkage flux of the stator core 54 is increased. Since the magnetic flux 54a gradually increases, the magnetic fluxes 52a and 54a partially saturate in the inter-pole yoke 58 at an early stage. The total magnetic flux of the magnetic fluxes 52a and 54a is added by the pole yoke 58 to saturate the pole yoke 58, that is, to partially saturate a part of the stator core 54, so that the armature current sharply increases. So Figure 1
In the case of 0 (a), the timing of the rapid increase of the armature current comes earlier.

On the other hand, in the state shown in FIG. 10B, the magnetic flux 52b generated in the stator core 54 by the rotor magnet 52 and the magnetic flux 54b generated in the stator core 54 by the application of the inspection voltage 53b, ie, in the opposite direction. Since the direction is the subtraction direction, the two magnetic fluxes 52b, 54b are determined by the sum of the two magnetic fluxes 52b, 54b.
Is partially saturated by the gap yoke 58 in FIG.
It is slower than the case of Therefore, in the case of FIG. 10B, the timing of the rapid increase of the armature current comes later.

As described above, the timing at which the armature current suddenly increases (increases speed) when the inspection voltages 53a and 53b are energized differs depending on the stop position of the rotor magnet 52. The polarity of the slave magnet 52 can be determined. Specifically, the inspection voltage 53
When a predetermined time has elapsed after the start of energization of a and 53b, it is determined whether the timing of the sudden increase of the armature current is early or late based on whether the armature current has reached a predetermined value.

As shown in FIG. 8D and FIG.
When a high signal is output from the Q output of the binary counter DL1 and a low signal is output from the Q bar output, the transistors Qu and Qy of the inverter circuit 3 are turned on and the transistors Qv and Qx are turned off based on the output. Thus, when the inspection voltage is applied to the brushless motor 51, an armature current flows through the armature winding 53 (FIGS. 8A and 9A). The armature current flows through the shunt resistor Rs of the current detection circuit 4 (shunt current) (FIGS. 8B and 9).
(B)), converted into a voltage and output to the comparison circuit 23. In the comparison circuit 23, the comparator CP3
The AC component of the output voltage of the current detection circuit 4 is
5 volt reference voltage. As a result of the comparison, if the AC component of the output voltage of the current detection circuit 4 is higher than the reference voltage of 5 volts, a high output is output from CP3. Conversely, if the output voltage of the current detection circuit 4 is lower, CP3 is output.
(FIG. 8 (c), FIG. 9 (c)).

Since the Q output of the binary counter DL1 is also input to the input terminal A of the monostable multivibrator MM2, a predetermined time determined by the capacitor C11 and the resistor R18 after the rising of the Q output of the binary counter DL1. Thereafter, the Q bar output of the monostable multivibrator MM2 rises (FIGS. 8 (e) and 9 (e)). The output of the comparator CP3 at the rise of the Q bar output is
It is latched by the D-latch DL2 of the conduction direction setting circuit 24. The latched data is output from the Q output of the D latch DL2, and the inverted data is output from the Q bar output.
A1 and NA2 (FIG. 8 (f), FIG. 9)
(F)).

As shown in FIG. 8, when the rotor magnet 52 is stopped in the state shown in FIG.
a, the magnetic flux 54 is applied to the stator core 54 in the adding direction.
a interlinks (occurs). Therefore, the magnetic flux of the stator core 54 saturates at an early stage, and the timing of rapid increase of the armature current comes at an earlier stage. Therefore, when the output of the Q bar of the monostable multivibrator MM2 rises, the comparator CP3 has a high output. In such a case, the Q output of the D latch DL2 is high (FIG. 8 (f)).
The bar output goes low.

As shown in FIG. 9, when the rotor magnet 52 is stopped in the state shown in FIG. 10B, the magnetic flux 54b is applied to the stator core 54 in the subtraction direction by the application of the inspection voltage 53b. Interlinks (occurs). Therefore, the stator core 5
The saturation time of the magnetic flux of No. 4 is later than that in the case of FIG. 10A, and the timing of the rapid increase of the armature current also comes later. Therefore, when the output of the Q bar of the monostable multivibrator MM2 rises, the comparator CP3 remains at the low output, so that the Q output of the D latch DL2 is low (FIG. 9 (f)) and the Q bar output is high. is there.

The energization direction setting circuit 24 waits for an output while the switch SW2 of the switching circuit 25 is in the check mode CK. At the moment when the switch SW2 is switched from the check mode CK to the start mode ST, a signal for setting the energizing direction of the first wave is output from the energizing direction setting circuit 24 to the set terminal S and the reset terminal R of the binary counter DL1. . According to this signal, the first energization start wave is set in a direction in which the magnetic flux generated in the stator core 54 by the energization is added to (the same direction as) the magnetic flux of the rotor magnet 52 (see FIG. 10A). Thus, by energizing the first wave in the addition direction (by energizing the first wave so as to generate a polarity (a depolarization direction) opposite to the polarity of the rotor magnet 52), the rotor magnet 52 Can be started by applying a large torque to the motor.

More specifically, in the case of FIG. 10A, high is output from the NAND circuits NA1 and NA2 of the conduction direction setting circuit 24 to the set terminal S of the binary counter DL1, and low is output to the reset terminal R. Therefore, the first wave of the energization start is set such that the Q output of the binary counter DL1 is high and the Q bar output is low. On the other hand, in the case of FIG. 10B, the NAND circuits NA1 and NA2 of the energization direction setting circuit 24
Low is output to the set terminal S of the binary counter DL1, and high is output to the reset terminal R.
The first energization start wave is set so that the output is low and the Q bar output is high.

When the switch SW2 of the switching circuit 25 is switched to the start mode ST, the analog switch AS3 is turned off, the output of the check wave oscillating circuit 21 is cut off, and the analog switch AS2 is turned on instead. The output of the cut-off comparator CP1 is output to the binary counter DL1 of the distribution circuit 9 and the monostable multivibrator MM1. Therefore, thereafter, every time the commutation timing is detected by the comparator CP1,
A high commutation command 65 is output from the comparator CP1 to the binary counter DL1 and the monostable multivibrator MM1 to perform a commutation operation, and the single-phase brushless motor 51
Is driven.

Next, the operation of the starting compensation circuit 7 for determining the rotation direction of the brushless motor 51 will be described with reference to FIGS. As described above, the starting compensation circuit 7
Is the comparator C when the brushless motor 51 is started.
The rotation direction of the brushless motor 51 is determined by changing the decreasing gradient of the commutation target voltages 60 and 60 ′ supplied to P1. The decreasing gradient of the commutation target voltages 60 and 60 'can be switched in two ways by turning on or off the switch SW1. The brushless motor 51 includes a switch SW
When the switch 1 is off, it rotates in the forward direction (FIGS. 4 and 5), and when the switch SW1 is turned on, it rotates in the reverse direction (FIGS. 6 and 7).

Referring to FIGS. 4 and 5, switch SW1
A description will be given of the forward rotation operation when the switch is off. As a result of the polarity determination of the rotor magnet 52, the first wave of the energization start is performed such that a magnetic field is generated in the direction of addition of magnetic flux (direction of depolarization), that is, as shown in FIG. When the energization of the first wave is started, the rotor magnet 52 moves in the direction of arrow R,
That is, rotation starts in the forward direction. Since the armature current changes with this rotation, the output voltage 63 of the current detection circuit 4 becomes as shown at the point D.

Thereafter, when the rotor magnet 52 further rotates (FIG. 5B), the output voltage 63 of the current detection circuit 4 also becomes the point E.
To the state of. When the rotor magnet 52 rotates to the position shown in FIG. 5C, the output voltage 63 of the current detection circuit 4 changes to the state at the point F, and the commutation target voltage 60 output from the starting compensation circuit 7 Exceeds the value of. Then, the high signal 65 (commutation command) is output from the comparator CP1 (FIG. 3G), and the commutation operation is performed. Since the direction of energization to the armature winding 53 is switched by the commutation, the direction of the magnetic field applied to the rotor magnet 52 becomes the direction of arrow B (FIG. 5D), and thereafter, the forward direction (arrow) (R direction) rotation is continued.

When the switch SW1 is turned off, even if the rotor magnet 52 is stopped at a position rotated by 180 degrees from the state shown in FIG. Since it is performed in the direction of adding magnetic flux (the direction of depolarization),
As in the case of FIG. 5, the motor 51 rotates in the forward direction (the direction of the arrow R). Thus, when the switch SW1 of the starting compensation circuit 7 is turned off, the brushless motor 51 always rotates in the forward direction.

Referring to FIGS. 6 and 7, switch SW1
The reverse rotation operation when is turned on will be described. As a result of the polarity determination of the rotor magnet 52, the first wave of the energization start is performed so that a magnetic field is generated in the direction of addition of magnetic flux (direction of depolarization), that is, as shown in FIG. When the energization of the first wave is started, the rotor magnet 52 moves in the direction of arrow R,
That is, rotation starts in the forward direction. Since the armature current changes with this rotation, the output voltage 63 ′ of the current detection circuit 4 is changed.
Is shown at point G.

Thereafter, when the rotor magnet 52 further rotates (FIG. 7B), the output voltage 63 'of the current detection circuit 4 also changes to the state of the point H. Then, when the rotor magnet 52 rotates to the position shown in FIG. 7C, the output voltage 63 ′ of the current detection circuit 4 changes to the state at the point I. Since the capacitance of the capacitor of the start compensation circuit 7 when the switch SW1 is on is larger than that when the switch SW1 is off, the commutation target voltage 60 ′ when the switch SW1 is on is
At this time, the value is sufficiently larger than the point I. Therefore, the commutation operation is not performed at the point I, and the rotor magnet 5
7 reaches the state of FIG. 7D, stops the forward rotation, and starts rotating in the reverse direction (the direction of the arrow L).

When the reverse rotation proceeds, the current detection circuit 4
Also changes to the state at the point J. Then, when the rotor magnet 52 reversely rotates to the position shown in FIG. 7E, the output voltage 63 'of the current detection circuit 4 reaches the point K. Only at this time, the commutation target output from the starting compensation circuit 7 is started. Exceeds the value of voltage 60 '.

Therefore, at the point K, the comparator CP1 outputs the high signal 65 (commutation command) (FIG. 3 (g)), and the commutation operation is performed. Due to the commutation, the direction of energization to the armature winding 53 is switched, and the rotor magnet 52
The direction of the magnetic field applied to the arrow B becomes the direction of arrow B (FIG.
(F)), the rotation of the brushless motor 51 in the reverse direction (the direction of the arrow L) is maintained.

When the switch SW1 is ON, even if the rotor magnet 52 is stopped at a position rotated by 180 degrees from the state shown in FIG. Since it is performed in the direction of adding magnetic flux (the direction of depolarization),
As in the case of FIG. 7, the motor 51 first makes a half rotation in the forward direction, then reverses and rotates in the reverse direction (the direction of the arrow L).
Thus, when the switch SW1 of the starting compensation circuit 7 is ON, the brushless motor 51 always rotates in the reverse direction.

Finally, the commutation operation of the single-phase brushless motor 51 during steady operation will be described. In the steady operation, the switch SW2 of the switching circuit 25 is set in the start mode S
Since the operation is performed in the state of T, the analog switch AS2 is on and the AS3 is off. Therefore, instead of the output of the check wave oscillation circuit 21, the comparator CP
1 is output to the binary counter DL1 of the distribution circuit 9 and the monostable multivibrator MM1.

The current detection circuit 4 is connected to the comparator CP1.
And the output voltage of the sampling circuit 5 which has been amplified approximately 1.8 times by the amplifier circuit 6 is output. The two output voltages are compared by the comparator CP1, and when the output voltage of the current detection circuit 4 becomes larger than the output voltage of the amplifier circuit 6, a high signal 65 (commutation command) is output from the comparator CP1 to the binary counter DL1 and the monostable multi-stage. Output to vibrator MM1 (FIG. 3 (g)).

The binary counter DL1 of the distribution circuit 9 receives the rising pulse of the commutation command 65 and inverts each output. As a result, the on / off states of the transistors Qu, Qv, Qx, Qy of the inverter circuit 3 are inverted, and
The direction of current supply to the armature winding 53 of the single-phase brushless motor 51 is switched (FIG. 3A), and a commutation operation is performed. By this commutation operation, an armature current in the opposite direction flows through the armature winding after the current recirculation action of the armature winding ends (FIG. 3B). The armature current flows through the shunt resistor Rs of the current detection circuit 4 (FIG. 3C), and is converted into a voltage.
As an output voltage of the current detection circuit 4, the comparator CP1
And output to the sampling circuit 5 (FIG. 3D).

On the other hand, the high signal 6 is output from the comparator CP1.
5, the monostable multivibrator MM1 receives the rising pulse of the high signal 65 and outputs a one-shot sampling command 66 (FIG. 3).
(H)). The sampling command 66 is output to the reset circuit 10 and resets the voltage level of the non-inverting input terminal of the comparator CP1 to 0 volt. Since the high signal 65 of the comparator CP1, that is, the commutation command, is output when the input voltage at the non-inverting input terminal becomes larger than the input voltage at the inverting input terminal, every time the high signal 65 (commutation command) is generated. By resetting the input voltage at the non-inverting input terminal, the output 65 (commutation command) of the comparator CP1 and the sampling command 66 can be reliably reset for each commutation operation.

The sampling command 66 is output to the sampling circuit 5. In response to the sampling command 66, the analog switch AS1 of the sampling circuit 5 is turned on, and the output voltage of the current detection circuit 4 is changed to the capacitor C1.
Is input to The sampling command 66 is set so that the shunt current (FIG. 3 (c)) ends at a position where the rising tendency is once weakened (usually in the range of 1/2 to 2/3 of the commutation period). The output voltage of the current detection circuit 4 is stored in the capacitor C1 at the falling timing of the sampling command 66 (off timing of the analog switch AS1) (FIG. 3E). The voltage value stored in the capacitor C1 is output to the amplifier circuit 6 as an output voltage of the sampling circuit 5, amplified by approximately 1.8 times by the amplifier circuit 6, and output to the comparator CP1 (FIG. 3 (f)). ).

The comparator CP1 compares the output voltage of the amplifier circuit 6 (FIG. 3 (f)) with the output voltage of the current detection circuit 4 (FIG. 3 (d)). Is larger, the high signal 6 which is a commutation command
5 is output to the distribution circuit 9 and the monostable multivibrator MM1 (FIG. 3 (g)). Thereafter, based on the high signal 65 (commutation command), the above-described operation is repeated, and the commutation operation is performed. Thereby, the single-phase brushless motor 51
Is continuously rotated.

The present invention has been described based on the embodiments. However, the present invention is not limited to the above embodiments, and various improvements and modifications can be made without departing from the gist of the present invention. Can easily be inferred.

For example, in the single-phase brushless motor 51 of the present embodiment, as shown in FIG. 1, the stator core 54 including the main leg portion 57 and the yoke 58 between the poles is integrally formed of the same core. However, as shown in FIG. 11, the pole yoke may be made of ferrite 59 having a square magnetization characteristic. Such ferrite 59
, The saturation magnetic flux density is about four times lower than that of the iron core, and the saturation of the magnetic flux occurs rapidly. Therefore, as shown in FIG.
By using the ferrite 59 for the pole yoke, the total cross-sectional area of the pole yoke 59 can be reduced to approximately four-tenths or less of the cross-sectional area of one main leg 57. Therefore,
Compared to the case where the pole yoke 59 is formed of an iron core, the cross-sectional area can be increased and the inductance change can be increased, so that the magnetic pole position when the single-phase brushless motor 151 is stopped is correspondingly increased. The determination can be made easily. Also, the mechanical strength can be improved.

In FIGS. 1 and 11, the gap yoke 5
Each of the pair 8 and 59 is formed as a pair (two). However, as long as the main legs 57 are connected, a single pole yoke may be formed. When one pole yoke is formed, the cross-sectional area of the single pole yoke is approximately one-tenth or less of the cross-sectional area of the minimum portion of the main leg (the pole yoke is a square-shaped yoke). Any ferrite having magnetizing characteristics may be used if it is about 4/10 or less. When the pole yoke is constituted by one, the single-phase brushless motor has one closed slot structure,
The other has an open slot structure.

Further, two positions that maintain a predetermined declination θ from the locked position where the magnetic flux axis X of the rotor magnet 52 and the magnetic flux axis Y generated in the stator core 54 by energizing the armature winding 53 coincide with each other. As a configuration for stopping the rotor magnet 52, the magnetic pole 55 of the stator core 54 may be formed in an elliptical shape instead of the notch 56, or an auxiliary magnet with a weak magnetic force may be provided in the stator core 54. You may do it. The armature winding 54 may have a meson tap as long as it is a single winding.

[0100]

According to the single-phase brushless motor and the magnetic pole position determining circuit of the single-phase brushless motor of the present invention, the magnetic pole position of the rotor magnet is determined without rotating the rotor magnet when the motor is stopped. There is an effect that can be. Therefore, the single-phase brushless motor can be quickly started in a desired direction. Also, by managing the number of steps when driving the single-phase brushless motor, the amount of rotation of the rotor magnet can be grasped,
The rotation amount can be used for various controls.

[Brief description of the drawings]

FIG. 1 is a sectional view of a single-phase brushless motor according to an embodiment of the present invention. (A) shows a state where the N pole of the rotor magnet is located on the left side and stops, and (b) shows a state where the N pole of the rotor magnet is located on the right side and stops.

FIG. 2 is a circuit diagram of a sensorless drive circuit of a single-phase brushless motor.

FIG. 3 is a diagram illustrating voltage and current waveforms when a single-phase brushless motor is driven. (A) is a figure which showed the voltage waveform applied to an armature winding. (B) is a diagram showing a waveform of a current flowing to the armature winding due to the application of the voltage. (C) is a diagram showing a waveform of a current flowing through the shunt resistor of the current detection circuit. (D) is a diagram showing an output voltage waveform of the current detection circuit. (E) is a diagram showing an output voltage waveform of the sampling circuit. (F) is a diagram showing an output voltage waveform of the amplifier circuit. (G) is C
FIG. 3 is a diagram showing an output voltage waveform of P1. (H) is MM
FIG. 3 is a diagram illustrating an output voltage waveform of FIG.

FIG. 4 is a diagram showing C in an OFF state of SW1 of a start compensation circuit.
FIG. 4 is a diagram showing an output voltage waveform to P1.

FIG. 5 is a diagram showing a rotation state of a rotor magnet in a state where SW1 of a start compensation circuit is off. (A) is a figure showing a starting state of forward rotation. (B) is a diagram showing a state where the forward rotation has advanced. (C) is a diagram showing a state in which the forward rotation is further advanced. (E) is a diagram illustrating a state in which a commutation operation is performed in accordance with further forward rotation in the forward direction.

FIG. 6 shows C in the ON state of SW1 of the start compensation circuit.
FIG. 4 is a diagram showing an output voltage waveform to P1.

FIG. 7 is a diagram illustrating a rotation state of a rotor magnet when an SW1 of a start compensation circuit is in an ON state. (A) is a figure showing a starting state of forward rotation. (B) is a diagram showing a state where the forward rotation has advanced. (C) is a diagram showing a state in which the forward rotation is further advanced. (D)
FIG. 9 is a diagram illustrating a state in which the rotation proceeds to the reverse rotation as a result of the further forward rotation. (E) is a diagram showing a state in which the reverse rotation has advanced. (F) is the figure which showed the state where the commutation operation | movement was performed according to the further reverse rotation.

FIG. 8 is a diagram showing each current-voltage waveform at the time of determining the polarity of the rotor magnet in the state of FIG. 10 (a). (A)
FIG. 3 is a diagram showing a waveform of a current flowing to an armature winding.
FIG. 3B is a diagram illustrating a waveform of a current flowing through a shunt resistor of the current detection circuit. (C) is a diagram showing an output voltage waveform of the comparison circuit. (D) is a diagram showing an output voltage waveform at the output terminal Q of DL1. (E) is a diagram showing an output voltage waveform at the output terminal Q of MM2. (F)
FIG. 7 is a diagram showing an output voltage waveform at an output terminal Q of DL2.

FIG. 9 is a diagram showing each current-voltage waveform at the time of determining the polarity of the rotor magnet in the state of FIG. 10B. (A)
FIG. 3 is a diagram showing a waveform of a current flowing to an armature winding.
FIG. 3B is a diagram illustrating a waveform of a current flowing through a shunt resistor of the current detection circuit. (C) is a diagram showing an output voltage waveform of the comparison circuit. (D) is a diagram showing an output voltage waveform at the output terminal Q of DL1. (E) is a diagram showing an output voltage waveform at the output terminal Q of MM2. (F)
FIG. 7 is a diagram showing an output voltage waveform at an output terminal Q of DL2.

FIG. 10A is a diagram showing a state of a magnetic flux generated in a stator core when energizing an armature winding in a state where the N pole of the rotor magnet is positioned on the left side and stopped. is there. FIG. 4B is a diagram illustrating a state of a magnetic flux generated in the stator core when the armature winding is energized in a state where the N pole of the rotor magnet is positioned on the right side and stopped.

FIG. 11 is a sectional view of a single-phase brushless motor according to a second embodiment.

[Explanation of symbols]

REFERENCE SIGNS LIST 1 brushless motor drive circuit 3 inverter circuit (part of test voltage supply circuit) 4 current detection circuit 9 distribution circuit (part of test voltage supply circuit) 21 check wave oscillation circuit (part of test voltage supply circuit) 23 comparison circuit (Part of the polarity discrimination circuit) 24 Energization direction setting circuit (part of the polarity discrimination circuit) 51 Single-phase brushless motor 52 Rotor magnet 53 Armature winding 54 Stator core 56 Notch (notch) 57 Main leg 58,59 Pole yoke MM2 Monostable multivibrator (part of polarity discrimination circuit)

Claims (6)

[Claims] A stator core on which an armature winding is wound;
In a single-phase brushless motor comprising a rotor magnet and the rotor magnet being stopped at a predetermined angle from the lock position, the stator core includes a pair of main legs, and a pair of main legs. An inter-pole yoke connecting the legs, wherein the cross-sectional area of the inter-pole yoke is approximately 1 times the cross-sectional area of the main leg.
A single-phase brushless motor characterized in that it is formed to be 1/0 or less. 2. The stator core includes the pair of main legs and a pair of interpole yokes connecting the pair of main legs at two different locations, respectively. And a pair of pole yokes, the rotor magnet is enclosed in a closed slot structure, and the total cross-sectional area of the pair of pole yokes is approximately one tenth of one cross-sectional area of the main leg. The single-phase brushless motor according to claim 1, wherein the motor is formed as follows. 3. The inter-electrode yoke is made of ferrite having a square magnetizing characteristic, and the cross-sectional area of the inter-electrode yoke made of the ferrite is approximately 4/10 or less of the cross-sectional area of the main leg. The single-phase brushless motor according to claim 1 or 2, wherein the single-phase brushless motor is formed as follows. 4. A notch formed in the stator core for stopping the rotor magnet at a predetermined angle from a lock position, wherein the pole yoke is adjacent to the notch. The single-phase brushless motor according to any one of claims 1 to 3, wherein the single-phase brushless motor is formed. 5. An inspection voltage applying circuit for applying an inspection voltage to the armature winding, a current detecting circuit for detecting a current flowing to the armature winding by applying an inspection voltage by the inspection voltage applying circuit, 5. A polarity discriminating circuit for discriminating a magnetic pole position when the rotor magnet is stopped based on an increasing speed of the armature current detected by the current detecting circuit. A magnetic pole position discriminating circuit of the single-phase brushless motor described in the above. 6. The magnetic pole of a single-phase brushless motor according to claim 5, wherein the test voltage energizing circuit outputs an alternating voltage having a frequency higher than a frequency at the time of driving the single-phase brushless motor as a test voltage. Position determination circuit.