HU223242B1 - Despreading of direct sequence spread spectrum communications signals - Google Patents

Abstract

The present invention relates to a correlator for processing a direct sequentially distributed telecommunication signal comprising a single phase component [I (t)] and a perpendicular phase component [Q (t)]. The control unit (155) of the correlator of the present invention rotating the phase of the complex scattering sequence [s (k)] of the received signal, outputing a first control sequence and a second control sequence, receiving a single phase component [I (t)] of the received signal and negating the first control sequence in a controlled manner (150i), and a second negating unit (150q) receiving the received orthogonal phase component [Q (t)] and negating in a manner controlled by the first control sequence, and receiving the same phase and perpendicular phase signals from the first and second negating units (150i, 150q). having phase and perpendicular phase signals in a manner controlled by the second control sequence has an exchange unit (152). The invention further relates to a method for processing a direct sequence spread spectrum received telecommunication signal having a single-phase component [I (t)] and a perpendicular phase component [Q (t)] to remove the complex sequence. In the method according to the invention, the phase of the complex scattering sequence [s (k)] is processed for the received signal, and the same-phase component [I (t)] of the received signal is negated. At the same time, the perpendicular phase [Q (t)] component of the received signal is negated, and the output signals of the same phase and perpendicular phase are exchanged in a manner controlled by the processed phase. Another object of the present invention is to provide a correlator for processing a direct sequence spread spectrum telecommunication signal having a logarithmic amplitude component phase component. The phase offset phase extractor of the received signal complex scattering sequence [s (k)], the sum of the received signal phase component and the extracted phase offset of the complex scattering sequence [s (k)] and the summed phase output, and the logarithmic amplitude of the received signal has a means for forming a logarithmic range complex signal. The present invention also relates to a method for processing a direct sequence spread spectrum received signal having a logarithmic amplitude component and a phase component. In this method, the phase offset of the complex scattering sequence [s (k)] of the received signal is obtained, the received phase phase component and the recovered phase offset of the complex scattering sequence [s (k)] are summed, and the summed phase is output, and the logarithmic amplitude component of the received signal and the summed f form a complex signal with a logarithmic range. The invention further relates to a further similar method in which the phase offset of the scattering sequence [s (k)] of the received signal complex is obtained, components with allogarithmic amplitude are collected, the received phase phase component and the complex scattering sequence [s (k)] are collected.

Description

FIELD OF THE INVENTION The present invention relates to a correlator for processing a direct-sequence spread spectrum received telecommunication signal comprising a single-phase component [I (t) j and a perpendicular phase component [Q (t) j. The correlates of the present invention comprise a control unit (155) for rotating the complex scattering sequence [s (k) j of the received signal, outputting the first control sequence and the second control sequence, receiving the same phase component of the received signal [I (t) j]. a first negation unit (150i) of negating and receiving a perpendicular phase component [Q (t) j of the received signal and a second negating unit (150q) negatively controlled by the first control sequence, and the first and second negating units (150i, 150q) a switching unit (152) for receiving phase and perpendicular phase signals, which exchanges the same phase and perpendicular phase signals in a manner controlled by the second control sequence.

The present invention also relates to a method for processing received direct-sequence spread spectrum telecommunication signals having the same phase component [I (t)] and the perpendicular phase component [Q (t)] to remove the complex sequence. In the method according to the invention, the received signal is processed with the [s (k)] phase of the complex scattering sequence and negated with the same phase component [I (t) j of the received signal. At the same time, the perpendicular-phase [Q (t)] component of the received signal is negated, and the same-phase and perpendicular-phase output signals are exchanged in a manner controlled by the processed phase.

The present invention further relates to a correlator for processing a direct-to-sequence received broadcast signal having a logarithmic amplitude component and a phase component. My correlates are a

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HU 223 242 B1

The length of the description is 16 pages (including 5 pages)

EN 223 242 B1 Phase converter extracting the phase offset [s (k)] of the received scatter signal, summing the phase component of the received signal and the extracted phase offset of the complex scattering sequence [s (k) j, and outputting the sum phase, and logarithmic amplitude of the received signal and means for forming a logarithmic complex signal containing its component and summed phase.

The present invention also relates to a method for processing direct-to-sequence transmitted-spectrum telecommunication signals having a logarithmic amplitude component and a phase component. In this method, the phase offset [s (k)] of the complex scattering sequence of the received signal is summed, the phase component of the received signal and the extracted phase offset of the complex scattering sequence [s (k) J are summed and output and the logarithmic amplitude component of the received signal and forming a logarithmic complex signal containing the summed phase.

The invention further relates to a similar method in which the phase offset [s (k) j of a complex scattering sequence of a received signal is obtained, the components of logarithmic amplitude are collected, the phase component of the received signal and the extracted phase offset of s (k) J , and emit an absolute value aggregate phase. The output absolute phase summed phase is collected and normalized to form a logarithmic complex signal containing the logarithmic amplitude component of the received signal and the collected normalized absolute phase summed phase.

In a similar procedure, the phase offset of the complex scattering sequence of the signal received in the signal is obtained and the components of logarithmic amplitude are collected. The phase component of the received signal and the extracted phase offset [s (k) j of the complex scattering sequence are summed, and an absolute value summed phase is output. The summed phases emitted are circularly averaged and the phases collected are emitted. They form a logarithmic complex signal containing the logarithmic amplitude component of the received signal and the collected phase.

The present invention relates to spread-spectrum telecommunications systems, and in particular to the recovery of so-called direct-sequence spread-spectrum telecommunications signals.

The mobile phone industry has seen tremendous business growth in both the United States and most of the world. Growth in major metropolitan areas has significantly exceeded expectations and exceeded system capacity. If this trend continues, the effects of rapid growth will soon reach even the smallest markets. One of the major problems of continuous growth is that the electromagnetic spectrum available to mobile phone service providers remains fixed while the customer base grows. Innovative solutions are needed to meet the demands of increasing capacity 40 while maintaining good quality service and not increasing prices.

Currently, channel access is primarily provided using FDMA (Frequency Division Multiple Access) and TDMA (Time Division Multiple Access 45) methods. In frequency division multiple access systems, the communication channel is a simple radio frequency band in which the signal transmission power is concentrated. In time division multiple access systems, a channel is a time slot in a periodic series of time intervals broadcast on the same radio frequency.

Spread spectrum (so-called spread spectrum) is a communication technique used commercially in wireless communication. Spread spectrum systems have been known since the Second World War. Its early applications were predominantly military (related to radar and intelligent radio interference). However, there is a growing interest today in commercial applications using spread spectrum systems, including digital cellular radio, terrestrial mobile radio and indoor and outdoor personal communication systems.

In a broadcast spectrum transmitter, a digital bitstream is scattered or spread from a base data rate to a broadcast data rate. This spreading operation applies a unique user digital code (a spread or tag sequence) to the bit stream, which increases its data rate while rendering it redundant. This application multiplies the digital bitstream with the digital code [or applies a logical XOR (Exclude OR) operation], The output result data sequence (called chip) is then used to create the output signal, so called QPSK, or Perpendicular Phase Shift Modulation or Quadrature Phase Keyboard ( QPSK = quadrature phase shift keying). This output signal is added to other similarly processed output signals and is transmitted over the telecommunications medium through multiple channels. The output signals of the plurality of users (channels) preferably share a telecommunication transmission frequency, with seemingly nested multiple signals, both in the frequency domain and in the time domain. Since the digital codes used are unique to each user, the output signals transmitted over the shared telecommunication frequency are also unique and can be distinguished by the appropriate processing techniques used in the receiver. In the spread spectrum receiver, the received signals are demodulated and the appropriate digital code of the affected user is used to recover the signal (for example, by multiplying the signal with the code, i.e., decoding the desired broadcast signal and returning to base data rate). When this digital code is applied to other broadcast and received signals, then

There is no reset and the signal retains the so-called chip rate. The reset operation is thus in fact a correlation procedure that compares the received signal with the corresponding digital code.

In many spread spectrum communication systems, the transmitted data series comprises two components: the in-phase or same-phase (1) component and the perpendicular-phase (Q) component. These components are generally considered to be real and imaginary parts of the complex signal. In the spread spectrum transmitter, a complex scattering sequence is used and the two components are modulated (perpendicular phase shift or phase shift modulation according to the procedure) and combined to form the output signal to be emitted. Since the received signal contains both the same phase component and the perpendicular phase component, the reset operation performed by the scattered receiver must correlate the received complex signal with the corresponding digital code (tag sequence). This is typically accomplished by two scalar correlators, one of which is provided with in-phase pulse sequences and the other with inputs of perpendicular-phase pulses. When complex scatter sequences are used, four scalar correlators are required, which, however, further complicates the correlation procedure.

In mobile communication systems, signals transmitted between two locations may suffer from echo distortion and time dispersion (dispersion). A multipath dispersion occurs when the signal is received by the receiver via multiple paths, rather than one, with the receiver receiving many different echoes with varying delay and amplitude. For example, this is typically caused by reflections from nearby mountain ranges or large buildings. When multiple time dispersions appear in a broadcast spectrum telecommunication system, the received signal is a mixture of several variants (or images) of the transmitted signal, which are transmitted along different paths (so-called rays). These versions of the broadcast signal are typically delayed relative to one another for periods less than one code signal. Macro-level changes and delayed cell switching during a call (so-called soft handoff) may cause greater delays.

The presence of multiple time dispersions in the case of a complex-transmitted scattered signal significantly complicates the reception and correlation procedures. In this case, for example, a so-called RAKE (comb or rake) receiver can be used to receive multiple beams of a signal (the name derives from the fact that it combines multiple contributions by applying a weighted amount). In this case, there is a correlation device for each image of the broadcast signal (a correlation device includes a single phase correlation portion and a perpendicular phase correlation portion). Each correlation device is tuned to the corresponding signal (beam) with a delay line. The received and time scattered signals are then weighted for each correlation device in proportion to the amplitude of each received signal, and the resulting signals are summed (by the same phase and perpendicular phase) and output to further processing.

Note that in some systems, the entire scattering sequence may in fact comprise a combination of multiple sequences. For example, in CDMA code division multiple access (CDMA code) telecommunications, as defined in the TIA IS-95 digital standard, downlink (network node-to-mobile) information is represented by a real tag sequence. The information is further encoded by same-phase and perpendicular-phase coding sequences (called scrambling). Therefore, the entire scattering sequence is the result of a mixture of complex coding sequences above the real scattering sequence. Applying known techniques, the receiver requires complex correlators that require sophisticated instrumentation. The complexity increases further when multiple channels (such as traffic and control) need to be reset at the same time.

Therefore, less sophisticated correlation arrangements are required when processing complex spread spectrum telecommunication signals that suffer from multipath time dispersion with complex broadcast and multichannel reception.

EP 0,658,985 (Sato) discloses a CDMA receiver. The receiver has a common signal processing unit and a plurality of channel signal processing units. Each of the beacon and channel signal processing units receives and processes the received signal. The common signal processing unit processes the received signal so that it can calculate the common values required for each channel signal processing unit to demodulate the spread spectrum. The channel signal processing unit then uses the calculated values to perform the necessary demodulation.

The present invention provides simpler arrangements for restoring direct sequence spread spectrum telecommunication signals. The correlator for receiving and processing the transmitted direct sequence spread spectrum telecommunication signals generally comprises a complex sequencing remover and a collecting and discharging unit.

The present invention relates, on the one hand, to a correlator for processing a direct-sequence spread spectrum received telecommunication signal comprising a single phase component and a perpendicular phase component. A control unit for transmitting the first control sequence and the second control sequence of a unit of a direct sequence spread spectrum received telecommunication signal, transmitting a first control sequence and a second control sequence, receiving a first control sequence negating the same phase component of a direct sequence spread spectrum received telecommunications signal, and a second negation unit receiving the perpendicular phase component of the direct sequence spread spectrum received telecommunication signal and negating in a manner controlled by the first control sequence, and receiving the same phase and perpendicular phase signals from the first and second negation units, controlled by a switching unit.

HU 223 242 Bl

The present invention also relates to a method for processing a direct-sequence spread spectrum received telecommunication signal having the same phase component and the perpendicular phase component to remove the complex sequence. The method of the present invention comprises the steps of: processing a phase of the complex scattering sequence for a direct sequence spread spectrum received telecommunication signal; negating the same phase component of a direct sequence spread spectrum received telecommunications signal; and exchanging the same phase and perpendicular phase output signals in a manner controlled by the processed phase.

The present invention also relates to a correlator for processing a direct-sequence spread spectrum received telecommunication signal having a logarithmic amplitude component and a phase component. A phase converter for extracting a phase offset of a complex spreading sequence of a direct sequence spread spectrum received telecommunication signal, a phase component of a direct sequence spread spectrum received telecommunication signal, and summing the sum of the phase outputs of a complex spreading sequence, and means for forming a logarithmic amplitude component of a received telecommunication signal with a logarithmic amplitude and a summed phase.

The present invention also relates to a method for processing a direct-sequence spread spectrum received telecommunication signal having a logarithmic amplitude component and a phase component, which also aims to remove the complex sequence. This process according to the invention comprises the following steps:

extracting the phase offset of the complex scattering sequence of the direct sequence scattered received telecommunication signal, summing the phase component of the received direct sequence scattered received signal and the extracted phase offset of the complex spreading sequence, and outputting the sum and forming a logarithmic complex signal containing the summed phase.

The present invention also relates to a further method for processing a direct-sequence spread spectrum received telecommunication signal having a logarithmic amplitude component and a phase component to remove the complex sequence. The latter process according to the invention comprises the following steps:

extracting the phase scatter of the complex scattering sequence of the received sequential spread spectrum telecommunication signal; summing the phase and normalizing it to form a logarithmic complex signal containing the collected logarithmic amplitude component of the direct sequence scattered received telecommunication signal and the collected normalized absolute value summed phase.

The present invention also relates to a method for processing a direct-sequence spread spectrum received telecommunication signal having a logarithmic amplitude component and a phase component to remove the complex sequence. The latter process according to the invention comprises the following steps:

extracting the phase scatter of the complex scattering sequence of the received sequential spread spectrum telecommunication signal; circulating and averaging the collected phases, and forming a logarithmic amplitude component of the received telecommunication signal of direct sequence spread spectrum and a logarithmic complex signal containing the collected phase.

The methods and apparatuses of the present invention will be better understood from the following detailed description and drawings, wherein:

Figure 1 is a block diagram of a spread spectrum telecommunication system, Figure 2 is a block diagram of a reset correlator used in a spread spectrum telecommunication system, Figure 3 is a block diagram of a spread spectrum telecommunication system shown in Figure 1 and illustrated in Figure 2 Fig. 4A is a more detailed block diagram of a correlator; Fig. 4A is a block diagram illustrating a first embodiment of an improved complex sequencing unit of the present invention; 5B and 5B. Figure 6 is a block diagram showing a second embodiment of the improved complex sequence removal unit of the present invention; Figure 7 is a block diagram showing a first embodiment of the improved correlator of the present invention; Figure 9 is a block diagram showing a second embodiment of the improved correlator of the present invention. Figure 9 is a block diagram showing a further embodiment of the improved correlator of the present invention.

Refer to Figure 1, which is a block diagram of a spread spectrum telecommunications system 100. An information stream is received on line 101. The spreader 102 then encodes the received information stream with a much higher rate tag (or

EN 223 242 B1), thereby generating a scattered or broadcast data sequence of "chips" on line 103. The combination of a higher rate scattering sequence with a lower rate information stream is often called scrambling or coding. The information stream can be scrambled (coded) by performing a negative or operation logically combining the information stream and the scrambling sequence (this is essentially equivalent to arithmetic multiplication if the bits are minus or plus one). Other forms of dispersion are also known. For example, a set of M bits can be scattered by selecting an element from the set of N codewords with bits, where N = 2 ^M. A set of codewords can be an orthogonal set such as a Walsh or Hadamard codeword. Although not illustrated, the broadcast 102 may perform a plurality of scrambling procedures, some of which are common to (or shared by) all broadcast channels before transmitting the broadcast data sequence to line 103.

The scrambled data sequence is then modulated by modulator 104 onto the radio frequency carrier. If the scrambled stream code signals are binary, modulator 104 performs BPSK binary phase shift keying. However, if the scrambled stream code symbols are complex, modulator 104 performs QPSK orthogonal phase shift modulation (phase keying) or offset QPSK. The modulated scattered data sequence is then transmitted to the radiating antenna 106 for transmission by electromagnetic waves.

Receiver antenna 108 receives the signal energy of the broadcast modulated broadcast data and transmits this energy to receiver 110. Receiver 110 amplifies, filters, mixes, and converts the received radio signal (receive signal) to a complex baseband signal, which has both a single phase (I) component and a perpendicular phase (Q) component, as needed. These components are usually sampled at least once per chip period and optionally stored in temporary memory.

The complex baseband signals are transmitted to one or more correlators 112 which correlate the data pulse sequences with the known scattering sequence. This is also called reset because correlation combines multiple scattered data values in a coherent way, restoring their simple information value when the reset sequence is accurately timed to the received pulse sequence. The output correlations are provided to one or more detectors 114 which restore the original information stream. The type of detector used depends on the characteristics and complexity of the radio channel. It may include channel estimation and coherent RAKE combinations, or differential receiver and combinations as needed.

Refer to Figure 2, which is a block diagram of correlator 112 used in the spread spectrum telecommunications system of Figure 1. Via line 121, complex chip pulse sequences are sent to the complex baseband output to the complex sequencing unit 120. The complex sequence deletion unit 120 multiplies these chip pulse sequences by conjugation of an element of the complex scattering sequence [s (k) j, essentially forming the corresponding reset sequence. The resulting chip pulse sequences are transmitted through line 123 to the complex collection and emptying unit 122. The complex collection and evacuation unit 122 collects and then outputs the correlation values (denoted by X) once per code period.

Refer to Figure 3, which shows a more detailed block diagram of the correlator 112 used in the spread spectrum telecommunications system of Figure 1 and shown in Figure 2. On line 121i, the chip phase pulse sequences [denoted by I (t)] of the same baseline component of the complex baseband received by the complex sequence removal unit 120 receive each of the two multipliers 132 and 134. The multiplier 132 is fed by the same phase component of the scattering sequence (denoted by i (k)) and the multiplier 134 is fed by the negation of the perpendicular phase component of the scattering sequence (denoted by -q (k)). Similarly, a

On line 121 q, the chip-divided pulse sequences of the perpendicular phase component of the complex baseband signal received by the complex sequence deletion unit 120 are denoted by the two multipliers 136 and 138, respectively. The multiplier 136 is fed by the same phase component of the scattering sequence [denoted by i (k)] and the multiplier 138 is fed by the perpendicular phase component of the scattering sequence [denoted by q (k)]. Since the scattering sequences i (k) and q (k) are typically plus one or minus one, the multipliers 132, 134, 136, and 138 may be designed as programmable inverters that output either the received pulse sequence values or their negatives. k) and q (k).

The outputs of the multipliers 132 and 138 are summed by the summer 140 and output from the complex sequencing unit 120 on line 123i as the generated single phase chip pulse sequence signals (denoted by I '(t)). The output single phase signals are then output to the complex -collection unit collects. Similarly, the outputs of the multipliers 134 and 136 are summed by the summer 140 and output from the complex sequencing unit 120 on line 123q as perpendicular phase result chip pulse signals [Q '(t)]. The output perpendicular phase signals are then 122 complex collection and emptying units. The complex collection and evacuation unit 122 is emptied and reset once per code period, providing both the imaginary and the real portion of the correlation values (denoted by X). It should be noted that in cases where a

The complex collecting and evacuating units 122 run at twice the speed of the multipliers 132, 134, 136 and 138, and the aggregators 140 and 142 are not required.

Corresponding correlator 112 performs two operations accordingly. The first operation performs the function of the complex sequencing unit 120. The second operation performs the function of the complex collection and evacuation unit 122. The present invention reduces the complexity of both the first and second operations.

Refer to Figure 4, which is a block diagram showing a first embodiment of the improved complex sequencing unit 120 'of the present invention. The complex

The scrambling sequence (or coding, scrambling) sequence can be written as follows:

s (k) = i (k) + jq (k) (1) where i (k) is the same phase component of the scattering sequence and q (k) is the perpendicular phase component of the scattering sequence. Both i (k) and q (k) plus one or minus one. 5A. As illustrated in FIG. 6A, the complex scattering sequence s (k) may assume four possible set values. Rotating these values by forty-five degrees and dividing the square root by two gives the rotated complex scattering sequence values, which can take the values of plus one, minus one, plus j, minus j. as shown in Figure. The improved complex sequence deletion unit 120 'of the correlator A112 preferably uses this relationship to simplify the complex sequence deletion process, taking advantage of the fact that the rotated sequence has zero or zero real sequence. It should be noted that forty-five degrees of rotation is not the only possible rotation and therefore the present invention encompasses all other preferred rotations such as one hundred thirty, two hundred twenty-five and three hundred and fifteen degrees.

The complex sequence deletion method essentially involves multiplying the received complex base band signal by the reset sequence, wherein the reset sequence comprises a conjugate of the complex scattering sequence. In the improved complex sequence removal unit 120 ', the negation unit 150 performs a portion of multiplication of the received complex baseband with the rotated reset sequence by selectively negating or negating the received complex baseband value. The same phase negation unit 150i receives a chip pulse sequence [labeled I (t)] of the same phase component of the complex baseband connected to line 121i. The 150q perpendicular phase negation unit, coupled to line 121q, receives a chip division pulse sequence [denoted by Q (t)] of the perpendicular phase component of the complex baseband.

The improved complex sequence deletion unit 120 'further includes a switch 152 for multiplying the received complex baseband signal and the rotated complex spread sequence, which selectively replaces or omits corresponding phase and perpendicular phase pulses of the received complex baseband values. The switch 152 includes a first switching element 158i associated with the same phase line 156i, which can be controlled to selectively switch, creating a first position link between the same phase line 156i and the line 123i which is the same phase of the resulting chip pulse code signal. (t)]. At the same time, the first switching element 158i provides a second positioning circuit between the same phase line 156i and the line 123q, which outputs the perpendicular phase (denoted by Q '(t)) of the resulting chip pulse code. The switch 152 includes a second switching element 158q connected to the perpendicular phase line 156q and adjustable for selective switching. The second switching element 158q establishes a first position switch between the perpendicular phase line 156q and the line 123q, which outputs the perpendicular phase [denoted by Q '(t)] of the resulting chip pulse code. Alternatively, it establishes a second position link between the perpendicular phase line 156q and the line 123i, which outputs the same phase code signal [denoted by I '(t)] of the resulting chip pulse sequence.

The selective operation of the negating units 150i and 150q and the first and second switching elements 158i and 158q is set by the control unit 155. The control unit 155 receives the complex spreading sequence s (k) and outputs the first and second control sequence signals on line 154 to the negation units 150i and 150q and to the first and second switching elements 158i and 158q, which are essentially s'. (k) multiplying the values of the rotated complex scattering sequence by forty-five degrees of conjugation with s' (k). Note that it is not necessary to calculate the spin complex scattering sequence s '(k) or the conjugate of s' (k). Instead, the logic formed in the control unit 155 processes the complex scattering sequence s (k) to control the same-phase negation unit 150i through a first-phase first-sequence sequence of the same phase to negate the same-phase component of the complex baseband if s (k) -1 + j or 1. is in the form of + j [i.e., the sequence conjugated to s' (k) is -1 or -j]. Otherwise, negation is not performed by the same-phase negation unit 150i [i.e., s' (k) sequence 1 desire], The logic constructed in the control unit 155 similarly controls the orthogonal phase negation unit 150q through a perpendicular phase first-sequence component of the complex baseband signal. to negate if s (k) is in the form -1 + j or -1-j [i.e., the rotated sequence s' (k) is -1 yearning]. Otherwise, the negation is not performed by the perpendicular phase negation unit 150q [i.e., the rotated sequence s' (k) is 1 or -j]. As for the switch 152, the logic contained in both the first and second switching elements 158i and 158q positions the second control sequence signal when s (k) is in the form 1-j or -1 + j [i.e., the rotated sequence s' (k) is 1 or -1]. Accordingly, the same phase line 156i is in the 123i line and the 156q perpendicular phase line to line 123q. Otherwise, both the first and second coupling members 158i and 158q are moved to the second position [i.e., the rotated sequence s' (k) is j or -j]. Accordingly, the same phase line 156i is connected to the perpendicular phase line 123q, and the perpendicular phase line 156q to the line 123i. In practice, negatives can be implemented such that, for example, both negated and non-negated values are always generated, and then a two-position selector switch can be used to select a value from the output. Such a device can be economically manufactured on a small scale using CMOS silicon integrated circuit technology.

Refer to Figure 6, which is a block diagram illustrating a second embodiment of the improved 120 "complex sequencing unit of the present invention. A ferry

The lex number of B1 (e.g. A + jB) can be expressed either in Cartesian (x, y) or polar (R, Θ) form. Conversions between these shapes can be easily performed using the equations x = Rcos (9) and y = Rsin (0). The log-polar form, where r = log (R), is a variant of the aforementioned forms. As discussed above and shown in Figure 5, there are four possible values for the complex scattering sequence s (k). Since each of these four values has a unit amplitude in the complex scattering sequence, it is not necessary to calculate the amplitude change in the scattering sequence for signal processing. However, the phase of the complex scattering sequence changes, causing changes in the phase values of the reception complex baseband signal. Therefore, the phase offset values of +45, + 135 °, + 225 ° (-135 °), and + 315 ° (-45 °) are added to the received complex baseband signal, depending on the values of the complex scattering sequence. Similarly, with respect to FIG. The phase offset values of +0, + 90 °, + 180 ° and + 270 ° (-90 °) are added to the received complex baseband signal. Preferably, the improved 120 "complex sequence deletion unit of the correlator 112 uses fixed values of phase offset values to simplify the complex sequence deletion process.

The complex sequence deletion method essentially consists in multiplying the received complex base band signal by the complex spreading sequence. Recalling the log-polar form discussed above, it is noted that, in the logarithmic range, a multiplication such as that performed in connection with the complex sequence deletion process becomes an addition. Accordingly, in a second embodiment of the improved 120 "complex sequence deletion unit, the received signal is plotted in a log-polar fashion, and the complex sequence deletion method is performed by modulating the phase portion of the log-polar complex base band signal and phase offsets defined by the complex scattering sequence. phase is added or subtracted. The results of these procedures can then be converted back to Cartesian output, if necessary.

Accordingly, the improved 120 "complex sequence deletion unit includes a 170 log-polar signal processing unit disclosed in U.S. Patent No. 5,048,059, issued September 10, 1991 to Paul W. Dent, entitled" Log-Polar Signal Processing ". signal processing. The log-polar signal processing unit 170 receives a signal Rf or lf connected to line 121 and outputs a logarithm of the amplitude of the received signal on line 172. The log-polar signal processing unit 170 further processes the received signal and outputs its extracted phase to line 176. The log polar signal processing unit 170 may include a series of saturation amplifiers to limit the signal and provide waveforms of logarithmic amplitude. The limited signal is used to determine phase values using any known method, such as that described in U.S. Patent No. 5,148,373.

The improved 120 "complex sequence removal unit further comprises a phase converter 178 which receives the complex scattering sequence s (k). The phase converter 178 processes the complex scattering sequence to obtain four possible phase offsets (see Figures 5A and 5B) and to output phase offsets on line 180 therefrom. The logic in the phase converter 178 preferably uses a table to generate the values of the phase offsets of the oh (k). These correspond to the conjugate of phase s' (k). There is also a summing / subtracting or adding 182 in the improved 120 'complex sequence removal unit for combining the extracted phase of the complex baseband signal on line 176 and the phase offset on line 180 to create a phase shift based on the phase of the complex scattering sequence. This removes the phase shift caused by radiation scattering. The addition (or addition / subtraction) input 182 is typically a fixed point integer, allowing the use of an integer modulo arithmetic unit that can be easily constructed and used.

The logarithm of the amplitude of the received signal output from the A170 log-polar signal processing unit (line 172) and the phase shift values from the addition or summing / subtracting 182 are processed by the converter 184. This converts the decoded log polar values to Cartesian values and outputs them to the same phase output on line 1231 [denoted by I '(t)] and to the perpendicular phase output on line 123q [Q' (t) marked] if a Cartesian shape is required for further processing.

Refer to Figure 7, which is a block diagram illustrating a first embodiment of an improved correlator 112 'of the present invention. The correlators 112 'shown in Figures 2 and 3 are essentially arithmetic averages, by means of collection and discharge circuit means, of the complex conjugate of the received signal and the complex scattering or coding sequence. Instead, the improved correlator 112 'computes geometric and multiplicative averages. In this respect, it is noted that the multiplicative mean is an arithmetic mean of logarithmic values. Accordingly, in the improved correlator 112 ', both the phase and the amplitude logarithm of the received complex base band signal are used in the correlation procedure.

Similar to FIG. 6, the improved correlator 112 'of FIG. 7 includes a log-polar signal processing unit 170 which receives, in line with line 121, an Rf or lf signal and outputs a logarithm of the amplitude of the received complex baseband signal 176. outputs a phase of a received complex baseband signal. For the multiplicative averaging discussed above, the logarithmic amplitudes of the received complex baseband signal are output at line 172 from the log-polar signal processing unit 170 to the collecting and discharging unit 190, which performs summation in the logarithmic range. The A190 collection and discharge unit is emptied and erased once per code period, providing numerically averaged logarithmic amplitude values once per code period.

The improved correlator 112 'further comprises a phase converter 178 which receives the complex scattering sequence s (k). The phase converter 178 processes the complex scattering sequence to extract

The two possible phase offset values are shown in Figs. 5A and 5B. From these, the phase converter 178 outputs a phase offset signal on line 180. A phase summer or aggregator 182 summarizes the phase detected on line 176 of the complex baseband signal and the phase offset on line 180, thereby generating a phase shift based on the phase of the complex scattering sequence in the complex baseband signal.

The sum of the phase values is not as obvious as the sum of the logarithmic amplitudes due to the cyclical nature of the phase (i.e. 0 ° and 360 ° are the same). Note that after complex sequence deletion, for phases having a value near 0 ° (or 0 radians), a plus one value signal is transmitted, while for phases having a value near 180 ° (or π radians) a minus one value is transmitted. The multiplicative averaging discussed above is applied to the phase of the received complex baseband by outputting the phase of the received complex baseband shifted by the complex scattering sequence from the phase sum or 182 in the log-polar range, -180 ° and 180 ° (-π and π radians) or other equivalent form. This offset phase is processed by the absolute converter 192 and the collection and discharge unit 194 after the adder 182.

The collection and discharge unit 194 is emptied and cleared once per code period, transmitting numerically averaged absolute values once per code period to the normalization unit 196.

The absolute value converter 192, the acquisition and discharge unit 194, and the normalization unit 196 convert the β-phase according to the following equations:

γ = Σ | Θ (λ) + Φ (*) | (2)

K = 0 β = - (3)

Where 0 (k) is the detected phase of the received complex base band signal emitted on line 176, 4> (k) is the phase offset of the conjugate of the complex scattering sequence s' (k) emitted on line 180 and N is the number of values to which the averaging applies.

The numerically averaged logarithmic amplitude values emitted from the 190 collecting and unloading units and the numerically averaged phase values emitted from the 196 normalization units are processed by the converter 184, which converts the recovered signal from a log-polar to a Cartesian, and the correlation values (X) the imaginary portion being properly output to the 198i and 198q lines. This step essentially converts the log-polar arithmetic mean of the amplitudes and phases to Cartesian multiplicative or geometric mean.

Refer to Figure 9, which is a block diagram illustrating another embodiment of the improved correlator 112 'of the present invention. The detailed description of the reference numerals referring to like elements of Figure 7 is not repeated. The outputs of the phase summer or summator 182 are circularly averaged. Circular averaging means calculating the mean of the phase sinus and the mean cosine of the phase. The outputs of the phase summer or summator 182 are processed in a sine / cosine converter 302 which provides both sine and cosine values. In the acquisition and discharge unit 304, the sine and cosine values are collected separately for a complete period, and then the averaged sine and cosine values are emptied. The averaged sinusoidal and cosine values, as well as the average logarithmic amplitude values from the collecting and unloading unit 190, are transmitted to the formator 306, which converts these quantities into a form suitable for further processing, such as Cartesian or log polar. For example, to convert to a log polar value, an average angular value is constructed from the averaged sine and averaged cosine values using an arc tangent function. The averaged identical phase and the averaged perpendicular phase can also be generated by converting the averaged logarithmic amplitude values to the averaged amplitude values and multiplied by the corresponding averaged sine and averaged cosine values.

In a CDMA (Code Division Multiple Access) spread spectrum telecommunications system, the entire broadcast sequence or tag sequence may occur as a combination of multiple component sequences. Further, multiple channels are known to share one or more common component sequences. For example, in the TIA IS-95 downlink (central transceiver), each channel shares a common complex broadcast sequence called a control sequence. The channels are then spoken individually using different Walsh codewords, which are real or non-complex sequences. However, one channel is not data modulated, so it provides a reference or clean control channel, which can be used to estimate channel load estimates. To demodulate the tagging sequences recorded in a conventional receiver, such as the IS-95 standard, there are separate correlators 112 (see Figures 3 and 4) for each channel. Note that in this example, the individual correlators correlate the complex recovered data with a real Walsh codeword.

Refer to Figure 8, which is a block diagram showing another embodiment of the improved correlator of the present invention. When the broadcast channels share a common broadcast or coding sequence, the receiver 200 has a common recovery or decoding unit 202 assigned to each of the receiving channels to be demodulated. Therefore, the complex chip pulse sequence received on the line 204 for multiple channels is processed by the common recovery or decoding unit 202, removing the common sequence. The outputs of the common reset or decode unit 202 are then processed by a plurality of non-common or unique decoding or reset units 206, which are correlated with different sequences, passing different correlation values at their output to different reception channels (denoted by X _A and X _B ). These individual decoding or resetting units 206 may include a correlator 112 'as shown in FIG. You are the unique decoder 206

The restoration units may also include the 120 'and / or 120' complex sequence removal units shown in Figures 4 and 6. In the previous IS-95 downlink example, the common sequence comprises a complex coding (scrambling) sequence and, accordingly, the common restoring or decoding unit 202 is a matched decoder. In the individual decoding or restoring units 206, Walsh code correlators can be used to work out each received channel and correlate different sequences to provide different correlation values.

For example, an IS-95 based CDMA telephone would have multiple RAKE branches or signal paths, each signal path comprising a common decoding unit and two unique decoding and restoring units, one for the traffic channel and one for the control channel. When two individual decoding and restoring units have to correlate with different Walsh codes, they can share most of the processing with each other. This is possible because half of the bits of one Walsh code are always the same as the corresponding bits of another Walsh code, while the remaining bits are mutually inverted. Therefore, for a common reset unit 202, it is advantageous to calculate the sum (average) of the same bits and to separately calculate the sum for the bits that differ in the two codes. The output is based on the two averages per code signal period. The individual reset units 206 then properly calculate the sum of the two averages and the differences, significantly reducing the calculations.

An example will illustrate how the process is simplified. Consider the following two Walsh code words:

1: 1111111100000000, and

2: 1001011001101001.

Matching bits:

1-1-11-0-0-00, with different bits:

11-1-1-00-0-0, where comparisons were made based on the bit polarities of Walsh codeword 1. In accordance with the foregoing, the first average of the matching eight bits is calculated by reversing the sign of the value to be averaged if the matching bit is zero. The second average of the different eight bits is calculated similarly. If these two averages are added together, the resulting sixteen-bit average represents the correlation with the first of the two Walsh codes, since the bit polarity of the first code is applied to the different bits. Subtracting the second average from the first, the signal of each value contributing to the second average is virtually inverted and thus identified in the second Walsh codeword by the different bits that are inverted relative to the elements of the first codeword to obtain correlation with the second code.

The two correlations with two different Walsh codes can thus be performed in a single operation, which gives two intermediate results per signal period. These results are further combined with reduced rate processing (1/8 rate in this example) to produce the two correlations sought. The expense can be reduced even when many different

Multiple correlations need to be made with the Walsh code. This ultimately forms a fast Walsh transformation structure if all correlations with the Walsh code have to be calculated. For example, U.S. Patent No. 5,356,454 discloses an efficient circuit for calculating fast Walsh transformations.

Although embodiments of the apparatus and method of the present invention are illustrated by the accompanying drawings and illustrated by the above detailed description, the invention is not limited to these embodiments, and many changes, rearrangements and substitutions may be made within the scope of the present invention. The claims are defined and defined.

Claims (24)

PATENT CLAIMS What is claimed is: 1. A correlator for processing a Direct Sequence Spread Spectrum (DSSS) received telecommunications signal having a single phase component [I (t) j and a perpendicular phase component [Q (t)], characterized in that: a control unit (155) for transmitting the first sequence of sequences and the second control sequence of the complex broadcast sequence [s (k) j of the received broadcast signal received in the direct sequence spread spectrum, receiving the same phase component of the received direct signal in the direct sequence spread spectrum [I (t)]; and a first negation unit (150i) negatively controlled by the first control sequence, and a perpendicular phase component [Q (t) j) of a received telecommunication signal received in the direct sequence spread spectrum, and a second negation unit (150q) negatively controlled by the first control sequence; the first and second negating units (150i, 150q) having a switching unit (152) for receiving the same-phase and perpendicular-phase signals and for exchanging the same-phase and perpendicular-phase signals in a manner controlled by the second control sequence. The correlator of claim 1, wherein the first control sequence comprises a single phase first control sequence for controlling negation by the first negation unit (150i) and a perpendicular phase first control sequence for controlling negation by the second negation unit (150q). . The correlator of claim 1, wherein the switching unit (152) has a single-phase output (123i) and a perpendicular-phase output (123q), further comprising: a single-phase switch (158i) receiving the same-phase signal and connecting the same-phase output (123i) or the perpendicular-phase output (123q) in a manner controlled by the second control sequence, and the perpendicular-phase signal and the perpendicular-phase output (123q); a perpendicular phase switch (158q) for switching the same phase output (123i) in a manner controlled by the second control sequence. 4. A method of receiving a direct-sequence spread spectrum received telecommunication signal having a single phase component [I (t) j and a perpendicular phase component [Q (t) j EN 223 242 B1 to remove the complex sequence, comprising the steps of: processing the [S (k) j phase of the complex scattering sequence received for the direct sequence spread spectrum received telecommunication signal, negating the same phase component of the direct sequence spread spectrum received telecommunication signal [I (t) j, orthogonal to the direct sequence spread spectrum received telecommunication signal The [Q (t)] component is negated and the same phase and perpendicular phase output signals are exchanged in a manner controlled by the processed phase. 5. A correlator for processing a direct sequence spread spectrum received telecommunication signal having a logarithmic amplitude component and a phase component, characterized in that: a phase converter (178) for extracting the complex scattering sequence [s (k) j of the direct sequence scattered telecommunication signal [178], the phase component of the direct scattered received spectrum signal and the complex scattering sequence [s (k) and a means for generating a logarithmic amplitude component of a direct-sequence spread spectrum received telecommunication signal and a logarithmic-range complex comprising the summed-phase received signal (<2> (k)}). 6. The correlator of claim 5, further comprising a converting converter (184) for receiving said complex signal and output in Cartesian coordinates. The correlator of claim 5, characterized in that: a first collecting and discharging unit (190) for collecting and emitting logarithmic amplitude components, a second collecting and discharging unit (194) for collecting and normalizing the summed phases of the output absolute, wherein the adder (182) is formed to dispense the absolute value of the summed phase; and the logarithmic complex signal generating device is configured to include collected and released logarithmic amplitude components and a collected and emitted aggregate phase. The correlator of claim 7, wherein the first collection and discharge unit (190) is configured to calculate an arithmetic mean of the collected logarithmic amplitude components. The correlator of claim 7, wherein the second collection and discharge unit (194) is configured to calculate an arithmetic mean of the summed phases of the normalized absolute value collected. The correlator of claim 7, further comprising a converter (184) for receiving the formed complex signal and converting it to output in Cartesian coordinates. 11. The correlator of claim 5, wherein: a first collecting and discharging unit (190) for collecting and emitting logarithmic amplitude components, means for circularly averaging the emitted aggregate phase and emitting the accumulated phase, a means for forming a logarithmic complex signal, and emitting and emitting logarithmic amplitude formed. The correlator of claim 11, wherein the first collection and discharge unit (190) is configured to calculate an arithmetic mean of the collected logarithmic amplitude components. A correlator according to claim 11, characterized in that said circularly averaging means converts (302) a sine / cosine converter for converting an output summed phase to a sine and cosine value, and a numeric mean of the sine and cosine values collecting and collecting the sine and cosine values. a second collecting and discharging unit (304) for calculating and discharging the collected phase. A correlator according to claim 11, characterized by means (184) for converting the complex signal to output in Cartesian coordinates. A method of processing a direct sequence spread spectrum received telecommunication signal having a logarithmic amplitude component and a phase component to remove the complex sequence, comprising the steps of: extracting the phase scatter [s (k)] of the complex scattering sequence of the received sequential spread spectrum telecommunication signal [<t> (k)], obtaining the phase component of the scattering spectrum of the direct sequence scattered received signal and s extracting the complex scattering sequence [s (k)] [<b (k)] is summed and the summed phase is emitted and a logarithmic amplitude component of the received sequential spread spectrum telecommunication signal and a logarithmic complex signal containing the summed phase are formed. 16. The method of claim 15, wherein the formed complex signal is converted to output in Cartesian coordinates. 17. A method of processing a direct sequence spread spectrum received telecommunication signal having a logarithmic amplitude component and a phase component to remove the complex sequence, comprising the steps of: extracting the phase scatter of the complex scattering sequence [s (k) j [<|) (k)] of the direct sequence scattered telecommunication signal, collecting the components having logarithmic amplitude, the phase scattering component of the direct sequence scattered received signal and (k)] summed the phase offset [4> (k)] and outputs an absolute summed phase, the absolute value summed phase emitted is collected and normalized, and a logarithmic amplitude of the logarithmic amplitude of the received direct-spectrum scattered received signal is collected. forming a logarithmic complex signal containing an absolute summed phase. HU 223 242 Bl 18. The method of claim 17, wherein computing the logarithmic amplitude components to be emitted computes an arithmetic mean of the logarithmic amplitude components collected. 19. The method of claim 17, further comprising computing an arithmetic mean of the accumulated normalized absolute phases collected during the collection and normalization of the absolute values to be emitted. 20. The method of claim 17, wherein the complex signal is converted to output in Cartesian coordinates. A method of processing a direct sequence spread spectrum received telecommunication signal having a logarithmic amplitude component and a phase component to remove the complex sequence, comprising the steps of: extracting the phase offset [s (k) j of the direct sequence scattered telecommunication signal [φ (1)], collecting the components of logarithmic amplitude, the phase component of the direct sequence scattered received telecommunication signal and the complex ) The extracted phase offset of J [φ (ΐ)] is summed, and an absolute value of the summed phase is output, the summed phases emitted are averaged and the collected phases are output, and the direct sequence scattered spectrum phase logarithmic complex signal. 22. The method of claim 21, wherein computing the logarithmic amplitude output components is computing an arithmetic mean of the logarithmic amplitude components collected. 23. The method of claim 21, wherein said circular averaging comprises: converting the output summed phases to sine / cosine values and calculating the arithmetic mean of the sine and cosine values collected and outputting the collected phase. 24. The method of claim 21, wherein the complex signal is converted to output in Cartesian coordinates.