GB2325590A - A signal switch for transmit-receive switching of a portable telephone - Google Patents

Abstract

A broad-band low insertion loss switch 200 for use in transmit/receive switching in a portable telephone has a transmit port 202, a receive port 230 and an antenna port 214. In transmit mode a signal from port 202 is split between transmission lines 206 and 208 and recombined at port 214. Phase shifter 220 imparts a zero phase shift so that the split signals are in phase at port 214. Phase shifter 226 imparts a 180 degree phase shift so that signals travelling along lines 222 and 224 are out of phase and cancel at port 230 so that no signal arrives at the receiver. In receive mode phase shifter 220 imparts a 180 degree phase shift so that signals from port 214 split between lines 216 and 218 arrive at port 202 out of phase and cancel, whereas the signals arriving at port 230 are in phase and re-combine to form the received signal. Phase shifter 220 is described in detail and comprises microstrip lines coupled across a slotline. The switch has a multi-layer configuration and can be used over GSM and PCN frequencies. In satellite communications the antenna port 214 may be replaced by two arms of a bifilar or quadrifilar helix antenna.

Description

SIGNAL SWITCH The present invention concerns a signal switch, particularly a broad band, low insertion loss switch for providing coupling from a first port and a second port to either a third port or a fourth port. The invention has particular, but not exclusive application to use in digital portable and cordless telephones. The invention finds further application, for example, in the fields of base stations for mobile radio telephony and radar systems.

In the field of portable telephones, particularly portable digital telephones, small size and long battery life are key requirements. In order to provide an economical telephone handset, some form of switching is required between transmission and reception of signals for the following reason. When the handset is operating to transmit the signals, a signal having a power as high as 2 Watts is transmitted from a transmitter circuit via an antenna. When the handset is operating to receive signals, however, it must be capable of picking up a signal whose strength is as low as -106 dBm. If the power levels produced by the transmitter circuit were to be applied or leaked to the receiver circuit there is a strong possibility that permanent damage to the receiver circuit would occur. Leaked signals from the transmitter to the receiver can also affect the ability of the receiver to pick up low level signals, i.e. its sensitivity. In order to avoid this, telephone handsets are provided with a transmit/receive switch. This circuit switches the antenna of the handset to either the transmitter circuitry or receiver circuitry as appropriate. In this way, the output power of the transmitter circuit is never applied across the sensitive receiver circuitry. Prior art transmit/receive switches typically comprise a pair of diodes or Field Effect Transistors (FETs) possibly in conjunction with a length of transmission line. The high insertion loss of such arrangements causes deterioration of the receiver sensitivity and the transmitter performance with consequent reduction in the talk time of the handset.

It is an object of the present invention to provide a signal switch which ameliorates this disadvantage.

According to the present invention, there is provided a signal switch for providing coupling from a first port and a second port to either a third port or a fourth port, the switch comprising: means for coupling the third port to a first intermediate port ar d means for coupling the third port to a second intermediate port; means for coupling the first intermediate port to the first port and means for coupling the second intermediate port to the second port, wherein one of the means for coupling the first intermediate port to the first port and the means for coupling the second intermediate port to the second port comprises a first phase shifter for providing substantially 0" phase shift or substantially 1800 phase shift; means for coupling the first intermediate port to the fourth port; and means for coupling the second intermediate port to the fourth port via a second phase shifter for providing substantially 1800 phase shift.

The first and second intermediate ports may simply comprise the meeting of three transmission lines on a printed layout.

The first and second ports may be coupled together to provide an antenna port. Alternatively, the first and second ports may be coupled to respective arms of a bifilar or quadrifilar helix antenna.

According to a particular aspect of the present invention, the second phase shifter comprises a second slotline defined by a ground plane; third microstrip means arranged to cross the slotline; fourth microstrip means arranged to cross the slotline; means for connecting the fourth microstrip means to the ground plane on a first side of the second slotline; fifth microstrip means arranged to cross the second slotline; means for connecting the fifth microstrip means to the ground plane on a second side of the second slotline opposite the first side; first switching means for switching the fourth microstrip means to couple a signal between the fourth microstrip means and the second slotline; and second switching means for switching the fifth microstrip means to couple a signal between the fifth microstrip means and the second slotline.

There are three well-known portable digital telephone systems, the Group Speciale Mobile (GSM or GSM 900), The Personal Communications Network (PCN or DCS 1800) and the Digital European Cordless Telephone (DECT). The GSM system operates at around 900 megahertz (MHz) while the PCN operates at around 1800 MHz. The particular aspect of the invention described above may therefore be used to provide a transmit/receive switch which operates over the wavebands of all of these telephone systems. This permits a practicable mobile handset operable on all three bands to be provided.

This refinement of the present invention is based upon an understanding of the behaviour of signals when transferring between a stripline (or microstrip) transmission line and a slotline. Stripline transmission line comprises a conductor and two ground planes, one above and one below the conductor. Ground planes are isolated from the conductor by dielectric layer, possibly air but usually a synthetic dielectric such as RT/Duroid (a registered Trade Mark of the ROGERS Corporation).

Microstrip is similar but comprises only one ground plane. Slotline, on the other hand, comprises a narrow gap in a ground plane. Signals propagating along a stripline or microstrip line (herein referred to collectively as microstrip means) can be coupled into (or out of) a slotline when the microstrip means traverses the slotline. The microstrip means must be grounded on one side of the slotline. The phase of the signal coupled into a microstrip means from a slotline (or vice versa) is dependent upon which side of the slotline the microstrip means is earthed. This is a broad band phenomenon having the additional advantage of low phase dispersion.

The present invention may thus be used in this aspect to provide a signal switch having broad band performance. The fourth and fifth microstrip means may be coupled together on the opposite side from which they may be grounded to the slotline ground plane. Since the fourth and fifth microstrip means may be grounded on opposite sides of the slotline, a signal on these coupled lines can be arranged to provide 0 of phase shift (for example by activating the first switching means) or 1800 of phase shift (by activating the second switching means). In a transmit/receive switch according to this aspect of the invention, the first phase shifter comprises such an arrangement.

This aspect of the invention has the further advantages of excellent higher harmonic rejection in the GSM and PCN wavebands, a higher third order intercept point, increased power handling capability and bandstop response at the second harmonic frequencies of the PCN band.

Further preferred features of the invention are defined in the accompanying claims.

The present invention will now be described, by way of example, with reference to the accompanying drawings, in which: Figure 1 shows a schematic diagram of a switch in accordance with the present invention; Figure 2 is a perspective view of a slotline and microstrip arrangement helpful in describing the operation of elements used in embodiments of the present invention; Figure 3 shows a schematic diagram of a phase shifter for use with one aspect of the present invention; Figures 4(a), (b) and (c) show a printed realization of a switch in accordance with the present invention; Figures 5(a) and (b) show a printed realization of a second phase shifter for use of the present invention in each GSM and PCN wavebands; Figure 6 shows a graph of measured reflection coefficient at the transmitter port against frequency for the switch shown in Figure 4; Figure 7 shows measured antenna port reflection coefficient against frequency for the switch shown in Figure 4; Figure 8 shows a graph of measured transmitter port to antenna port (upper curve) and antenna port to receive port (lower curve) transmission coefficient against frequency for the switch shown in Figure 4; Figure 9 shows the transmit port to receive port isolation against frequency with the antenna port loaded for the switch shown in Figure 4; Figure 10 shows a graph of reflection coefficient against frequency for the phase shifter shown in Figure 5; Figure 11 shows a graph of transmission coefficient against frequency for the phase shifter shown in Figure 5; Figure 12 shows the relative phase against frequency between the two switching states of the phase shifter shown in Figure 5; Figure 13 shows a graph of relative amplitude against frequency between the two states of the phase shifter shown in Figure 5; and Figure 14 shows a schematic diagram of an alternative embodiment of the present invention.

Figure 1 shows a switch 200 comprising a transmitter port 202 (third port), an antenna port 214 (combination of first and second ports) and a receiver port 230 (fourth port). The port 202 is connected to a short section of 50Q transmission line 204 which is split between a 70Q transmission line 206 and a 70Q transmission line 208. Another end of the transmission line 206 is connected to the first end of a further section of 70Q transmission line 222 and a section of transmission line 216.

The point 210 at which the lines 206, 216 and 222 meet is regarded as a first intermediate port. Another end of the transmission line 208 is connected to a section of 70S2 transmission line 226 and a section of transmission line 218. Another end of each of the lines 216 and 218 are connected to the antenna port 214. The transmission line 224 is interrupted by a 1800 phase shifter 226 which will be described in more detail hereinafter. The transmission line 218 is interrupted by a phase shifter 220 which may provide substantially 0 or substantially 1800 phase shift and will be described in greater detail with reference to Figures 3 and 5 hereinafter.

The point at which the lines 208, 212 and 224 meet is regarded as a second intermediate port 212. Another end of each of the lines 222 and 224 are connected together and coupled via a short section of 50Q transmission line 228 to the receive port 230 (fourth port). While meanders are shown in the transmission lines in the schematic circuit diagram, these may not be necessary if the overall size of the switch is sufficient having regard to the frequency of operation and the dielectric constants of the dielectric materials used to realize the switch.

The circuit has two modes of operation, as follows. In a first mode, the phase shifter 220 is regarded as yielding zero phase shift and a transmitter output signal is applied to the transmitter port 202. The transmitter output signal is split between the transmission lines 206, 208 and arrives in phase at the first intermediate port 210 and the second intermediate port 212. The phase shifter 226 inverts the signal at the second intermediate port 212 so that the two signals arriving at the receive port 230 via transmission lines 222 and 224 are out of phase.

These two signals therefore cancel and no appreciable power level is applied to the receiver port (and hence the receiver, not shown). Since the phase shifter 220 is providing 0 of phase shift (or substantially 0 ), the signal from the first intermediate port 210 and the second intermediate port 212 combine in phase at the antenna port 214 via the transmission lines 216 and 218. The transmitter signal applied to the port 202 is therefore applied, substantially unattenuated, to the antenna (not shown) connected to the antenna port 214. Thus the switch achieves the requirements of a transmit/receive circuit in transmit mode, i.e. isolation from the receiver and low loss coupling to the antenna.

In the receive mode, the phase shifter 220 is arranged to provide 1800 of phase shift. In this mode, a signal at the antenna port 214 travels along the transmission lines 216 and 218. These signals are at a much lower level than the transmitter signals. The signals are provided at first intermediate port 210 (in-phase) and at the second intermediate port 212 (out-of-phase). These signals recombine via the transmission lines 206 and 208 but, since they are in out of phase, they cancel. From the first intermediate port 210 and the second intermediate port 212, the received signal travels towards the receiver port 230 via the transmission lines 222 and 224. The signal travelling along transmission line 224 undergoes a further phase change of substantially 1800 in the phase shifter 226. Thus the signals arrive at the receiver port 230 substantially in-phase and thus re-combine to provide an input to the receiver circuitry of the handset. Some losses do occur in the phase shifter 220 and the phase shifter 226. However, by splitting the signal received by the antenna in half to provide two signals, the losses across the phase shifters are approximately halved and overall losses are reduced by about 25%.

Some further optimisation such as introducing different lengths of transmission line at 216, 218 and providing inequal power splitting at the antenna port could be carried out. One realization of the circuitry shown in Figure 1, for use in both the GSM and PCN wavebands is described with reference to Figure 4.

Figure 2 now describes a basic phenomenon upon which elements of the present embodiment are based. Figure 2 shows a perspective view of a first section 10 and a second section 12 of a ground plane which defines a slotline 14. Above the ground plane, and crossing the slotline 14 are a first microstrip line 16 and a second microstrip line 18. The microstrip lines are arranged above the ground plane with a finite separation therefrom. Microstrip line is usually supported with respect to the ground plane by a dielectric material (not shown for clarity). The first microstrip line 16 is shorted to the portion 10 of the ground plane by a conducting stud (or "via") 20. The second microstrip line 18 is connected to the second portion 12 of the ground plane with a via 22.

One aspect of the behaviour of this arrangement is as follows.

If a microwave input signal is applied at point X to the section of the microstrip line 16 distant from the via 20, this signal will be coupled into the slotline 14. The exact mechanism by which this occurs is known, but is complex and beyond the scope of the present description. The behaviour of microstrip line 18 is the reverse of that of microstrip line 16. A signal Y taken from the end of the microstrip line 18 distant from the via 22 provides a signal which has been coupled-out of the slotline 14. Because the microstrip line 18 is grounded on the opposite side of the slotline 14 from the grounding of the microstrip line 16, the signal Y taken from the microstrip line 18 is in anti-phase (or has a 1800 phase shift) with respect to the input signal X. The first phase shifter of the present invention may exploit this phenomenon by providing switching means to derive signals either in phase or out-of-phase from the slotline 14.

Figure 3 shows a schematic diagram of an embodiment of a phase shifter 220 used in accordance with a refinement of the invention. A ground plane 32 is shown which may extend further in any or all directions.

Within the ground plane is provided a slot line 34 which is open (e.g.

36) at each end. The oval gap 36 in the ground plane is a means of illustrating that the slotline is open (or effectively so).

An input port 38 of the circuit is coupled to a microstrip line (or stripline) 40 which crosses the slotline 34 and is earthed to the ground plane on the right-hand side of the slotline at 42. A microstrip line (or stripline) 44 crosses the slotline 34 and coupled in succession to diode 46 and earth 48. A microstrip line (or stripline) 50 crosses the slotline 34 and is coupled in succession to diode 52 and earth at 54. The earth 48 of microstrip line 44 is on the opposite side of the slotline from the earth 42 while the earth 54 of the microstrip line 50 is on the same side of the slotline 34 as the earth 42. The microstrip lines 44, 50 are combined together to provide an output port 56.

Some switching circuitry is required in order to switch the diodes 46, 52 on and off. This circuitry, however, has been omitted from the present diagram for reasons of clarity. It will be discussed in more detail with reference to Figure 5 later. The operation of the phase shifter shown in Figure 3 is as follows.

A signal is imagined to be applied to the input port 38. This signal propagates along the line 40 which is earthed at 42. Since the line 40 crosses the slotline 34 and is earthed on one side thereof, the input signal is coupled into the slotline 34. If we imagine that diode 52 is conducting and diode 46 is off, then a signal is coupled into the line 50.

The line 50 is a microstrip line (or stripline) which crosses the slotline 34 and is earthed 54 on one side thereof. As discussed with reference to Figure 2, a signal will be coupled from the slotline 34 to the microstrip line 50. Since the line 40 and the line 50 are earthed at 42 and 54 on the same side of the slotline 34, the output signal at 56 will be in-phase with the input signal at 38 (00 of phase shift). Since the diode 46 is off, no signal is coupled into the microstrip line 44. The output signal from the port 56 is thus an in-phase signal.

In the opposite circumstances, the diode 52 is off while the diode 46 is on. A signal is still imagined to be applied to the input port 38. The signal is coupled into the slotline 34 as described previously but instead of being coupled out of the slotline by the microstrip line 50, it is coupled out by the microstrip line 44. Since diode 52 is off, no signal is coupled into the microstrip line 50 and the signal on microstrip line 44 is coupled to the output 56. The microstrip line 44 is earthed on the opposite side of the slotline 34 from the earth 42 of the microstrip line 40. As a consequence, the signal coupled into the microstrip line 44 from the slotline 34 is in anti-phase with the input signal applied to the port 38.

Consequently, simply by switching either diode 46 or diode 52 on, a 0 or 1800 phase shift can be provided. This phase shift gives good broadband performance over an octave of frequency (see experimental results discussed later).

By use of this phase shifter in a switch according to the present invention, a broadband transmit/receive switch can be realized.

Figures 4(a), (b) and (c) show a practical, printed realization of the switch shown schematically in Figure 1. Figure 4(a) shows a plan view of a printed stripline layout 300 together with a slotline in a ground plane 306 and its image (with respect to plane 300) in a ground plane 304.

The transmitter port 202 is connected via striplines 206, 208 to vias V1 and V2 respectively. A via is a pin located substantially perpendicular to the plane of the paper for connecting one level of a multi-level arrangement to another one. The vias V1 and V2 are connected to microstrip circuitry which is described with reference to Figure 4(b). Via V1 equates with the first intermediate port in Figure 1 and via V2 equates with the second intermediate port. On the stripline layout, a stripline 222 proceeds from the via Vitro the receive port 230. A stripline 224 proceeds from the via V2 to the RX port 230. However, the stripline 224 is interrupted by a phase shifter as shown in Figure 2.

A first section of the line 224 is terminated on the left-hand side of a slotline 310 using a via V3 to both of the ground planes 304 and 306.

The slotline 310 is open at either end, for example as depicted by the oval 312. A remaining section of the line 224 is earthed to both of the ground planes 304 and 306 by means of via V4 on the right-hand side of the slotline 310. This section is connected to the receive port 230. The stripline 224, slotline 310 and vias V3, V4 operate as a 1800 phase shifter as described with reference to Figure 2. The relative locations of the ground plane 306 and the stripline layout 300 will be described in further detail with reference to Figure 4(c).

Figure 4(b) shows a microstrip layout including the lines 216, 218, the antenna port 214 and the 0"/180" phase switch 220. The microstrip line 216 extends between via V1 (also connected to the stripline layout 300) and the antenna port 214. The microstrip line 218 extends between the via V2 and the antenna port 214 and includes the 0 /180 phase shifter 220 which will be described in greater detail with reference to Figure 5. The dimensions of the layout d3 and d4 are shown in this figure.

Figure 4(c) shows the relative positions of the stripline layout 300 and the microstrip layout 302. The layer 307 corresponds to the microstrip lines of the phase shifter 220. Also shown in the figure are ground planes 304 and 306 which serve as upper and lower ground planes to the stripline layout 300 and includes the slotlines used in the 0 /180 phase shifter 220 (see Figure 5). Further, layer 304 serves as a ground plane to the microstrip layout 302. The overall dimensions d3 and d4 are 15 mm for the particular dielectric used. The stripline shown in Figure 4(a) has a length between the transmitter port 202 and the via V1 of 13.36 mm. The microstrip line 216 has a length of 7.85 mm and the microstrip line 218 has a length of 10.15 mm. Where possible, the same reference numerals as those used in Figure 1 have been used for corresponding features of the arrangement shown in Figure 4. The various layouts and ground planes are separated by RT/Duroid 6010 dielectric which has a dielectric constant of 10.2 and a specified loss tangent (tan d) of 0.0023 at 10GHz.

While one of the diodes in this embodiment is conducting while in receive mode, by using a low current PIN diode the effect on the standby time of a portable telephone handset using such a transmit/receive switch is small.

Figure 5 shows an actual implementation of the phase shifter 220 described with reference to Figure 3. Figure 5(a) shows a plan view showing the components of the arrangement, while Figure 5(b) shows a side view to indicate which components are arranged on which level.

The arrangement 60 includes an RF input port 62 connected to a first plate a capacitor 64. The capacitor has a typical value of 33 (33pF) picofarads to 68 picofarads (68 pF). This capacitor is to isolate an input signal applied to port 62 from the DC signals discussed later. A second plate of the capacitor 64 is connected to a via V4 which is a conducting pin extending substantially in the plane of the paper for connecting the capacitor 64 to a microstrip line 66 which is on a different level within the device. A first portion 68 of a conductive plane and a second section 70 of the conductive plane comprise a slotline ground plane.

The slotline 72 is defined between these two portions of ground plane.

The stripline 66 crosses the slotline 72 and is connected to the slotline ground plane 70 by means of a via V1. The microstrip line 66 thus couples an input signal into the slotline 72. A further microstrip line 74 crosses the slot line 72 and is attached at one end to an anode of a diode 76 and at another end to a resistor 78 and a first plate of a capacitor 80. The cathode of the diode 76 is connected to the slotline ground plane 68 by means of a via V3. The other end of the resistor 78 is connected to a first DC control port 82 while the other plate of the capacitor 80 is connected to a further section of microstrip line 83.

Another end of the microstrip line 82 is connected to a microstrip line 84 by means of a via V5.

A further section of microstrip 86 crosses the slot line 72 and is connected at one end to the anode of a diode 88 and at another end to a resistor 90 and a first plate of a capacitor 92. The cathode of diode 88 is connected to slotline ground plane 70 by means of a via V2. Another end of the resistor 90 is coupled to a second DC control port 94 while another plate of the capacitor 92 is connected to a section of microstrip line 96. Another end of microstrip line 96 is connected to the microstrip line 84 by means of via V6. The microstrip line 84 further comprises an output RF port 98 which is coupled to the vias V5 and V6.

As stated above, not all of these components are at the same level.

Figure 5(b) shows the five layers of the device. The layer 100 comprises the microstrip ground plane which extends completely under the circuitry of the device. The microstrip line ground plane 100 is separated from the microstrip line circuitry by a dielectric layer 102. In the present example the dielectric layer comprises RT/Duroid 6010 which has a dielectric constant of 10.2 and a specified loss tangent, tand, equal to 0.0023 at 10ghz. These layers are each 1.9 mm thick The layer 104 contains the microstrip line 84 and the slotline ground plane where portions 68 and 70 are arranged. The microstrip lines 66, 74 and 86 together with diodes 76 and 88, microstrip line sections 83 and 92, and capacitors 80 and 92 are all arranged on layer 108. This layer is connected by means of vias V1, V2 and V3 to the slotline ground plane (portions 68, 70) on layer 104. The vias V1, V2 and V3 may further extend to connect the portions 68, 70 of the slotline ground plane to the microstrip ground plane (common ground) 100. For use in this arrangement shown in Figure 4 the ground plane 100 corresponds with layer 304, the layer 104 corresponds with the layer 302 while the layers 108 and 307 are equivalent.

In operation an RF input signal is applied to port 62 and coupled out at port 98. If a voltage of 10 Vdc is applied to DC control port 82 then a current will flow through the resistor 78, the microstrip line 74, the diode 76 and the via V3 to ground. The resistor 78 has a resistance of 1 kOhm and is provided to limit the current through the diode 76. When a DC control voltage is applied to the port 82 the diode 76 will conduct, effectively shorting one end of the microstrip line 74 to the slotline ground plane 68. A signal from the slotline 72 will thus be coupled into the microstrip line 74 in accordance with the principles discussed above.

This signal will pass through the capacitor 80 and the microstrip line 82, via V5 and microstrip line 84 to the RF output port 98. The capacitor typically has a capacitance of 33 to 68pF and is provided to isolate the DC control signal applied at port 82 from the RF output port. Because the microstrip line 74 is earthed to the slotline ground plane on the opposite side of the slot line 72 from the grounding of microstrip line 66 (when diode 76 is conducting), the signal available at RF output port 98 will be in anti-phase to that at RF input port 62 (1800 phase shift).

When a DC control voltage of 10 Vdc is applied to DC control port 94 a current flows via resistor 90, microstrip line 86, diode 88 and via V2 to slotline ground plane 70. The resistor 90 is provided to limit the current through the diode 88 and has a value of 1 kOhm. Thus, when a signal is applied to RF input port 62, a signal is coupled from the slotline 72 into the microstrip line 86. This signal flows via capacitor 92 through the microstrip line 96, via V6 and microstrip line 84 to RF output port 98. The capacitor 92 is provided to isolate the RF output port 98 from the DC control signal applied at port 94 and may have the same capacitance as capacitor 80. The operation is analogous to that described above for a DC control voltage applied to port 82. However, since the microstrip line 86 is connected to the slotline ground plane 70 on the same side of the slotline 72 as the microstrip line 66, the signal at RF output port 98 will be in-phase with a signal applied to RF input port 62.

Therefore, whether the phase switch imposes no phase change (00 phase shift) or 1800 phase shift can be determined by application of control voltages to either port 94 or port 82.

The dimensions of the device shown in Figure 5 are as follows: dl = 6 mm, d2 = 8 mm, Figure 6 shows a graph of measured transmit port reflection coefficient (dB) against frequency when the receive port is open. The horizontal axis is marked in MHz while the vertical axis is marked in dB. The GSM transmit waveband is identified by arrows 1 and 2 while the PCN transmit waveband is identified by arrows 3 and 4. The graph indicates the good performance attained in practice. The actual reflection coefficients at the point marked on the graph are: 1 880MHz -15.548dB 2 915MHz -17.270dB 3 1710MHz -26.873dB 4 1785MHz -30.937dB Figure 7 shows the antenna port reflection coefficient against frequency with the transmit port open for

The GSM receive bands are indicated by the arrows 1, 2 while the PCN receive bands are indicated by the arrows 3, 4. The measured reflection coefficients of these four points are as follows: 1 935MHz -11.112 dB 2 960 MHz -11.803 dB 3 1805 MHz -17.758 dB 4 1880 MHz -17.665 dB Figure 8 shows the measured transmit port to antenna port and the antenna port to receiver port transmission coefficient (dB) for the device shown in Figure 4. The horizontal axis is marked in MHz while the vertical axis is marked in dB. The GSM transmit wavebands are identified by the arrows 1, 2 while the PCN transmit wavebands are identified by the arrows 3, 4. The upper curve represents the measured transmit port to antenna port attenuation. It can be seen that this is less than 0.5 dB over the relevant GSM and PCN bands. The attenuation at the marked pointed is as follows: 1 880 MHz -.446 dB 2 915 MHz -.3978 dB 3 1710 MHz -.4453dB 4 1785 MHz -.4787 dB.

The lower curve represents the antenna port to receiver port insertion loss against frequency. This again shows a very low level of attenuation.

Figure 9 shows the transmit port to receive port isolation with the antenna port loaded for the device shown in Figure 4. The horizontal axis is marked in MHz while the vertical axis is marked in dB. The GSM receive bands and the PCN receive bands are indicated by the arrows 1, 2 and 3, 4 respectively. The curve shows that the isolation figures for this arrangement are sufficient. The measured attenuations at these four points are as follows: 1 935 MHz -17.753 dB 2 960 MHz -18.025 dB 3 1805 MHz -38.692 dB 4 1880 MHz -39.948 dB Figures 10 to 13 show test results obtained from testing of the phase shifter shown in Figure 5. Figure 10 shows a graph of the reflection coefficient in dB against frequency measured at the input port, when a 50 Ohm termination is connected to the output port. Since the device is reciprocal, the same results will apply if the "input" and output ports are interchanged. The horizontal axis represents frequency in MHz while the vertical axis is marked in decibels (dB). The graph shows the GSM frequency band between arrows 1 and 2 while to the right of the graph, the arrows 3 and 4 show the PCN frequency band. The measured attenuations at these four points are as follows: 1. 880 MHz -10.177 dB 2. 960 MHz -11.418 dB 3. 1710 MHz -22.285 dB 4. 1880 MHz -32.377 dB Figure 11 shows a graph of transmission coefficient in dB against frequency for the 1800 phase change mode. The insertion loss is shown in dB on the vertical axis while the frequency is shown in MHz on the horizontal axis. The arrows 1, 2 and 3, 4 show the GSM band width and the PCN band width respectively. The actual attenuations at these four points are as follows: 1. 880 MHz 0.6515 dB 2. 960 MHz 0.6456 dB 3. 1710 MHz 0.6091 dB 4. 1880 MHz 0.7155 dB This graph shows a consistently low level of attenuation across all of the frequency bands of interest.

Figure 12 shows the relative phase between the 0 and 1800 phase switching modes against frequency. The vertical axis represents degrees while the horizontal axis is calibrated in MHz. The curve actually shows the difference in phase between the output in the 0 phase state subtracted from the output in the 1800 (anti-phase) state. It can be seen that the relative phase is very close to 1800 The variation of the phase difference does not exceed i5 over an octave of bandwidth. The small offset from 1800 can be compensated with a small section of microstrip lines. The four arrows 1, 2 and 3, 4 delineate the GSM and PCN frequency bands as before. The actual relative phase changes at these four points are as follows: 1. 880 MHz 173.340 2. 960 MHz 173.310 3. 1710 MHz 171.390 4. 1880 MHz 175.860 This level of phase change is well within acceptable limits for use in a transmit/receive switch as discussed previously.

Figure 13 shows a graph of the relative amplitude of the output of the phase shifter in the 0 state and the 1800 state. As before, the vertical axis is marked in decibels and the horizontal axis is marked in MHz.

The four arrows 1,2 and 3,4 delineate the GSM and PCN wave bands as discussed previously. The actual variation in amplitude between the two states at these four points is: 1. 880 MHz 0.2418 dB 2. 960 MHz O.2878 dB 3. 1710 MHz -0.0939 dB 4. 1880 MHz 0.3682 dB These variations are very small and well within acceptable limits.

Figure 14 shows an alternative embodiment of the present invention with particular application to an Intermediary Circular Orbit (lCO) User Terminal for Mobile Satellite Communications. Figure 14 differs from Figure 1 in that instead of microstrip line 216 and microstrip line 218 being connected to provide antenna port 214, they are connected to respective windings of a bifilar(or quadrifilar) helix antenna (not shown).

In the case of a bifilar antenna, the microstrip line 216 and the microstrip line 218 are connected to the two arms of the antenna. The two arms of the antenna are fed in anti-phase to create circular polarisation meaning that the arrangement shown in Figure 14 acts both as a switch and a balanced feeding arrangement for a mobile antenna. The operation of the arrangement will be the reverse of that shown in Figure 1 so that in transmit mode the phase shifter 220 will provide 1800 and in receive mode it will provide 0" of phase shift.

A quadrifilar helix antenna could be used instead of a bifilar helix antenna. In this case the microstrip line 216 and the microstrip line 218 will each be connected to provide a pair of lines having relative phases of 90". The four arms of the quadrifilar antenna are then connected to these two pairs of lines which provide signals at relative phases of 0 , 90 , 1800 and 270".

The invention is not restricted to the embodiments described since this can be modified to utilise different components, different sizes (for example to accommodate different frequency ranges) and different materials as is known to those skilled in the art. Possible variations include different switching devices (e.g. FET's or bipolar transistors) and the use of an arrangement based on more microstrip line and less stripline. The polarity of the switching diodes and the polarity of the control signal applied thereto could both be reversed. The in-phase and anti-phase sections of stripline may be reversed with respect to input stripline (66, Figure 3(a)). Alternative phase shifters could be used.

The present invention also encompasses any novel feature disclosed herein, implicitly or explicitly, as will be apparent to the skilled person.

Claims (15)

1. A signal switch for providing coupling from a first port and a second port to either a third port or a fourth port, the switch comprising: means for coupling the third port to a first intermediate port and means for coupling the third port to a second intermediate port; means for coupling the first intermediate port to the first port and means for coupling the second intermediate port to the second port, wherein one of the means for coupling the first intermediate port to the first port and the means for coupling the second intermediate port to the second port comprises a first phase shifter for providing substantially 0 phase shift or substantially 1800 phase shift; means for coupling the first intermediate port to the fourth port; and means for coupling the second intermediate port to the fourth port via a second phase shifter for providing substantially 1800 phase shift.

2. A switch as claimed in Claim 1, wherein the first port and the second port are coupled together to comprise an antenna port, the third port comprises a transmitter port and the fourth port comprises a receiver port.

3. A switch as claimed in Claim 1, further comprising a bifilar or quadrifilar helix antenna, wherein the first and second ports are connected to separate arms of the antenna.

4. A switch as claimed in Claim 1, Claim 2 or Claim 3, wherein the second phase shifter comprises a coupling from a first microstrip means to a second microstrip means via a first slotline, which slotline is defined by a ground plane and arranged to cross the first and second microstrip means.

5. A switch as claimed in any one of the Claims 1 to 4, wherein the means for coupling the third port to the first intermediate port is substantially 1/4 wavelength long at an upper operating frequency.

6. A switch as claimed in any one of Claims 1 to 5, wherein the means for coupling the third port to the first intermediate port is substantially 1/8 of a wavelength long at a lower operating frequency.

7. A switch as claimed in any one of Claims 1 to 6, wherein the first phase shifter comprises: a second slotline defined by a ground plane; third microstrip means arranged to cross the second slotline; fourth microstrip means arranged to cross the second slotline; means for connecting the fourth microstrip means to the ground plane on a first side of the second slotline; fifth microstrip means arranged to cross the second slotline; means for connecting the fifth microstrip means to the ground plane on a second side of the second slotline opposite the first side; first switching means for switching the fourth microstrip means to couple a signal between the fourth microstrip means and the second slotline; and second switching means for switching the fifth microstrip means to couple a signal between the fifth microstrip means and the second slotline.

8. A switch as claimed in Claim 7, wherein the first and second switching means comprise diodes.

9. A switch as claimed in Claim 8, further comprising a resistor connected to each diode and wherein each diode is controlled by a DC voltage applied via a respective resistor.

10. A switch as claimed in any one of the Claims 7 to 9, wherein the microstrip means comprise striplines.

11. A switch as claimed in any one of the Claims 7 to 9, wherein the microstrip means comprise microstrip lines.

12. A switch as claimed in any one of the Claims 1 to 11, wherein the means for coupling the third port to the first intermediate port and the means for coupling the third port to the second intermediate port, the means for coupling the first intermediate port to the fourth port and the means for coupling the second intermediate port to the fourth port comprise 70Q transmission line.

13. A switch as claimed in Claim 12, wherein the first port, second port, third port and fourth port have substantially 50Q impedance.

14. A switch as claimed in any one of the Claims 1 to 13, wherein the switch comprises a printed arrangement.

15. A signal switch substantially as herein described with reference to the accompanying drawings.