GB2204751A - Discharge lamp circuits - Google Patents

Abstract

A.C. mains, coupled through a simple filter F to a full wave rectifier bridge B supplies power to an inverter circuit having d.c. input terminals 12, 13, coupled by a bypass capacitor C0. A primary high frequency circuit includes series resonant capacitor C and inductor L components, the capacitor C being in parallel with a primary inductor L1 which, through the resonant inductor L, is in series with a diode D1 that blocks the flow of direct current from the positive rectifier output terminal 12 to the negative terminal 13. To establish the resonant frequency current in the primary circuit, a switch S1 is connected anti-parallel to the diode D1. A drive circuit 14 drives the switch S1 to execute switching cycles at a rate equal to the series resonant frequency of L and C. The high frequency current is propagated in a secondary circuit including a discharge lamp 11 in series with a current limiting capacitor C1, an inductor L1' which is part of an autotransformer formed by L1 and L1', and a further inductor L3 which is inductively coupled to the primary inductor L1 or inductor L so as to cancel harmonics of the resonant frequency of L and C1 the inductor L3 being of a size which with the capacitor C1 is series resonant at an unwanted harmonic of the said resonant frequency. The switch S1 may be a gate turn off thyristor, a gate turn off thyristor in series with a transistor (30), (Figure 10), or an FET (Figure 8). The autotransformer may provide voltage step-up or current step-up (Figure 5) for use with low and high pressure lumps respectively. Alternatively, an isolating transformer may be used (Figure 6), in which case the primary circuit resonance inductor may be provided by leakage reactance of the transformer and the associated resonance capacitor by the capacitive reactance, reflected into the primary circuit via the transformer, of a capacitor & connected in parallel with the secondary L1' (Figure 7). <IMAGE>

Description

DISCHARGE LAMP CIRCUITS This invention relates to discharge lamp circuits and especially to circuits for supplying high frequency current to a discharge lamp.

It is known that operation of discharge lamps by means of high frecuencies results in improvement in the luminous efficiency in most cases and in the elimination of the flicker that occurs at low frequencies due to cyclic reignition. By high frequency in this context is meant a frequency of substantially 1 kilohertz or higher.

In the particular case of fluorescent tubes, if the tube is supplied with current at a frequency of greater than about 1 kilohertz, the ionization state of the gas is not able to follow the changes in the lamp current, with the result that a nearly constant plasma densitv and a nearly constant electrical impedance are maintained in the tube, so that the dynamic electrical characteristics of the tube tend to become linear and the waveforms of tube current and voltage more closely sinusoidal.

Operation of low pressure discharge lamps at high frequency results in lower electrode losses and, usually, a gain in positive column radiation, in comparison with lower frequency operation.

High frequency operation of discharge lamps can improve lamp run up time and result in smoother operation, so that longer lamp lifetime may be achieved. Furthermore, a better power factor can be achieved without the use of a large correction capacitor.

Choice of operating frequency must be made also to avoid acoustic resonance in the arc tube of certain lamps.

High frequency discharge lamp circuits are known which are powered by a.c. mains supply. The known circuits of this type have a high frequency inverter supplied from a bridge rectifier with a large reservoir capacitor to smooth the rectified alternating current. The capacitor is an electrolytic capacitor. Rectifiers with smoothing capacitors produce a peaky, distorted waveform.

Heavy filtering is required to make this waveform suitable for driving the high frequency inverter. Consequently conventional ballast chokes are employed in each supply line to the inverter to provide undisturbed line current and an acceptable power factor.

Such ballast chokes are large and result in the supply circuit being ahout the same size and weight as other conventional discharge lamp supply circuits. A result of the known form of high frequency circuit for use with an a.c. mains supply is that such circuits are only available commercially for lower power fluorescent tubes or fluorescent lamps specially made for operation with such circuits and having a high running voltage and therefore a low lamp current.

Disadvantages of the known high frequency discharge lamp circuits powered bv a.c. mains supply are limitation to efficiency improvement on account of switching losses and dissipation in resistor-caDacitor snubber circuitry, and mains pollution with radio frequency interference due to the generation of nearly rectangular current waveforms by the inverter circuit. Such known circuits may also involve complex transformer design and control circuitry for continuous operation under changing lamp and fault conditions.

According to the present invention there is provided a discharge lamp circuit comprising a rectifier for full wave rectifying mains alternating current and an inverter for converting the output of the rectifier into a high frequency current to pass through a discharge lamp, characterised in that the rectifier is connected directly to direct current input terminals of the inverter, bypass capacitor means couples the said input terminals together at the high frequency generated by the inverter, and the inverter comprises a primary circuit which includes the bypass capacitor means, primary inductor means, resonating capacitance effectively in parallel with the primary inductor means, resonating inductance effectively in series with the resonating capacitance, undirectional conducting means connected in series with the bypass capacitor means and the primary inductor means and arranged to prevent current flowing therethrough from the positive direct current input terminal of the inverter to the negative direct current input terminal of the inverter by way of the primary inductor means, and switching means connected to provide a current path bypassing the unidirectional conducting means and adapted to operate so as to execute switching cycles at a rate substantially equal to the series resonance frequency of the resonating capacitance and resonating inductance, and a secondary circuit including output terminals to be connected to the electrodes of a discharge lamp, capacitive impedance means arranged to limit lamp current, first inductive means inductively coupling the secondary circuit to the primary inductor, and second inductance means arranged to couple lamp current inductively to the primary inductor means or to the resonating inductance, the capacitance of the capacitive impedance means and the inductance of the second inductance means being series resonant at substantially the second or a higher harmonic of the series resonance freauency of the said resonating capacitance and resonating inductance, and the arrangement being such that the coupling of the second inductance means to the primary inductor means or to the resonating inductance opposes Dropagation of the said harmonic in the secondary circuit.

Preferably the high frequency is above substantially 20 kilohertz.

Several low cost solid state circuits embodying the invention will now he described with reference to the accompanying drawings.

The circuits described are for operating a range of low and high pressure discharge lamps up to powers of a few kilowatts. These circuits convert a.c. mains supply into high frequency for lamp control. They feature a nearly pure sinusoidal waveform with a minimal mains input current distortion and minimal radio frequency interference. Some of them employ a single semiconductor switch which minimises the cost of the semiconductors and associated circuitry. The circuits described are dual resonant arrangements and incorporate a circuit which carries out the function of both a switching regulator and an inverter. The circuit arrangement generates a high frequency sinusoidal current at its output, so there is far less main pollution and radio frequency interference than in, for example, a conventional switch mode power regulator.

Because only a decoupling capacitor is used after a bridge recifier, and not a large reservoir capacitor as in known circuits, the mains line current flowing to the input is also sinusoidal and undistorted with the result that the circuit is capable of converting sufficient mains power to run high voltage discharge lams without exceeding the accepted limits of mains harmonic generation. A secondary circuit gives a negative feed back effect reducing high frequency harmonics.

In the accompanying drawings: FIG. 1 is a circuit diagram of part of a first embodiment of the invention: FIG. 2 is a circuit diagram of a second embodiment; FIG. 3 is a detailed circuit diagram of a drive circuit which is part of the embodiment of Fig. 2; FIG. 4 is a more detailed circuit diagram of a drive circuit suitable for use in the embodiment of Fig. 2; FIG. 5 is a circuit diagram of a third embodiment; FIG. 6 is a circuit diagram of part of a fourth embodiment; FIG. 7 is a circuit diagram of a fifth embodiment, FIG. 8 is a circuit diagram of a sixth embodiment, FIG. 9 is a circuit diagram of a drive circuit which is part of the embodiment of Fig. 8, and FIG. 10 is a circuit diagram of a modification of the drive circuit of Fig. 3.

Fig. 1 shows part of a lamp operating circuit embodying the invention. In the circuit of Fig. 1, a full wave rectified voltage Vo, produced by a full wave rectifier (not shown) driven by an a.c.

mains supnly giving 240 volts at 50 hertz, is applied across a capacitor CO having a capacitance which is less than 5 microfarads.

The circuitry to the right, in Fig. 1, of the capacitor CO is connected to a discharge lamp 11 which is supplied, in operation, by the lamp operating circuit, with a high frequency current, i.e. an a.c. current having a frequency of substantially 1 kilohertz or higher. The value of the capacitance of the capacitor CO is such that this capacitor acts as a bypass capacitor for the high frequency, so that no high frequency current passes into the rectifier or into the mains supply. The circuitry includes a series resonant combination of an inductor L and a capacitor C with a switch S1 connected across the series resonant combination and a diode D1 connected in parallel with the switch.Current is supplied to the capacitor C through an inductor L1 connected between one input terminal 1.2 and the unction point between the series resonant inductor L and capacitor C, the other electrode of the capacitor C being connected directly to the other input terminal 13 . The diode Dl is poled to hlock the passage of direct current from the rectifier through the inductors. The switch S1 is operated by a driving circuit (not shown) at a rate equal to the series resonance frequency of the combination LC.The mark/space ratio of the switching of the switch S1 is 1:1 and the switch S1 is open during a part of each cycle of series resonance of the combination LC in which a high voltage, for example substantially 1000 volts appears across the switch Sl. The resonance of the combination LC generates the high frequency current which energises the lamp 11. Thus the frequency of the current supplied to the lamp 11 is the series resonance frequency of the combination LC.

The inductor L1 with the capacitor CO and the combination LC with the switch S1 and diode D1 form a high frequency primary circuit in the supply circuitry for the lamp 11. The inductor L1 together with an extension inductor L1', a capacitor C1, the lamp 11, a feedback inductor L3 and the capacitor CO form the high frequency secondary circuit in the supply circuitry. The inductor L1 with its extension L11 act as an autotransformer at the high freauencv to provide a suitably high voltage for starting and operating the lamp 11. A suitable voltage for a low pressure sodium discharge lamD is in the range 600 to 700 volts. The value of the capacitor C1 is chosen to limit the lamp current to a suitable value for the type of lamp.

The feedback inductor L3 is magnetically coupled to the resonance inductor L by being wound on the same core. The purpose of the feedback inductor L3 is to feed back the second or higher harmonics of the fundamental high frequency lamp current in such a phase relation as to effect cancellation of those harmonics. Close magnetic coupling of the feedback inductor L3 to the resonance inductor L is required. The harmonics thus fed back are at frequencies determined by the series combination of the lamp current limiting capacitor C1 and the feedback inductor L3.

The magnitude of the high frequency voltages generated can be varied by varying the conduction time of the switch S1 and by varying the operating frequency of the switch S1- The magnitude decreases as the conduction time of S1 is reduced, and also decreases as the operating frequency of the switch S1 is varied from the series resonance frequency of the combination LC. These two effects can, if desired, by used to adjust the magnitude of the high frequency voltage.The effect of varying the operating frequency is a result of the frequency of the high frequency currents generated being determined by the operating rate of the switch S1. If the operating rate of the switch S1 departs from the series resonance frequency of the combination LC, the high frequency currents generated are at the operating rate of the switch S1 and are presented with a corresponding higher non-resonant impedance by the combination LC.

The circuit of Fig. 1 generates self-maintaining high frequency oscillations, provided that the inductance of the primary inductor L1 of the autotransformer is greater than the inductance of the resonance inductor , and the capacitance of the decoupling (bypass) capacitor CO is greater than the capacitance of the resonance capacitor C.

When the switch S1 is operated, current in the resonance inductor L oscillates sinusoidally about zero at the resonant frequency of the series resonant combination LC, so that the resonant frequency is = 2 n LC The oscillatory current I generated is equal to the voltage across CO divided by the impedance of the combination LC, i.e.

I = VO/Z sin t For stable self oscillation, the minimum value of the voltage across the switch S1 must reach zero during each cycle.

The voltage across the resonant capacitor C can be equal to between 2Vo and 3Vo, depending upon the conduction period during each cycle of lamp current.

The voltage VL across the lamp 11 depends mainly on the ratio of L1' to L1. The autotransformer L1, L1' provides a voltage step up for lamp ignition or stepdown for a lamp having a lower arc voltage.

The series resonant combination of the inductor L3 and the current limiting capacitor C1 have values of L3 and C1 chosen to resonate at approximately the second harmonic of the fundamental frequency of the series resonant combination LC.

The switch S1 need conduct only in the direction which allows direct current to pass through the inductor L from the input terminal 12. Fig. 2 shows a lamp operating circuit embodying the invention and connected to drive a sodium discharge lamp 11, and having a gate turn off thyristor as the switch S1.

The circuit of Fig. 2 has a.c. main input terminals Q and n to be connected respectively to the live and neutral output terminals of a 240 volts a.c. mains supply. A simple filter F formed by an inductor and a capacitor may be included between the a.c. input terminals Q and n and a bridge rectifier B that provides a full wave rectified output voltage with the negative bridge output terminal grounded. The circuitry between the terminals 12 and 13 and the lamp 11 is the same as that in Fig. 1 except for the position of the high frequency resonance capacitor C which in the circuit of Fig. 2 is connected in parallel with primary L1, of the autotransformer L1, L11.

A drive circuit 14 is connected to the gate terminal of the gate turn off thyristor S1, and to a point a at the end of the primary L1 connected to the terminal 12. All that is required of the drive circuit 14 is that it cause the switch S1 to execute onoff cycles at the rate which is the series resonance frequency of the combination LC. The point a to which the drive circuit is connected is the positive direct current output terminal of the full wave rectifier B. The circuit 14 is also connected to the grounded output terminal 13 by a connection not shown in Fig. 2. Fig. 3 shows details of the drive circuit 14. The output terminal 15 of the circuit 14 is connected to the gate terminal of the thyristor S1' The drive circuit 14 includes an inductor L1(2) which consists of a few turns of wire coupled to the inductor L1* When the lamp 11 is operating, high frequency power is coupled into the drive circuit 14 by the inductor L1(2)' and a suitable low voltage produced at positive supply line 16 by the action of a diode D2, limiting resistors R2 and R3, a bypass capacitor CD, and a zener diode Z arranged as shown. The supply line 16 serves an oscillator 17 and a transistor output circuit 18. The oscillator 17 generates a square wave at the series resonance frequency of the combination LC, and the output circuit 18 drives the gate turn off thyristor S1 in response to the oscillator output.

To start the drive circuit 14, and hence to start the lamp 11, the circuit 14 is connected to the point a from which, at the initial application of mains power to the circuit of Fig. 2, direct current passes through a limiting resistor R1 and a thermistor Rt, having a positive temperature coefficient, to the capacitor CD.

Since the thermistor is initially cold, its resistance is initially low, and the capacitor CD charges rapidly to a level at which the oscillator 17 starts and the drive output circuit 18 triggers the gate turn off thvristor S1 so that the generation of the high frequency in the circuit of Fig. 2 begins. The oscillator 17 and drive output circuit 18 remain powered by direct current from the point a until the thermistor Rt has warmed sufficiently for its resistance to reduce the flow of current from the point a to a negligible value. During this time, the high frequency power coupled in by the inductor L1(2) increases to a level sufficient to maintain energisation of the oscillator 17 and the drive output circuit 18, so that the lamp operating circuit with the drive circuit 14 is then self maintaining, i.e. maintains its operation.

The residual small current drawn through the resistor R1 and the thermistor Rt is sufficient to keep the thermistor Rt in a high resistance state.

The drive circuit 14 is designed to ensure that the initial current to the gate turn off thyristor S1 is large enough for the turn on losses to be low, and that a low impedance is presented to the gate of the thyristor S1 to effect turn off, so that gate turn off losses are low.

Fig. 4 shows a detailed example of the drive circuit 14. The oscillator 17 in this example makes use of a Signetics 555C timer 17'. The transistor output circuit 18 has three NPN transistors Tr1, Tr2 and Tr3, and one PNP transistor Tr4 connected as shown.The power consumption of the drive circuit of Fig. 4 when driving a 6 to 8 amp gate turn off thyristor is approximately 0.75 watts.

Circuit component values for the circuit of Fig. 2 when used to operate different discharge lamps will now be given.

1. Operation of a 135 watt SOX lamp CO = 3.3 microfarads C = 0.007 microfarads C1 = 0.01 microfarads L = 0.65 millihenrys: wound on ferrite E core measuring 25 millimetres by 25 millimetres by 7 millimetres, or wound on a ferrite bobbin measuring 20 millimetres by 20 millimetres.

L1 = is millihenrys L1' = 1.5 millihenrys L1 and L1, are wound to form an autotransformer on a ferrrite E core type N 32-120-25 The bridge B has four diodes of the type 1N5626 The switch S1 is a gate turn off thyristor of the type BT 157 150C)R The diode D1 is the type BY229.

2. Operation of a 90 watt SOX lamp.

CO = 3.3. microfarads C = 0.05 microfarads C1 = 0.01 microfarads L = 0.65 millihenrys: wound on ferrite E core or a ferrite bobbin as in 1 above.

t1 = 18 millihenrys L1( = 1.2 millihenrys L1 and L1' are wound to form an autotransformer on two ferrite U cores of type N 34-015-25.

The bridge B, switch S1 and diode D1 are as in 1 above.

3. Operation of a 125 watt MBF lamp.

CO = 4.7 microfarads C = 0.01 microfarads C1 = 0.08 microfarads L as for 1 and 2 above L1 = 18 millihenrys L1' = 4.5 millihenrys L1 and L1' are wound to form an autotransformer on a ferrite E core.

The bridge has four diodes of the type 1N5626 The switch S1 is a gate turn off thyristor of the type BTW581300.

The diode D1 is the type BY229.

In the three examples given above, the following approximate efficiencies have been found: 135 watt SOX lamp run with 1 amp current.

S1 2.0 watts D1 0.5 watts Drive circuit 14 0.75 watts B 0.75 watts L, L1, L1 and L3 3.50 watts other losses 0.50 watts Total loss 8.0 watts Total wattage 143 watts Efficiency 94%.

90 watt SOX lamp run with lamp current.

Total wattage 98 watts Efficiency 92%.

125 watt MBF lamp run with lamp current Total wattage 134 watts Efficiency 93.

The lower switching losses are attributed to a low rate of change of current.

The thermistor Rt in the drive circuits of Fig. 3 and 4 can be replaced by a small thyristor arranged to be turned off when CD becomes charged to a suitable voltage.

An advantage can be obtained with the drive circuit of Fig. 4 bv suitable choice of the value of resistance presented by the resistor R1. If the lamp 11 does not ignite, because it fails or is faulty, the current flowing in the inductor L1 is low and, with a sufficiently high value of resistance for the resistor R1, the resulting voltage applied to the drive circuit by the inductor L1(2) is not sufficient to maintain operation of the drive circuit. The initial current drawn through the resistor R1 and the thermistor Rt causes only momentary operation of the drive circuit, and is rapidly reduced by the resulting rise in temperature of the thermistor Rt.

Thus the drive circuit, and hence the whole lamp circuit, remains in a passive state in which negligible current is drawn from the mains supply and no voltage appears across the large terminals since the mains frequency is blocked by the capacitor C1.

Fig. 5 shows an alternative lamp operating circuit which embodies the invention and, like Fig. 2, uses an autotransformer.

Parts of the circuit of Fig. 5 which have already appeared in Fig. 2 are given the same reference numerals or letters as in Fig. 2.

The circuit of Fig. 5 differs from that of Fig. 2 in that the inductor L11 is connected between the capacitor C1 and the point a, i.e., the positive d.c. output terminal of the bridge rectifier B.

Consequently, whereas in Fig. 2 the lamp high frequency current passes through the autotransformer inductors L1 and L1' in series, so that their voltages are additive, in Fig. 5 the primary winding L1 of the autotransformer is bypassed and the lamp high frequency current circulates around a circuit consisting of, in series, the lamp 11, capacitor C1, inductor L1', bypass capacitor Co, and inductor L3. This arrangement results in a lower starting voltage applied to the lamp 11 but allows a higher lamp current to circulate for a given switch S1. The circuit of Fig. 2 does not allow such a high lamp current as the circuit of Fig. 5 because the switch S1 in Fig. 2 conducts part of the high frequency lamp current, the inductor L being smaller than the inductor L1 with which it is in parallel at the high frequency. The circuit of Fig. 5 thus provides current step-up as opposed to voltage step-up, and is useful for operating a high pressure mercury or high pressure sodium lamp, a lower tube running voltage and a higher arc current being required for such lamps compared with low pressure sodium lamps and cold cathode fluorescent lamps.

A modification of the circuit of Fig. 5 is shown in Fig. 6 in which complete isolation of the lamp circuit from the resonating/switching circuit is achieved by arranging the inductors L1 and L11 as primary and secondary windings respectively of a transformer T without d.c. connection between its primary and secondary circuits. The transformer T must be substantially without leakage. The inductor L3 is provided as a further secondary winding of the transformer T. High pressure mercury lamps can be run by this modification. However, high pressure sodium or mercury halide lamps require a high voltage starting pulse of 2 kilovolts to 4 kilovolts if the lamps are standard single ended lamps and an additional modification to provide such an ignition pulse is then required in the form of an ignition circuit.

Fig. 7 shows an alternative circuit embodying the invention in which the high frequency resonant components are provided by the leakage reactance of a ferrite core transformer T1 having the inductor L1 as primary winding and the inductors L1' and L3 as secondary windings, and the reflected capacitive reactance of the capacitor C which in this circuit is connected in parallel with the inductor L1'. The inductance of the inductor L1 is, in this circuit, preferably at least twice the leakage inductance of the transformer T1. The series resonance frequency of the bypass capacitor CO with the inductor L1 must be sufficiently low compared with the switching rate of the switch S1 for there to be no excitation of this low frequency in the circuit.In particular, the series resonance frequency of the bypass capacitor CO with the inductor L1 must not be a subharmonic of the high frequency which is to be generated. The reflected capacitance which appears across the primary inductor L1 is the value of the capacitor C multiplied by the square of the turns ratio of L1 to L1', which is in accordance with the conventional transformer theory. The values of the capacitor C1 and the inductor L3, which are intended to ensure that harmonics of the fundamental high frequency lamp current are cancelled, must be chosen with regard to the presence of the leakage of the transformer T1.

The respective windings of the inductors L1 and L1' are physically separated on the core of the transformer T1 and their separation is adiusted to provide the required value of leakage inductance in the primary circuit for resonance at the desired high frequency.

An advantage of the circuit of Fig. 7 is that all the windings can be wound on a single ferrite core.

As in the case of the circuit of Fig. 5, the circuit of Fig. 7 can be modified to provide complete d.c. isolation of the high frequency lamp circuit from the resonating/switching circuit by omitting the connection between the high frequency lamp circuit and the grounded terminal 13.

The leaky transformer T1 can be constructed with two separate core portions so that tuning to the desired resonance frequency can be carried out by relative movement of the core portions towards or away from each other. For example, the inductors L1 and L1' may be formed by respective windings on respective E-cores arranged to face each other at the ends of their core limbs. The inductor L3 is, in such an arrangement, formed by a winding on the core of the inductor L1.

Fig. 8 shows the circuit of an embodiment which has the same basic configuration as the circuit of Fig. 1 and uses a transistor as the switch Sl. The transistor S1 is a high voltage power field effect transistor of a MOS FET type. The inductive surge across the transistor during switching is reduced by the provision of a series combination of a zener diode Z2 and a resistor R5 which is connected in narallel with the transistor S1. A drive circuit 20, which is a modification of the drive circuit 14 of Fig. 4 and is shown in detail in Fig. 9, drives a second MOSFET 21. The second transistor 21 is connected to act as a switch across the gate and grounded terminal of the switch transistor S1.The drive circuit 20 receives power from only the point a, and does not include a thermistor or any equivalent component for cutting off the supply of current from the point a. The oscillator 17 of the drive circuit 20 is coupled by a resistor capacitor coupling to the gate terminal 22 of the second transistor 21. The gate terminal 23 of the switch transistor S1 is coupled by a resistor R4 to the positive supply line of the oscillator 17 and hence is resistively coupled to the point a through three resistors: R1, R6 and R4. The voltage on the supply line of the oscillator 17 is stabilised by a capacitor 24 and a zener diode 25, the capacitor 24 coupling the junction point between the resistors R1 and R6 to ground, and the zener diode being connected between the supply line and the ground line of the oscillator 17.The timer 17' of the oscillator 17 and the second transistor 21 draw very low current from the point a, and the high frequency current variation in the resistor R1 due to the switching of the second transistor 21 is prevented by the bypass capacitor CO from reaching the bridge rectifier B. The low power requirements of the drive circuit 20 and the second transistor 21 offset the additional losses occurred by the zener diode Z2 and the resistor R9-. The circuits of Figs. 8 and 9 drive a 55 watt SOX discharge lams 11 at 120 kilohertz.

Figure 10 shows a drive circuit which is a further modification of the drive circuit of Fig.3. In the circuit of Fig.10, a bipolar switching transistor 30 is connected in series with the main switching element S1 which is a gate turn off thyristor. The base of the transistor 30 is driven by the oscillator 17, which can be the same oscillator circuit as in Fig.4 where it drives a transistor Tr1. Power for the oscillator 17 is supplied continually from the point through the resistor R1. When the transistor 30 is turned off by the oscillator 17, current flowing into the thyristor S1 passes through the turn-off gate of the thyristor and charges up a small capacitor 31 of, for example, 82 pico farads, and also, through a diode 32, passes into the larger capacitor CD, for example, 100 microfarads. The voltage at the cathode of the thyristor S1 is prevented from rising above a predetermined level by a clamping diode 33 connecting the thyristor cathode to the cathode of the zener diode Z.

When the thyristor S1 is not conducting and the transistor 30 is turned on by the oscillator 17, the capacitor 31, which has been charged to a predetermined voltage by the diverted thyristor current at turn off and current from the point cr through the resistor R1 and a resistor 34 in parallel with the diode 32, discharges through the turn-on gate of the thyristor S1, thereby rendering the thvristor S1 conducting again. The voltage across the capacitor 31 does not rise sufficiently during the conducting phase of the transistor 30 to prevent the turn-off gate of the thyristor S1 from going negative when the transistor 30 turns off and the thyristor current is diverted to the capacitors 31 and CD.

Claims (10)

1. A discharge lamp circuit comprising a rectifier (B) for full wave rectifying mains alternating current and an inverter for converting the output of the rectifier (B) into a high frequency current to pass through a discharge lamp, characterised in that the rectifier (B) is connected directly to direct current input terminals (12,13) of the inverter, bypass capacitor means (cho) couples the said input terminals (12,13) together at the high frequency generated by the inverter, and the inverter comprises a primary circuit which includes the bypass capacitor means (cho), primary inductor means (L1), resonating capacitance (C) effectively in parallel with the primary inductor means (L1), resonating inductance (L) effectively in series with the resonating capacitance (C), unidirectional conducting means connected in series with the bypass capacitor means (cho) and the primary inductor means (L1) and arranged to prevent current flowing therethrough from the positive direct current input terminal (12) of the inverter to the negative direct current input terminal (13) of the inverter by way of the primary inductor means (L1), and switching means (S1) connected to provide a current path bypassing the unidirectional conducting means (D1) and adapted to operate so as to execute switching cycles at a rate substantially equal to the series resonance frequency of the resonating capacitance (C) and resonating inductance (L), and a secondary circuit including output terminals to be connected to the electrodes of a discharge lamp (11), capacitive impedance means (C1) arranged to limit lamp current, first inductance means (L1') inductively coupling the secondary circuit to the primary inductor (L1), and second inductance means (L3) arranged to couple lamp current inductively to the primary inductor means (L1) or to the resonating inductance (L), the capacitance of the capacitive impedance means (C1) and the inductance of the second inductance means (L3) being series resonant at substantially the second or a higher harmonic of the series resonance frequency of the said resonating capacitance (C) and resonating inductance (L), and the arrangement being such that the coupling of the second inductance means (L3) to the primary inductor means (L1) or to the resonating inductance (L) opposes propagation of the said harmonic in the secondary circuit.

2. A discharge lamp according to claim 1, wherein the resonating inductance (L) is provided by an inductor connected in series with the primary inductor means (L1) and the unidirectional conducting means (D1).

3. A discharge lamp circuit according to claim 1 or 2, wherein the resonating capacitance (C) is provided by a capacitor connected in parallel with the primary inductor means (L1).

4. A discharge lamp circuit according to any preceding claim, wherein the primary inductor means (L1) and the first inductance means (L1') are respectively portions of an autotransformer.

5. A discharge lamp circuit according to claim 1, wherein the first inductance means (L1') is a secondary winding of a transformer (T) having the primary inductor means (L1) as a primary winding, and the secondary circuit is isolated from the primary circuit.

6. A discharge lamp circuit according to claim 5, wherein the resonating capacitance is provided by reflection of capacitive impedance (L) connected in parallel with the first inductance means (L1?).

7. A discharge lamp circuit according to claim 5 or 6, wherein the resonating inductance is provided by leakage reactance of the transformer (T1).

8. A discharge lamp according to any preceding claim, wherein the switching means is a gate turn off thyristor (S1) and a drive circuit (14) is provided for controlling the switching action of the thyristor (S1), the drive circuit (14) including means (17,18) for generating a switching signal supplied to the gate of the thyristor (S1), first means (R1,Rt) for supplying power to the generating means (17,18) for a predetermined duration starting at energisation of the rectifier (B), and second means (L1(2)Dz) for supplying power to the generating means (17,18) during operation of the thyristor (S1).

9. A discharge lamp circuit according to claim 8, wherein the first means for supplying power to the generating means (17,18) comprises a conductive path for direct current from the rectifier (B), and the conductive path includes a thermistor (Rt) having a positive temperature coefficient such that the thermistor (Rt) substantially prevents flow of direct current from the rectifier (B) to the generating means (17,18) during operation of the switching means (S1) except in the said predetermined duration.

10. A discharge lamp according to claim 1, wherein the switching means comprises a field effect transistor (S1) adapted to be switched by an oscillator (17) supplied with direct current from the rectifier (B).