GB1605271A

GB1605271A - Radio signalling equipment

Info

Publication number: GB1605271A
Application number: GB1380175A
Authority: GB
Inventors: S Calvard; A P Morgan; E J Davis; J D Maines; G L Moule
Original assignee: SECR DEFENCE; UK Secretary of State for Defence
Current assignee: SECR DEFENCE; UK Secretary of State for Defence
Priority date: 1975-04-03
Filing date: 1975-04-03
Publication date: 1987-06-10

Description

(54) IMPROVEMENTS IN OR RELATING TO RADIO SIGNALLING EQUIPMENT (71) I, SECRETARYOFSTATEFOR DEFENCE. LONDON, do hereby declare the invention, for which I pray that a patent may be granted to me, and the method by which it is to be performed, to be particularly described in and by the following statement: The present invention relates to receivers for a radar or telecommunications system particular to apparatus for enhancing the probability of detection of a signal at a receiver.

In known radar apparatus, it is common to use a bi-phase modulated carrier wave as a transmitted signal and to provide an identically coded sequence of signals which can be delayed and correlated with signals reflected from a target and received at a receiver to give the required target range information. The modulation of the carrier wave is achieved either by phase shift keying techniques in which the keying operations are controlled by the output of a digital code generator, or injection phase locking the carrier wave oscillator to a stored replica of the previous transmitted pulse in an interrupted carrier wave coherent radar system such as described in the co-pending Application No. 25344/72 Specification No. 1,432,541.

In known telecommunications apparatus, it is common to use a binary signal representation of data for transmission and to code each 1-signal and each 0-signal of the representation with a separate code sequence. In this way each 1-signal and each 0-signal of a given duration is sub-divided in a predetermined way into a number of much shorter duration 1-signals and 0-signals. The effect of this is to widen or spread the frequency spectrum of the transmitted signals so that reception and decoding by an unwanted recipient is difficult.

In the telecommunications apparatus it is also common to use a bi-phase modulated carrier wave as the transmitted signal where the bi-phase modulations are controlled by the coded data signal.

It is an object of this invention to provide a receiver for a radar or telecommunications system in which the bi-phase coded signal is received and detected in a manner which is relatively insensitive to the frequency of the return signal (ie doppler frequency shifts have relatively little effect on the detection probability) and in which the signal-to-noise ratio of the received signal is enhanced significantly above that achieved in alternative doppler independent receiver circuits for example that described in Application No.

56322/72, Specification No. 1,445,572.

According to the present invention, a receiver for a radar or telecommunications system for receiving signals phase-modulated by a predetermined regular digit-signal rate, includes: a decoding means for applying a sequence of phase-shifts, complementary to those given by the code sequence, to the received signals; a first signal path and a second signal path having a common input connected to an output of the decoding means; a delay device connected in the first signal path, having a multitude of output tappings disposed so that the delays between adjacent tappings are integral multiples of the bitduration of the code signals, and means for combining signals from the output tappings;; and signal mixer means connected to receive at one input the combined signals from the delay device and at a second input signals from the second signal path, wherein the signal mixer means is arranged to derive an autocorrelation signal the magnitude of which is dependent on the relative phases of the received signals.

The delay device may be an acoustic delay line having a plurality of transducers equispaced along its length, the distance between any two adjacent transducers being an integral multiple of the distance propagated by acoustic waves in the delay line in one bitduration of the code-signals, these transducers forming the output tappings of the delay line and being connected to a common output.

According to one aspect of the invention the receiver is associated with transmitting apparatus comprising a pulse generator circuit for generating pusles at the desired pulse repetition frequency (PRF) of the radio signalling equipment, a surface wave acoustic delay line, two input transducers, one at each end of the surface wave acoustic delay line an integral number of acoustic wavelengths at an intermediate frequency (IF) apart and each connected to an output of the PRF generator, and an output transducer positioned on the delay line between the two input transducers and separated from each by an integral number of acoustic wavelengths at the said IF.

The PRF generator may be an IF surface wave acoustic oscillator, such as is described in co-pending Application No. 7880/73 Specification No. 1,451,326, or in the journal Ultrasonics - May 1974 issue at page 115 in the paper by M.F. Lewis "Surface Acoustic Wave Devices and Applications", driving a step recovery diode impulse generator. The output of the delay line may be connected to a random or pseudo-random bi-phase modulating circuit which is synchronised by clock pulses derived from the output of the step recovery diode impulse generator.

An embodiment of the invention as applied to an interrupted carrier wave phase-coherent proximity-fuze radar will now be described, by way of example only and with reference to the drawing accompanying the provisional specification which shows the transmitter and receiver circuits of a proximity-fuze radar in schematic circuit diagram form.

In the drawing the IF stages of a proximityfuze radar carried for example in the warhead of a Surface-to-Air missile (SAM), comprises a surface acoustic wave oscillator 1 providing signals at 50 MHz which are applied to a divide-by-ten circuit 2 to produce a 5 MHz reference signal. A step recovery diode impulse generator circuit 3 is connected to the output of the divide-by-ten circuit 2, so as to produce sharply rising pulses corresponding to each positive peak of the 5 MHz signal hence determining the pulse repetition frequency (PRF) of the radar system. The output of the step recovery diode impulse generator circuit 3 is connected to two input transducers 4 and 5 at the extremities of a lithium niobate surface acoustic wave delay line 6 and also to the clock input of a pseudo-random code generator circuit 7.The input transducers 4 and 5 on the lithium niobate surface acoustic wave delay line 6 are separated by an integral number of acoustic wavelengths at a frequency of 100 MHz. Each pulse from the step recovery diode impulse generator circuit 3 causes the transducer 4 and the transducer 5 to resonate at 100 MHz for 60 ns, ie 30% of the interval between successive pulses from the step recovery diode impulse generator circuit 3. An output transducer 8 is provided at an intermediate position on the delay line 6, at an integral number of acoustic wavelengths from each of the input transducers 4 and 5. A biphase modulator circuit 9 has one input connected to the output of the transducer 8 and another input connected to receive the output signals of the pseudo-random code generator circuit 7.The pseudo-random code generator circuit 7 produces a 1023 bit pseudorandom sequence of binary signals each bit of which is produced at its output at a time corresponding to the receipt of a clock pulse from the step recovery diode impulse generator circuit 3. Each 60 ns pulse of 100 MHz signal from the output of the transducer 8 will have its phase shifted by the bi-phase modulator circuit 9 according to the polarity of the binary signal received coincidentally from the pseudo-random code generator circuit 7. For example, the bi-phase modulator circuit 9 may give a 0 phase shift to one of the 60 ns pulses of 100 MHz signals when a 0-signal is received from the pseudo-random code generator circuit 7, and a n phase shift to another 60 ns pulse of 100 MIIz signals on receipt of a 1signal.

The output of the bi-phase modulator circuit 9 is connected to one input of an up-mixer circuit 10 another input of which is driven from a local oscillator circuit 11 which may be a surface acoustic wave device. The up-mixer circuit 10 maintains the pahse of the IF bi-phase modulated pulses but increases their frequency to a suitable microwave frequency for transmission. The output of the up-mixer circuit 10 is amplified in a radio-frequency power amplifier 12 and fed via a transmitreceive (TR) switch 13 to an aerial for transmission. The transmitted signal thus consists of a coded sequence of bi-phase modulated microwave pulses which will be reflected by a target 15 and received by the aerial 14 after a propagation delay dependent on the range of that target 15.The TR switch 13 is operated by pulses derived from the step recovery diode impulse generator circuit 3 so that for the first 60 ns following a pulse from the generator circuit 3 the transmitter is connected to the aerial 14 and for the remaining 140 ns between successive pulses from the step recovery diode impulse generator 3 the aerial is connected to the receiver. This is described as 30% duty cycle operation.

The receiver circuit is shown in the drawing; it includes a pre-amplifier circuit 16 which has an input connected via the TR switch 13 to the aerial 14 and has an output connected to a down-mixer circuit 17 which has another input connected to the output of the local oscillator 11. The down-mixer circuit 17 converts the received microwave frequency signals to the 100 MHz IF maintaining the phase relationship between successive pulses. The IF signal is amplified in conventional IF amplifier stages 18 and fed to one input of a demodulator circuit 19. Another input of the demodulator circuit 19 is connected via a range delay circuit 20 to the output of the pseudo-random code generator circuit 7.The demodulator circuit 19 operates in a similar manner to the bi-phase modulator circuit 9 ie the phase of each 60 ns pulse of 100 MHz signals received from the IF stages 18 is determined in the demodulator circuit 19 by the polarity of the delayed binary signal received coincidentally from the pseudo-random code generator circuit 7.

However, the demodulator circuit 19 applies phase shifts to the receive signals which are complementary to those applied by the bi-phase modulator circuit 9 in the transmitter.

Thus, if the example given above of the operation of the bi-phase modulator circuit 9 was the case, the demodulator circuit 19 would give a phase shift of one received signal when a 0-signal is received from the pseudo-random code generator circuit 7 and a 0-phase shift to another received signal on receipt of a signal.

When the target is at a range corresponding to the time delay given to the binary code sequence by the delay cicuit 20 all the phase shifts applied by the modulator 9 will be complemented by the phase shifts applied by the demodulator 19 so that the output from the demodulator circuit 19 will have no phase discontinuities. This technique is also described in the copending UK patent Application No.

56322/72, Specification No. 1,445,572.

The output of the demodulator circuit 19 is split into two channels. One channel is fed directly to a first input of a phase sensitive detector 21 while the other channel is fed to an input transducer on a quartz delay line 22 which has eighty output tappings, all connected to a common output. This common output is connected to a second input of the phase sensitive detector 21. The output of the phase sensitive detector 21 is connected via a low-pass filter 23, an integrator circuit 24 and a threshold circuit 25 to an output of the system which in this case is connected to a detonation circuit (not shown).

The eighty-taps delay line 22 and the phase sensitive detector 21 form an auto-correlator, and together with the circuits 23 to 25 form an arrangement similar to that described in copending Application No. 58827/71, Specification No. 1,404,590, in which each received pulse is first auto-correlated with a one-bit delayed version of the preceding pulse in order to overcome carrier frequency variations due, for example, to the Doppler effect. In this case however the one-bit delay is replaced in the auto-correlator by the eightytap delay line 22 which in total provides an eighty bit delay so that the phase sensitive detector 21 compares the phase of each pulse with the phase of a signal formed from a summation of the eighty preceding pulses. This is a pre-detection integration technique which provides significant signal-to-noise enhancement in the receiver.

It will be appreciated that in interrupted carrier wave systems such as that described in the copending Application No. 25344/72, Specification No. 1,432,541, the problem was to provide a transmitter which could be completely switched off during reception periods, so that the signal-to-noise ratio of the received signal was not impaired, and yet when switched on for the next transmission period would have its phase coherently preserved in relation to the previous transmission. In the apparatus described above however a carrierwave oscillator is not used and the requisite quiet reception periods and phase coherence is achieved by the impulse generator circuit 3 and the delay line 6.

The three transducer configuration of the lithium niobate surface acoustic wave delay line 6 is a commonly used technique for reducing the insertion loss which is associated with two-transducer delay-lines. Typically the insertion loss associated with conventional delay lines is halved by employing this threetransducer technique. Phase coherence between successive pulses of the transmitted signal is ensured because of the exact reproducibility of the response of the transducers 4 and 5 to the sharp pulses produced by the step recovery diode impulse generator 3. The transmitter therefore produces the 30% duty cycle microwave pulse train coded with bi-phase modulation according to the 1023-bit pseudo-random sequence generated by the pseudo-random code generator circuit 7.Between each transmited burst and the next a period 140 ns is allowed in which pulses may be received by the receiver after reflection from the target 15.

When the target is at the specified range the demodulator circuit 19 reduces all the phase coded signals received to a coherent phase relationship and after the auto-correlation process described for enhancing the signal-tonoise ratio of the signal the output of the lowpass filter 23 and the integrator 24 will gradually build up to a level which will eventually exceed the reference level applied to the threshold circuit 25. The output of the threshold circuit 25 is then used to detonate the SAM warhead.

It will now be appreciated that the arrangement described in the example embodiment provides not only the postdetection integration afforded by systems such as described in copending Patent No. 56322/72, Specification No. 1,445,572, but also gives predetection integration by the use of the multi-tap delay line 22. The combination of post detection and predetection integration gives more processing gain than a system employing only post detection integration and is less sensitive to Doppler frequency shifts than a system employing only predetection integration.In the embodiment an eighty-tap delay line was used; the number of taps used in any particular application is arbitrary but should be carefully chosen, because in general an N-bit IF pulse integrator (which in effect the tap delay line is) is quite frequency sensitive and the range of frequencies that it will successfully operate upon is only 1 wide to the 3 dB point, where N is the Nm number of taps or pulses to be integrated and T is the bit period. Thus in applications where Doppler frequency shifts are expected, the number of taps in the predetection integrator should be chosen to be as high as possible (to increase the predetection integration) while keeping the expected Doppler frequency shifts within the useful passband of the delay line integrator.

It will be noted that the system described in the example makes extensive use of surface acoustic wave devices. The oscillator 1, the local oscillator 11 and the delay lines 6 & 22 are all surface acoustic wave devices. This allows the radar fuze system to be made considerably smaller and lighter in weight than a comparable system made with alternative conventional devices, due to the 100,000 times smaller velocity of acoustic surface waves as compared with electromagnetic waves of the same frequency. Because the surface acoustic wave oscillator 1 which defines the pulse repetition frequency is based on the same substrate as the eighty-tap delay line 22 the surface acoustic wave functions are essentially insensitive to temperature changes.This is a very significant consideration and means that the substrate can be, for example, lithium niobate with its high coupling constant, ie low coupling losses but its high temperature coefficient which would otherwise be undesirable. Aluminium nitride on sapphire is another possibility enabling radio frequency surface acoustic wave components for the system to be fabricated without the use of an electron beam. These devices are planar structures making them very easy to integrate into miniturised electronic packages such as the system described here.

Known interrupted carrier wave coherent radar systems have sensitivities limited by inadequate suppression of the carrier during the reception periods but the system described hereinabove has a transmitter which is completely quiescent during reception periods and can provide carrier suppression of greater than 120 dB.

The method of demodulation of the IF pulses in the receiver gives scope for optimum predetection integration using the surface acoustic wave tap delay line 22 as described above. In general the greater the percentage of time spent in predetection integration compared with post-detection integration the more sensitive is the receiver. The predetection integration time ie the number of taps on the delay line may be limted by the expected Doppler shift as described above, or in practice by the maximum size of the surface wave substrate which can be obtained.

Although a specific example has been described with respect to a radar fuze the system of combined predetection and postdetection integration described above has application to communications. Slow data for transmission may be spread over a wide bandwidth which is a well-known anti-jam technique in communications. The improved signal-to-noise ratio provided by the receiver in the above system is of course extremely desirable in communications systems where unknown Doppler shifts are a problem.

Many variations and modifications of the above example will now suggest themselves to one skilled in the art. The fixed delay line 20 which determines the activation range can be replaced with a selector for selecting different outputs of the pseudo-random code generator circuit 7 and applying them to the demodulator 19. For example if this pseudo-random code generator circuit consists of a shift register with appropriate feedback to produce the pseudorandom sequence then the selector might be arranged to select different stages of the shift register according to the desired range making use of a digital selector to address the appropriate shift register stage. This ability to derive the range function digitally is an important aid in for example altimetry where range tracking is desired.Another use of the digitally selectable range is to combat multipath problems in a communications system where a programmable correlator is needed in the receiver. These features can be incorporated in the apparatus according to the invention without the severe penalty which occurs in coventional receivers, or the loss in processing gain which occurs in the Doppler insensitive post-detection processors.

Although surface acoustic wave delay lines have been used here there is no reason in principle why they may not be replaced by electromagnetic delay lines, for example the eighty-tap delay line 22 could in principle be a tapped delay formed of electromagnetic wave guide.

The signal integration components can be arranged to operate at intermediate frequency, the signals being up-converted to radio frequency for propagation and down-converted after reception as described above. However, with improvements in surface acoustic wave technology and integrated circuits it will almost certainly become possible to make the integration components operable at the required radio frequency if it is below 2 GHz.

WHAT I CLAIM IS: 1. A receiver for a radar or telecommunications system receiving signals phase-modulated by a predetermined code sequence of code signals having a

**WARNING** end of DESC field may overlap start of CLMS **.

Claims

**WARNING** start of CLMS field may overlap end of DESC **. integration. In the embodiment an eighty-tap delay line was used; the number of taps used in any particular application is arbitrary but should be carefully chosen, because in general an N-bit IF pulse integrator (which in effect the tap delay line is) is quite frequency sensitive and the range of frequencies that it will successfully operate upon is only

1 wide to the 3 dB point, where N is the Nm number of taps or pulses to be integrated and T is the bit period. Thus in applications where Doppler frequency shifts are expected, the number of taps in the predetection integrator should be chosen to be as high as possible (to increase the predetection integration) while keeping the expected Doppler frequency shifts within the useful passband of the delay line integrator.

predetermined regular digit-signal rate, the said receiver including: a decoding means for applying a sequence of phase-shifts, complementary to those given by the code sequence, to the received signals; a first signal path and a second signal path having a common input connected to an output of the decoding means; a delay device connected in the first signal path, having a multitude of output tappings disposed so that the delays between adjacent tappings are integral multiples of the bitduration of the code signals, and means for combining signals from the output tappings;; and signal mixer means connected to receive at one input the combined signals from the delay device and at a second input signals from the second signal path, wherein the signal mixer means is arranged to derive an autocorrelation signal the magnitude of which is dependent on the relative phases of the received signals.

2. A receiver as claimed in claim 1 and wherein the delay device is an acoustic delay line having a plurality of transducers equispaced along its length, the distance between any two adjacent transducers being an integral multiple of the distance propogated by acoustic waves in the delay line in one bitduration of the code-signals, these transducers forming the output tappings of the delay line and being connected to a common output.

3. A receiver for a radar or telecommunications system as claimed in claim 1 associated with transmitter apparatus wherein a pulse generator circuit is connected to two input transducers at opposite ends of a surface wave acoustic delay line having an output transducer at an intermediate position on the delay line which is an integral number of acoustic wavelengths at the operating frequency of the delay line from each of the input transducers.

4. A receiver and transmitter apparatus as claimed in claim 3 and wherein the pulse generating circuit comprises a surface wave acoustic oscillator connected to energise a step recovery diode at a predetermined pulse repetition frequency.

5. A receiver and transmitter apparatus substantially as hereinbefore described with reference to the drawing accompanying the provisional specification.

6. A receiver substantially as hereinbefore described with reference to the drawing accompanying the provisional specification.