FR2568386A1 - VOLTAGE REFERENCE CIRCUIT WITH BAND PROHIBITED - Google Patents

Abstract

THE INVENTION RELATES TO PRECISION VOLTAGE REFERENCE CIRCUITS. A FORBIDDEN BAND VOLTAGE REFERENCE CIRCUIT INCLUDES IN PARTICULAR A FORBIDDEN BAND 2 CELL WHICH DETECTS AN IMBALANCE IN THE CURRENTS FLOWING IN THE CELL AND WHICH PRODUCES AN IMBALANCE SIGNAL WHICH IS TRANSMITTED EXCLUSIVELY BY LARGE BAND 40 NP TRANSISTORSGEURN 41, 50, 51 IN ORDER TO OBTAIN A VERY STABLE AND VERY HIGH GAIN AND A VERY LARGE BANDWIDTH. THE ABSENCE OF ANY ACTIVE PNP TRANSISTOR IN THE SIGNAL PATH PROVIDES A SINGLE VOLTAGE REFERENCE CIRCUIT BUT HAVING A VERY LARGE BANDWIDTH, WITH STABLE AND VERY REPRODUCIBLE PERFORMANCE. APPLICATION TO ELECTRONIC CIRCUITS.

Description

The present invention relates to voltage reference circuits at

forbidden band, and she carries

  more particularly on such circuits having a

  bandwidth much higher than can be achieved with previous bandgap circuits.

  Intra-band circuits are commonly used

  terdite in integrated circuits to produce

  invariant reference points with respect to the temperature

  which are used internally in the integrated circuit for various purposes, such as for example

  reference voltages or bias voltages.

  The various inter-band voltage reference circuits

  known as known have various disadvantages. Most of these circuits are very complex and occupy a large fraction of the area of a semiconductor chip. Some

  reference voltage circuits with forbidden band

  They do not have an appropriate voltage gain and are therefore excessively sensitive to variations in load currents that the bandgap circuit must provide.

  to a charging circuit. Some of the reference circuits

  previous bandgap voltage are not capable of

  It can not be adjusted to produce higher reference voltages, higher value, independent of the temperature. The state of the art is generally indicated in U.S. patents. Nos. R-30586, 4,325,017, 4,249,122, 4,339,707 and 4,064,448. The known prior art circuit of the applicant who is closest

  of the invention is described in a patent application

  co-owned and assigned to the plaintiff, entitled "IMPROVED BANDGAP REFERENCE CIRCUIT", filed May 25

  1984 by Stephen R. Burnham and Paul M. Henry.

  All previous bandgap circuits

  closest to that of the invention use

  Lateral PNP sistors as active devices in signal channels of amplifier stages that amplify differences between differential input signals

  a bandgap cell. The use of

  Lateral PNP sistors severely limit the width of

  amplification stages, unless techniques requiring complex circuits are used, because the bandwidth of the lateral PNP transistors in current integrated circuit technology is limited to

  about a megahertz. This constitutes an important limitation

  previous bandgap circuits, as a result of

  modern applications of inter-band circuits

  they lead to their use in integrated circuits.

  who control systems that are subject to transient currents of high frequency. If the circuit

  bandgap has a low bandwidth, the

  transients present on a node voltage of

  reference generated by the bandgap and

  other critical circuits are filtered and tend to dampen slowly producing ripples, resulting in faulty circuit operation.

  in the controlled system. The reference circuits with

  banned must therefore have high bandwidths.

  for a reason which is exactly the same as that for which the voltage regulator circuits

  generally have a high bandwidth, that is,

  to say so that they can present a very recovery

  fast after the presence of transient charge currents.

  The amplifier stages with transistors PNP la-

  which are used in connection with certain circuits

  reference to previous bandgap voltage, may

  can be used conveniently with current technology

  bipolar integrated circuits, and some of these

  cooked, like the one mentioned in the patent application

  joined by Burnham / Henry, must be carefully designed to

  achieve sufficiently high earnings.

  There remains, therefore, a need for a cir-

  bake voltage reference to perfect bandgap

  simple and broad bandwidth, providing

  an increased reference voltage independent of the time

  which is very stable and has a response to transient load variations that are possible

  to predict in a very precise way.

  The object of the invention is therefore to provide a

  bake voltage reference with band gap type

  with a bandwidth much greater than

  that of reference circuits with a band

earlier.

  Another object of the invention is to provide a band gap voltage reference circuit which is economical, has a large bandwidth and can be accurately predicted recovery characteristics in the presence of transient variations of

load.

  Briefly, and in accordance with one of his

  embodiments, the invention provides a feedback circuit

  bandgap bandgap which includes a bandgap cell with first bandgap

  and second differential inputs, the first of these

  being connected so as to receive a voltage of

  reference, a resistive ratio circuit intended to

  a "thermal voltage" at the second differential input

  of the bandgap cell, a charging circuit

  in the bandgap cell, intended to produce

  an amplified signal representative of a neck imbalance

  in the bandgap cell, and circuits for amplifying the signal again to produce a

  output reference voltage, wherein the amber signal

  folded only goes through vertical NPN transistors in

  the charging circuit and the amplifier circuit. The cir-

  fired reference voltage with bandgap and large

  bandwidth also includes a feedback circuit

  which responds to the output reference voltage by producing a feedback signal and applying it to the resistive ratio circuit to establish and correct the thermal voltage. The embodiment of the invention which is described uses a PNP current mirror circuit as a constant current charging device.

  asymmetrical for one side of the

  said, an NPN-type transmitter charge circuit for isolating

  the asymmetrical output of the

  said and to apply his signal to the entrance of a floor

  amplifier having an active NPN transistor and a tran-

  constant current load sistor connected in current mirror, type PNP. The amplified signal is isolated by an emitter-charged NPN transistor and is applied to a feedback NPN transistor and divider circuit

  which performs a resistive division of the reference voltage

  output frequency that is applied to the signal circuit

  resistive. The resistive ratio circuit is adjusted

  to achieve a precise compensation of the negative temperature coefficient of the voltage VBE of the reaction NPN transistor. The invention will be better understood when reading

  the following description of an embodiment, given

  born as a non-limiting example. The rest of the description

  refers to the accompanying drawings in which :. -

  Figure i is a schematic diagram of a band gap voltage reference circuit; Figure 2 is a detailed circuit diagram of an embodiment of the invention; and

  Figure 3 is a graph useful for describing

  operation of the circuits of Figures 1 and 2.

  Now considering Figure 1, we see

  an inter-band reference voltage generator circuit

  1, which comprises a bandgap framed by the phantom lines 2. The bandgap cell 2 comprises an NPN transistor 33 whose base is connected to the conductor 101 and whose emitter is

  connected to a constant current source circuit 3.

  The emitter of the transistor 33 is also connected to the emitter of the transistor 34, the base of which is connected to the ground conductor 5. The four arrows symbolizing

  the transmitter of transistor 34 indicate that its transmission area

  The collector of transistor 33 is connected by means of a conductor 6 and a resistor 17 to a conductor which is four times larger than that of transistor 33.

  voltage supply + V. Similarly, the col-

  transistor 34 drive is connected to the + V driver

  by a conductor 7 and a resistor 18.

  The base of transistor 33 is connected by the

  conductor 101 and the resistor 21 to the feedback conductor

  103, and is also connected by the resistance

this 16 to the ground conductor 5.

  The conductors 6 and 7, which constitute the output terminals of the bandgap cell 2, are respectively connected to the positive and negative inputs.

  of the high gain amplifier 8. The output of the amplifier

  8 is connected to the conductor 37, to which

the output voltage VREF.

  A resistor 19 is connected between the lead

  37 and the driver 102, on which appears, a voltage

  bandgap VBG. A resistor 20 is connected between the conductor 102 and the ground conductor 5. The

  driver 102 is also connected to the base of the transistor.

  tor NPN reaction 36, whose collector is connected to

  + V, and whose transmitter is connected to the response conductor

103.

  It will be noted that the preceding structure, representing

  shown in Figure 1, is not intended to describe specifically

  the invention, but to facilitate the description of the func-

  fundamental circuit of the bandgap circuit, a particular embodiment of which constitutes the present

  invention and which is shown in FIG.

  The circuit of FIG. 2 is similar to that of the simplified circuit of FIG. 1. In FIG. 1, the basic operation is as follows. The amplifier 8 detects any low value difference between the voltages on the conductors 6 and 7, and varies VREF by giving it a value which is feedback and forces VTH to have a value for which the leads 6 and 7 are precisely at the same voltage. It is assumed that the values of the resistors 17 and 18 are equal. The very high gain of the amplifier

  8 forces the circulation of equal currents in

sistors 33 and 34.

  Resistors 19 and 20 fulfill the function of "proportionally increasing" the voltage

  VBG on the driver 102, to establish the reference value

  VREF. We can express this differently by saying

  that VREF is divided according to the relationship between the

  19 and resistance 20, to establish the value of the VBG band gap voltage that is applied to the

base of the reaction transistor 36.

  At this point, it must be borne in mind that one purpose of the forbidden band reference voltage circuit 1 is to

  produce for VREF a level that is independent of the time

ture.

  The base-emitter voltage of the reaction transistor

  However, the temperature is not temperature independent, and decreases in fact as the temperature increases, as shown in curve 39 in FIG.

  3, the desired zero variation of the inter-band voltage

  said VBG on the conductor 102 is indicated by the reference 38. The reference 44 designates the hatched region in the figure

  3 which represents the difference between curve 39 and

  be 38, and represents the value of the voltage at node 103.

  It will then be noted that the tension on the oonduct-

  101, that is to say the voltage VTH, is equal to (kT / q) Log (N), denoting by N the ratio between the area of the emitter of the transistor 34 and the area of the emitter of transistor 33. It can be seen that the value of VTH increases

when the temperature increases.

  Those skilled in the art will readily see that it is possible

  select the ratio of resistance 21 to resistance

  16, so that the variation of the reaction voltage

  the driver 103 has an equal slope but

  opposite to the slope of line 39 in FIG. 3, and thus compensates for the temperature coefficient precisely

  negative of the base-emitter voltage of the

36.

  At this point, we will describe the mode of

  detailed description of the invention with reference to FIG.

  Reference 1A designates the improved band gap voltage reference circuit of the invention. When

  it was possible, we used the same numerical references

  to designate identical or similar components

in Figure 2 and in Figure 1.

  For some practical reasons, the structure of the band gap cell 2 in FIG.

  little different from the structure shown in Figure 1.

  First, the resistors 17 and 18 represented on the

  Figure 1 has been deleted and there is instead a

  PNP side sistor 9, operating as a current source, whose emitter is connected to + V by a feedback resistor 10A, and whose collector is connected to the

  7. In combination with PNP transistors

  11 and 13, the side PNP transistor 9 constitutes a cir-

  baked current mirror PNP type, as one skilled in the art will note easily. The base of transistor 9 is connected

  to the emitter of transistor 13, whose collector is connected

  to -V. The base of the transistor 9 is also connected to the base of the transistor 11, the emitter of which is connected to + V by the feedback resistor IOB. The collector of the transistor 11 is connected to the base of the transistor 13 by means of the conductor 14, and it is also connected to the

  collector of the NPN transistor 3B, which is one of the

  current source torsion of an NPN-type current mirror circuit, comprising the NPN transistors 3A, 3B, 3C,

  3D, 3E and 3F. A current source 29 attacks the transistor

  tor connected in diode 3F, which establishes the voltage

  emitter of transistors operating as current source

in this current mirror circuit.

  If desired, resistance can be used.

  these counter-reactions in series with the transmitters of

  each of the NPN transistors operating as a current source, to improve their pairing, but, for convenience, these

  resistors are not shown in Figure 2.

  Those skilled in the art will readily note that the collection of

  current source PNP transistor 9 is present at

  collector of the NPN transistor 34A a much higher impedance

  than can be achieved by resistance

  readable in practice, as the resistor 18 shown in Figure 1, in the current state of the art technology.

integrated circuits.

  The resistor 17 of FIG. 1 is not used

  in the bandgap cell in Figure 2.

  An NPN transistor 10 whose base is connected to a voltage

  reference voltage whose level is greater than 3 diode voltage drops to that of the ground conductor 5, is

  connected by its emitter to the collector of transistor 33A.

  The function of transistor 10 is to equalize the voltages

  collector-base of transistors 33A and 34A.

  To increase the nominal voltage across the resistor 16, the expedient of adding a diode-connected transistor 33B identical to the transistor 33A in series with the emitter of the NPN transistor 33A has been used. Similarly, the transistor connected to

  diode 34B is connected in series with the transmitter of the

  sistor 34A, whose collector is connected to the conductor 7. The emitters of the transistors 33B and 34B are connected together and are also connected to the current source transistor 3A, whose emitter is connected to -V. In the

  preferred embodiment of the invention, the transmission area

  The transistor 33B is equal to that of the transistor 33A, and the emitter area of the transistor 34B is equal to the emitter area of the transistor 34A. It should be noted that the higher the ratio

  areas of the emitters of transistors 34A and 34B,

  the corresponding transmitter areas of the transistors

  tors 33A and 33B, is large, minus the attenuation of

  due to resistances 21 and 16 must be strong; and the lower the attenuation, the more the bandwidth

Closed loop is great.

  While the output signal of the bandgap cell shown in Figure i is applied

  in the form of a differential input signal to the amplifier

  8, in FIG. 2 the amplified output signal

  that the bandgap cell 2 produces on the conductor

  7 is taken as an output signal "dissy-

  metric "instead of a differential output signal, and

  it is applied to the base of a transistor with a charge of

  40, whose collector is connected to + V.

  The emitter of the emitter charging transistor or

  separator 40 is connected to the conductor 27, which is itself

  even connected to the collector of the current source transistor 3C. The base of the transistor 40 is also connected to a terminal of the compensation capacitor 12.

  separator transistor 40 is also connected by the con-

  driver 27 at the base of a transistor amplifier of se-

  cond floor, 41, which is an NPN transistor, according to the invention. The emitter of the NPN transistor 41 is connected to the ground conductor 5, and its collector is connected by the conductor 22 to the other terminal of the compensation capacitor 12, as well as to the collector of the PNP transistor current source 23. The transistors Lateral PNPs 23 and 25 form another PNP-type current mirror circuit. The emitters of these two PNP transistors operating as a current source are connected to + V by respective feedback resistors O10C and 10D. Their bases

  both are connected by the driver 26 to the

transistor 25 reader.

  The output signal of the second amplifier stage

  2, appears on the conductor 22 and is transmitted by the conductor 22 to the base of the transistor

  NPN with transmitter charge 50, whose collector is connected

  to + V, and whose transmitter is connected to the conductor 28.

  The collector of the PNP transistor source of cou-

  rant 25 is connected by the conductor 26 to the collector

  of the NPN transistor current source 3D.

  The driver 28 connects the transmitter of the transistor

  tor NPN to 50 emitter load at the collector of the NPN current mirror 3E. The conductor 28 is also connected to the base of the NPN transistor 51, the emitter of which drives the conductor 37 on which is produced the increased bandgap reference voltage, VREF, independent

  of the temperature. The collector of transistor 51 is

  connected to + V. The conductor 37 is connected to the collector of the reaction NPN transistor 36, the emitter of which is connected to the feedback conductor 103, as in FIG. 1. The resistors 19 and 20 and the conductor 102 are connected in the same manner as on FIG. Figure 1, and perform the same function of proportionally increasing VBG

to produce VREF.

  The continuous operation corresponding to the

  asymmetrical plating carried out by the forbidden band cell of FIG. 2 differs from the operation of the cell shown in FIG. 1, in which the PNP current mirror circuit comprising the transistors

  9, 11 and 13, allows the current I2 crossing the tran-

  sitors 34A and 34B have a value of 50 microamperes, as determined by the current Il, which is also 50 microamperes, the latter being produced by the NPN transistor 3B, operating as a source or mirror

  current, and by the current source 29.

  The NPN current mirror transistor 3A gives a value of 100 microamperes to the current 13 flowing

  in his collector. The current 14 which passes through the

  sistors 33A and 34A is forced to 50 microamperes.

  Any imbalance between 12 and 14 produced by a variation

  tion of the voltage VTH on the conductor 101 is therefore

  folded on the driver 7 with a gain of about 500, and it is again amplified on the driver 22 with an additional gain of about 2000, and a reaction

  is established in a manner similar to that described above.

  in relation to Figure 1, to produce the correct

  necessary on the driver 101.

  An original aspect of the invention above consists in the exclusive use of NPN transistors

  as active devices in the amplifying parts

  of the forbidden-band circuit of Figure 2,

taking transistors 40, 41, 50, 51 and 36. These transistors

  NPN tors have a cut-off frequency much higher than that of lateral PNP transistors which can be

  practice in the present state of the art,

  Picnic. The following table gives examples of values

  for the resistors and the capacitor of the mode of

of Figure 2.

TABLE I

  Resistance 10A 2 kiloohms Resistance 10B 2 kiloohms Resistance 10C 2 kiloohms Resistance 10D 2 kiloohms Resistance 16 700 ohms Resistance 21 5 kiloohms Capacitor 12 2.5 picofarads The values of resistors 19 and 20 are not critical, and they are simply chosen to circulate an appropriate bias current in the emitter load transistor 51, and to give the desired proportional increase ratio

  to go from the band gap voltage VBG to the value de-

  Sire, independent of the temperature, of VREF. The values

  their precise currents produced by the NPN transistors operating in current mirror 3C, 3D and 3E are not

nothing critical.

  Those skilled in the art will note that at ambient temperature

  biante, the value of VTH is about 71 millivolts, if we give the value 4 to N, that is to say the ratio between

  transistors of the transistors 34A and 34B, and

  res emitters of transistors 33A and 33B.

  It is typically desired that the value

  VBG is equal to the bandgap voltage of the silicone

  cium, which is about 1.14 volts. It is desirable that the sum of the base-emitter voltage VBE (36) of the transistor 36 and the voltage on the conductor 103 is equal to the bandgap voltage VBG. Figure 3 shows these two

  voltage components and their sum over a period of time

  theoretical temperatures. Mathematically, we can extrapolate the value of VBE (36), to obtain VBG at a temperature equal to 0 Kelvin. At this same temperature, the extrapolated value of VTH is equal to zero. The shaded part 44 of the drawing represents the voltage at the node 103, which is equal to R21 + R16 VTH, by designating by R16 and R21 the values of the R16 resistors 16 and 21. The appropriate choice of the ratio of the resistances and N allows the voltage at node 103 to increase as a function of temperature, in such a ratio

  that the voltage at the bandgap node 102 remains practically

  tically constant in the interesting range of 200

at 4000 Kelvin.

  It should be noted that with a precise analysis, the VBE component (36) has a slight curvature and that the band gap voltage varies slightly depending on

  temperature. These effects can be included in the design

  of the reference circuit of FIG. 2.

  It has been found that the embodiment of the invention

  The invention described above had a very wide bandwidth of about 30 megahertz. This is much greater than the amplifier bandwidth of all bandgap voltage reference amplifier circuits

  known in the prior art, all of which use amplifiers

  single-stage gain drivers, formed by current mirror PNP circuits or bootstrap-type feedback circuits, sometimes referred to as floating current mirror circuits. These last circuits have so

  characteristic a lower amplifier bandwidth

than one megahertz.

  The circuit shown in FIG. 2 does not require

  while not having a higher semiconductor chip area than single-stage PNP floating current amplifiers, and it is much simpler from a circuit analysis point of view. Improved bandwidth

  of the circuit represented in FIG.

  come two amplification stages instead of just one. Meadow-

  The first amplification stage comprises the right half of the amplification cell 2 and the PNP current source transistor 9, and the second stage comprises the NPN transistor.

  41 and the current source PNP transistor 23.

  greatly improved maneuvers result from the fact that the amplified signal does not pass through any of the current source PNP transistors; these transistors simply act as high impedance load devices. The cutoff frequency of vertical NPN transistors in

  bipolar integrated circuits is typically

  than greater than 500 megahertz. For PNP transistors

  side, the cut-off frequency is typically

  tick of about one megahertz only. As a result, high frequency circuits with side PNP transistors, and amplifiers comprising such circuits, have

  bandwidths limited to a value of the order of

  gahertz, unless you use techniques that make use of

to complex circuits.

  By way of comparison to show the simplicity of design and analysis of the circuit of FIG. 2 with respect to the above-mentioned amplification stages at

  "floating PNP current mirror", incorporated into cir-

  burned bandgap voltage generators of art

  previous, we can note that the gain of the circuit of the invention

  tion with its two stages of NPN amplification is

  simply gm1gm2RllR12, where gml is the transcon-

  ductance on the right side of the forbidden cell cell 2, by Rill 'impedance at node 7, by gm2 transconductance NPN transistor 41 and by R12 the impedance at node 22. This gain is very stable, it can be accurately predicted and

  it has a value of about 1,000,000. As mentioned earlier

  However, the gain of the floating PNP current mirror circuit can theoretically be close to infinity but, in practice,

  when we actually build these devices, phenomena

  Subtle effects in the operation of the circuit mean that the actual gain is frequently much lower than expected and often much less than 1,000,000. In addition, the bandwidth of the PNP current mirror circuits is still much less than the bandwidth. 30 megahertz of

present circuit.

  This extremely high bandwidth allows the compensation capacitor 12 to have a value

  very low, typically about 2 to 5 pico-

  farads only, compared to the beach of 30 to 40 picfarads

  compensation capacitors which are required to

  characteristic in reference circuits of

  previous bandgap.

  The recovery period of the reference circuit

  this band-gap of the present invention, under the effect of transient load variations output, is reduced by an order of magnitude or more, compared to

  pre-banded reference voltage reference circuits

  using PNP side transistors active in

  their amplification stages. It is therefore considered that the cir-

  baked-in-line of Figure 2 represents a

  much safer, less risky, which does not require

  no compromise in size, complexity or performance

  and still has a much higher bandwidth

  and much lower recovery times than

  known band gap voltage reference circuit.

  Although the invention has been described with reference to a particular embodiment, those skilled in the art will appreciate

  various modifications to the way in which

  described, without departing from the scope of the invention. For example, one can proceed in various ways other than those described, to obtain equal current densities in the NPN transistors located in each of the right sides

  and left of the bandgap cell.

Claims (15)

  A high bandgap bandgap reference voltage generator circuit, characterized in that it comprises, in combination: (a) a bandgap cell (2) having first and second differential inputs, the first input differential being connected to receive a first voltage of   reference (5); (b) a reporting circuit (16, 21) for   born to apply a thermal voltage (VTH) to the second differential input of the bandgap cell; (c) a charging circuit (9, 10, 11, 13) in the cell   bandgap (2), intended to produce a first   amplified signal representative of a current imbalance in the bandgap cell (2); (d) amplifying means (40, 41, 50, 51) responsive to the first amplified signal producing an output reference voltage (VREF), said amplifying means including a number of active transistors through which the first amplified signal; and (e) temperature responsive reaction means (36,103) responsive to the   output reference voltage (VREF) by producing a   reaction and applying this reaction signal   the reporting circuit (16, 21), the gear ratio   dependent on the feedback signal to compensate for the thermal voltage (VTH), so that the voltage   (VREF) is virtually independent of   the temperature sensitivity of the reaction means (36, 103), and all the active transistors (40, 41, 50, 51)   are NPN transistors, the amplification means do not   holding any transistor crossed by the first ampli-   which is not an NPN transistor.   2. Reference voltage generator circuit at   wideband bandgap according to the   cation 1, characterized in that the bandgap cell (2) comprises first and second NPN transistors (34A, 33A),   whose bases are respectively connected to the first   re and second differential inputs, and that the   charging circuit comprises a first PNP transistor sour-   current (9) whose collector is connected to the collec-   reader of the first NPN transistor (34A), and the first amplified signal is produced on the collector of the first   PNP transistor current source (9).   3. Reference voltage generator circuit   bandgap band according to the   2, characterized in that the amplification means   cation comprises a third NPN transistor (41) whose emitter is connected to receive the first   reference voltage (5), and a second PNP transistor   current (23) whose collector is connected to the collec-   reader of the third NPN transistor (41), and first coupling means (40) for coupling the collector of the first current source PNP transistor (9) to the base   of the third NPN transistor (41), in order to apply the first   first amplified signal at the base of the third transistor NPN (41).   4. Reference voltage generator circuit at   wideband bandgap according to the   3, characterized in that the temperature responsive reaction means comprises an NPN transistor of   reaction (36), whose emitter-base voltage decreases when   that the temperature increases, and that the circuit of   port comprises first and second resistors (16, 21) connected in series between the emitter of the reaction NPN transistor (36) and the first reference voltage (5), the   connection point (101) between the first and second   sistances being connected to the second differential input   the bandgap cell (2).   5. Reference voltage generator circuit at   wideband bandgap according to the claim   4, characterized in that the ratio of the first and second   resistance (16, 21) is selected to   compensate for a negative temperature coefficient of the NPN reaction sistor (36).   6. High Bandwidth Bandgap Reference Voltage Generating Circuit According to the Claim   cation 5, characterized in that it comprises a condensation   compensator (12) which is connected between the   third NPN transistor (41) and the collector of the   first PNP current source transistor (9).   7. Reference voltage generator circuit   bandgap band according to the claim   2, characterized in that the band cell   terdite (2) comprises a first diode-connected transistor (34B) whose base and collector are connected   to the emitter of the first NPN transistor (34A), and a second   diode-connected transistor (33B), whose base and collector are connected to the second NPN transistor (33A), and whose emitter is connected to the emitter of the first   transistor connected in diode (34B).   8. Reference voltage generator circuit   bandgap band according to the claim   3, characterized in that the first coupling means comprises a first emitter-charged NPN transistor (40) whose base is connected to the collector of the emitter.   first PNP current source transistor (9), and whose transmitting   is connected to the base of the third NPN transistor (41).   9. Reference voltage generator circuit at   wideband bandgap according to the   8, characterized in that the amplification means comprise a second emitter-charged NPN transistor (50) whose base is connected to the collector of the third NPN transistor (41), and a third emitter-charged NPN transistor (51) whose base is connected to the transmitter of the   second transistor with an emitter load (50), and whose emitter   is connected to the output reference voltage (VREF) and is also connected to the base of the NPN transistor reaction (36).   10. Reference voltage generator circuit   bandgap band according to the claim   9, characterized in that it comprises a third   resistor (19) which is connected between the transmitter of the third   second emitter load transistor (51) and the base of the   NPN reaction transistor (36), and a fourth resistor   which is connected between the base of the NPN feedback transistor (36) and the first reference voltage (5). 11. Reference voltage generator circuit   bandgap band according to the claim   6, characterized in that the geometric configuration   of a transistor belonging to the bandgap cell (2) and the geometrical configuration of the first current source PNP transistor (9) are selected to give a gain of about 500 for the bandgap (2).   12. Reference voltage generator circuit   bandgap band according to the claim   11, characterized in that the geometric configuration   of the third NPN transistor (41) and the geometrical configuration of the second current source PNP transistor (23)   are selected so as to give a gain of second   about 2000 for the amplification means.   13. Reference voltage generator circuit   bandgap band according to the claim   1, characterized in that all active NPN transistors (40, 41, 50, 51) are vertical NPN transistors in an integrated circuit.   14. Method for obtaining a reference voltage   this bandgap, characterized in that it comprises the following operations: (a) a band cell is established   prohibited (2) having first and second different entries   and connect the first differential input   at a first reference voltage (5); (b) applying a thermal voltage (VTH) to the second differential input of the band gap cell (2) by means of a resistive ratio circuit (16, 21); (c) producing on a first output node (7) of the cell   band (2) a first amplified signal   sensing current imbalance in the bandgap cell (2) by means of a first load circuit (9, 11, 13) connected to the first output node   (7); (d) amplifying the first amplified signal for   to obtain an output reference voltage (VREF), in   passing this first amplified signal through a set of active transistors (40, 41, 50, 51); and (e) producing a feedback signal in response to the output reference voltage (VREF), and applying this REF signal.   reaction to the resistive ratio circuit (16, 21) by   of an active reaction transistor (36) to compensate for the thermal voltage (VTH), so that the voltage   (VREF) is virtually independent of   the sensitivity to the temperature of the reaction transistor   (36), and all the active transistors (40, 41, 50, 51,   36) through which the first amplified signal or the   The reaction signal, as well as the active transistors of the bandgap cell (2), are vertical NPN transistors.   15. The method according to claim 14, characterized   in that the bandgap cell (2) comprises first and second NPN transistors (34A, 33A), the bases of which   are respectively connected to the first and second   differential paths, and in that the first load circuit comprises a first PNP current source transistor.   (9) whose collector is connected to the collector of the first   first NPN transistor (34A), the first amplified signal   rerouting on the collector of this first source PNP transistor current (9).