Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/0003—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
  • H04B1/0028—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at baseband stage
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/0003—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain
  • H04B1/0028—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at baseband stage
  • H04B1/0032—Software-defined radio [SDR] systems, i.e. systems wherein components typically implemented in hardware, e.g. filters or modulators/demodulators, are implented using software, e.g. by involving an AD or DA conversion stage such that at least part of the signal processing is performed in the digital domain wherein the AD/DA conversion occurs at baseband stage with analogue quadrature frequency conversion to and from the baseband
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04B—TRANSMISSION
  • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
  • H04B1/06—Receivers
  • H04B1/16—Circuits
  • H04B1/26—Circuits for superheterodyne receivers
  • H04B1/28—Circuits for superheterodyne receivers the receiver comprising at least one semiconductor device having three or more electrodes

Definitions

• the invention relates to a method for processing an analog signal whose frequency spectrum has, over a determined bandwidth, two main lobes separated by a frequency band where the power is negligible.
• the invention also relates to a device for processing a corresponding analog signal.
• the field of the invention is that of radio navigation by satellite.
• Radionavigation systems such as GPS, GLONASS, are positioning systems in three dimensions, based on the reception of signals emitted by a constellation of satellites.
• the signal transmitted by a satellite is typically composed of a carrier modulated by a spreading code and possibly data; BPSK modulation (English acronym for Binary Phase Shift Keying) which gives a carrier with phase jumps of ⁇ each time the binary code is changed, is commonly used in current systems.
• BPSK modulation English acronym for Binary Phase Shift Keying
• FIG. 1a shows a carrier of period T, a random binary spreading code of frequency F co of, the resulting signal, modulated according to a BPSK modulation (designated BPSK signal for simplicity) and the envelope of the corresponding frequency spectrum.
• the frequency spectrum of a BPSK signal has (in power) an envelope of the form IT -. 2, 1,, ⁇ , r -. sin ⁇ x. ,,
• FIG. 1b shows the signal resulting from the same carrier and the same spreading code, but this time modulated according to a BOC modulation (designated BOC signal for simplicity), and the envelope (in power) of the corresponding frequency spectrum, which is of the form 1 / F C ode • sinc 2 (
• the frequency spectrum of a BOC signal has two identical main lobes spaced on either side of f p (respectively -f p ), with each of the adjacent secondary lobes, as shown in FIG. 1b.
• BOC modulation can be considered to be BPSK modulation applied after having previously multiplied the carrier by a subcarrier whose frequency f sp is often a multiple of f p .
• the signal transmitted by the satellite is an analog signal which, after having traveled the distance between the satellite and the receiver, is converted by the receiver into a digital signal for further digital processing.
• This conversion includes a step of sampling the spectrum of the signal received by the receiver, followed by a digitization step. Sampling is carried out according to a sampling frequency fe.
• the sampling frequency fe must be greater than or equal to the bandwidth of the spectrum.
• the spectrum of a BOC signal, from which the lobes are separated has a wider frequency band than that of a BPSK signal, as illustrated in FIGS. 1a) and 1b): it follows that the sampling of a BOC signal is produced at a higher sampling frequency than that of a BPSK signal.
• the use of a high sampling frequency has the disadvantage of inducing an additional cost and an increase in consumption.
• the invention proposes a method for processing an analog signal whose frequency spectrum has, over a determined bandwidth, two main lobes separated by a frequency band where the power is negligible, mainly characterized in that it includes a sampling step according to a determined sampling frequency, and prior to this sampling step, a step consisting in carrying out a frequency translation of the two main lobes towards one another in order to reduce the width bandwidth and therefore the sampling frequency.
• This translation can be obtained by two methods.
• the step of translating the lobes can be obtained by multiplying the analog signal by a signal of the cos type ( ⁇ t), ⁇ being determined from the frequency of the subcarrier and the bandwidth of the main lobes; the translation of the main lobes having generated parasitic lobes, the method further comprises, prior to sampling, a step of filtering the translated lobes, with a view to eliminating the parasitic lobes.
• It preferably includes a prior step of converting the analog signal into baseband.
• the analog signal can be a signal modulated according to a BOC type modulation.
• the BOC signal comprising a carrier, a code and a subcarrier, respectively having determined frequencies
• the method comprises a step of digitizing the sampled signal and a step of demodulating the digitized signal based on the use of a locally generated code and subcarrier, the local code being generated from the frequency of the code, the local subcarrier being generated from the frequency of the subcarrier determined and reduced during of the lobe translation stage.
• the analog signal is for example a radio navigation signal.
• the subject of the invention is also a device for processing an analog signal whose frequency spectrum has, over a determined bandwidth, two main lobes separated by a frequency band where the power is negligible, characterized in that it comprises a frequency translation element of the main lobes towards each other capable of reducing the bandwidth.
• the invention relates to a receiver of a radio navigation system comprising such a device.
• FIG. 1a schematically represents a carrier of period T, a random binary spreading code equal to 1, -1, 1, 1, ..., and the resulting BPSK signal emitted, expressed as a function of time and the envelope of the corresponding frequency spectrum, expressed in power
• the figure 1b) schematically represents the same carrier code as those in FIG.
• FIGS. 3a), 3b) and 3c) schematically represent (expressed in power) the envelope of the frequency spectrum of the BOC signal of FIG.
• FIG. 5 schematically represents a first embodiment of a device for processing an analog signal according to the invention
• FIG. 6 schematically represents a second embodiment of a device for processing an analog signal according to the invention
• FIG. 7 schematically represents the control loop of the carrier and that of the code and the subcarrier in the case of a conventional BOC signal processing device
• FIG. 8 schematically represents an element for calculating the local phase common to the code generator and to the sub generator - carrier in the case of a conventional BOC signal processing device
• FIGS. 9 a) and 9 b) schematically represent the local code (fig 9a) and the local subcarrier (fig 9b) in function ion of the local phases expressed in chip, in the case of a device for processing a conventional BOC signal
• FIG. 9 a) and 9 b) schematically represent the local code (fig 9a) and the local subcarrier (fig 9b) in function ion of the local phases expressed in chip, in the case of a device for processing a conventional BOC signal
• FIG. 9a) and 9 b schematically represent the local code (fig 9a) and the local subcar
• FIG. 10 schematically represents the servo loop of the carrier and that of the code and the subcarrier in the case of a device for processing a BOC signal according to the invention
• FIG. 11 schematically represents an element for calculating the phase of the local code and an element for calculating the phase of the local subcarrier in the case of a device for processing a BOC signal according to the invention
• FIGS. 12 a) and 12 b) schematically represent the local code (fig 12a) as a function of the local phase expressed in chip and the local subcarrier (fig 12b ) as a function of the local phase expressed in cycles, in the case of a device for processing a BOC signal according to the invention.
• the method according to the invention aims to reduce the sampling frequency of a BOC signal.
• the BOC signal is conventionally converted to baseband, possibly passing through a prior conversion to intermediate frequency Fi.
• Bandpass filtering is generally applied before the conversion (s) so as to eliminate certain secondary lobes; low-pass filtering is generally applied after the conversion (s).
• the spectrum of the signal BOC in FIG. 1b has been represented, at the output of the receiver antenna (fig 2a), after its conversion into an intermediate frequency Fi (fig 2b) and then into the base band (fig 2c).
• the bandwidth of the spectrum is then Bj n itiaie or Bi.
• the signal BOC after its conversion to intermediate frequency Fi is a real signal whereas after its conversion to baseband, the signal which comprises an I channel and a Q channel (in quadrature with respect to the I channel), is complex.
• the side lobes of the frequency band located between the two main lobes are preferably eliminated by filtering in order to avoid aliasing during sampling.
• 0 b e or B
• the sampling frequency fe is greater than or equal to the bandwidth of the spectrum of the signal BOC, in this case Bi.
• a first analog method consists in multiplying the channels I and Q by a signal in cos (early) represented in FIG. 3b, ⁇ being of the form 2 ⁇ (fsp-fsientd) -
• the spectra before and after multiplication are respectively represented in FIGS. 3a and 3c; after multiplication, each lobe is then centered on a reduced subcarrier frequency, f succd - On af serie ⁇ BI / 2.
• a final filtering eliminates the parasitic lobes to avoid aliasing during sampling.
• Another method makes it possible both to translate the main lobes towards one another and to sample: this is obtained by carrying out sampling according to a specific sampling frequency fe s .
• This frequency fe s is determined from the following conditions, intended to avoid that during this specific sampling, there is an overlap between lobes.
• FIGS. 4a and 4b These conditions are illustrated in FIGS. 4a and 4b, on which the spectrum before sampling and the spectrum after sampling are respectively represented as desired, that is to say without overlapping of lobes. More particularly represented in FIG. 4b, the first and second main lobes corresponding to the line located at the frequency 0: to comply with the condition of non-overlap, the frequency band of this first lobe must be located below the frequency N.fe s and beyond the frequency (N-1/2). fe s , which results in conditions (1), (2) and (3).
• This digital method has the advantage of carrying out two steps (approximation of the lobes and sampling) in one and also makes it possible to avoid having to carry out by an analogical method the double multiplication by the signal cos ( ⁇ t).
• the method according to the invention can also be applied to analog “pseudo-BOC” signals obtained from two signals transmitted by the same source and synchronously, on two distinct and close frequencies, each signal being treated as a lobe of the spectrum of a signal BOC. This is for example the case for the Galileo system with signals transmitted in the frequency bands E1 and E2.
• main lobes are identical, but the invention also applies in the case where the main lobes are not.
• the analog signal is digitized.
• the analog signal thus converted into a digital signal is then processed according to the desired application.
• FIGS. 5 and 6 We will now describe an example of an analog signal processing device included in a receiver of a positioning system, shown in FIGS. 5 and 6.
• the analog signal whose carrier has a frequency fp is filtered by means of a bandpass filter 2 which can be a ceramic filter.
• the signal is then preferably amplified by a low noise amplifier 3.
• a signal is obtained whose spectrum corresponds to that of FIG. 2a, that is to say free of certain secondary lobes.
• the conversion to baseband of this amplified signal is obtained by multiplying it by means of a multiplier 4 on a first channel designated channel I by a signal of the form cos (2 ⁇ .fp.t) and by means of another multiplier 4 'on a second channel designated channel Q by a signal of the form sin (2 ⁇ .fp.t).
• the signals of the form cos (2 ⁇ .fp.t) and sin (2 ⁇ .fp.t) come from a local oscillator 5.
• the spectrum of the complex signal (channel I and Q) thus obtained is of the form of that of Figure 2c.
• the signal thus multiplied is filtered by means of a 6 or 6 'bandpass filter which can be an RC filter (comprising a resistance R and a capacitance C) or a surface wave filter (SAW filter in English) so as to eliminate the secondary lobes of the frequency band located between the two main lobes.
• a 6 or 6 'bandpass filter which can be an RC filter (comprising a resistance R and a capacitance C) or a surface wave filter (SAW filter in English) so as to eliminate the secondary lobes of the frequency band located between the two main lobes.
• SAW filter surface wave filter
• the implementation of the analog method is obtained by arranging as shown in FIG. 5, on each channel I and Q a multiplier 7 or 7 ′ capable of multiplying the signal by a signal of the form cos ( ⁇ .t) from the local oscillator 5, then a low-pass filter 8 or 8 'making it possible to eliminate the parasitic lobes as indicated in FIG. 3c.
• the signal obtained is then sampled by means of a sampler using a sampling frequency fe greater than or equal to 2BI and digitized by means of a digitizer which produces a digital signal, these sampler and digitizer being grouped in a converter 9 or 9 .
• the implementation of the digital method is obtained by directly disposing as shown in FIG. 6 on each channel I and Q a sampler using a sampling frequency fe s and a digitizer which produces a digital signal, these sampler and digitizer being grouped in a 10 or 10 'converter.
• the digital processing of the signal obtained on each of the I and Q channels is then carried out according to the desired application.
• a BOC signal consists mainly of a carrier, a subcarrier and a code.
• the purpose of signal processing is to demodulate the digitized BOC signal into carrier, subcarrier and code for retrieve the measurement of the propagation delay from the difference between the time for code transmission by the satellite and the time for code reception by the receiver.
• Demodulation is done by correlating the digitized BOC signal with locally generated carrier, subcarrier and code.
• carrier and code tracking loops the code loop including the tracking of the subcarrier; these loops control the local carrier, subcarrier and code phases with respect to the phases of the carrier, subcarrier and code of the BOC signal received, from measurements resulting from the correlations.
• the measurement of the delay on the code and of the initial Doppler effect is made in an acquisition phase also called hooking phase which consists in testing in open loop several hypotheses of position of the code and of the Doppler effect until what the result of the correlation indicates by a high energy level that the phase shift between the received signal and the local signal is minimal. Then, we refine the search and then close the loops.
• demodulation steps are obtained by means of a demodulator comprising servo loops, an example of which is shown in FIG. 7.
• a demodulator comprising servo loops, an example of which is shown in FIG. 7.
• the digital signal at the input of the servo loops is as we have seen. previously a complex signal comprising an I channel and a Q channel.
• the correlation of the received signal with the local signal is done first by multiplying by means of a multiplier 11 the digitized signal by a signal of the form e " ' ⁇ , ⁇ being the phase of the local carrier.
• the signal obtained is then multiplied by means of a multiplier 12 on a so-called point channel (hence the notation l p and Q p for point I channel and point Q channel) by a signal representative of the code and subcarrier modulation, and by summing the results of these multiplications obtained at different times by means of an integration-summation element 14.
• the signal representative of the code modulation and of the subcarrier was obtained by multiplying by means of a multiplier 13 , a signal representative of the code locally generated from ⁇ , by a signal representative of the subcarrier generated locally from ⁇ , ⁇ and ⁇ being respectively the phase of the local code and of the local subcarrier, which are in fai t identical in this case.
• phase discriminator 15 which deduces therefrom a carrier difference which is a real signal and which is injected into a carrier loop corrector 16.
• a phase calculation element 17 which can be a numerically controlled oscillator (“Numerically Controlled Oscillator” in English), calculates the phase ⁇ of the local carrier as a function of the carrier speed coming from the carrier loop corrector 16, and the frequency of the carrier without Doppler effect called frequency carrier's heel.
• Carrier speed is the propagation speed of the carrier measured at reception: we deduce the variation in frequency of the carrier due to the Doppler effect.
• This phase ⁇ thus enslaved is used by a carrier generator to generate a local carrier of the form e " "' ⁇
• the correlation of the received signal with the local signal is also done on a so-called delta channel (hence the notation l ⁇ and Q ⁇ for channel I delta and channel Q delta), by multiplying by means of a multiplier 21 the digitized signal multiplied by a signal of the form e " "' ⁇ by a so-called delta signal.
• This delta signal from a summator 20 is the difference of the signal representative of the code modulation and of the subcarrier which has undergone an advance by means of a device 18 making it possible to anticipate the signal relative to that of the channel. punctual and a delay by means of a device 19 making it possible to delay the signal compared to that of the punctual channel.
• the results of these multiplications obtained at different times are summed by means of an integration-summation element 22.
• phase discriminator 23 which deduces therefrom a code deviation which is a real signal and which is injected into a code loop corrector 24.
• An element of phase calculation 25 which can be a numerically controlled oscillator (“Numerically Controlled Oscillator” in English), calculates the phases ⁇ and ⁇ of the local code and the local subcarrier according to the code speed (identical to the speed of subcarrier) from the code loop corrector 24 and the stub frequency of the code.
• the code speed is the speed of propagation of the code measured on reception: we deduce the frequency variation of the code due to the Doppler effect.
• the phases ⁇ and ⁇ of the code and of the subcarrier which are identical, are thus enslaved then respectively used by a code generator 26 to generate the local code and by a subcarrier generator 27 to generate the local subcarrier .
• FIG. 8 shows the detail of a code phase calculation element 25. It includes a converter 30 of the code speed expressed in m / s, in a measurement expressed in Hz of the frequency variation due to the Doppler effect, the conversion being effected from the code chip; the phase calculation element further comprises a summator 31 of this measurement of the Doppler effect and the stub frequency of the code and an integrator 32 transforming this new frequency into a phase ⁇ .
• FIG. 9 represents a) the local code generated by the code generator 26 as a function of the local phase expressed in chip, the chip being the wavelength of the code;
• FIG. 9 b) represents the local subcarrier generated by the subcarrier generator 27 as a function of the local phase also expressed in chip, since the same element of phase calculation 25 was used for the two generators 26 and 27 .
• the sampling frequency used at the receiver has been reduced by means of a translation towards each other of the main lobes of the spectrum of the received signal.
• This translation reduced the frequency of the subcarrier which became f sest d-
• the frequency of the reduced subcarrier then being different (weaker) from the frequency of the code, it is thus necessary to dissociate the element of computation of the phase of the subcarrier which takes into account the frequency of the reduced subcarrier, of the element for calculating the phase of the code which takes into account the frequency of the code as shown in FIG. 10.
• FIG. 11 shows the detail of the phase calculation elements 25 and 28 respectively used for the code and for the subcarrier.
• the phase calculation element 25 used for the code is the same as that of FIG. 8.
• the phase calculation element 28 used for the subcarrier comprises a converter 33 for the code speed (which is the same as the subcarrier speed) expressed in m / s, in a measurement expressed in Hz of the frequency variation due to the Doppler effect, the conversion taking place from the wavelength of the subcarrier expressed in cycles; the phase calculation element further comprises a summator 34 of this measurement of the Doppler effect and of the reduced heel frequency of the subcarrier and an integrator 35 transforming this new frequency into a phase ⁇ .
• the Doppler effect is independent of the reduction in the frequency of the subcarrier which occurs only at the level of the receiver.
• FIG. 12 represents a) the local code generated by the code generator 26 as a function of the local phase expressed in chip;
• FIG. 12 b) represents the local subcarrier generated by the subcarrier generator 27 as a function of the local phase expressed in cycles, since a phase calculation element 28 specific for the subcarrier has been used upstream of the generator 27.

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• Engineering & Computer Science (AREA)
• Computer Networks & Wireless Communication (AREA)
• Signal Processing (AREA)
• Position Fixing By Use Of Radio Waves (AREA)
• Networks Using Active Elements (AREA)

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• 2002
  • 2002-12-17 FR FR0216000A patent/FR2848743A1/fr active Pending
• 2003
  • 2003-12-12 WO PCT/EP2003/050998 patent/WO2004055540A2/fr not_active Application Discontinuation
  • 2003-12-12 US US10/539,621 patent/US20060128343A1/en not_active Abandoned
  • 2003-12-12 EP EP03799563A patent/EP1581819A2/de not_active Withdrawn
  • 2003-12-12 CA CA002510191A patent/CA2510191A1/fr not_active Abandoned

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