Classifications

• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
  • H02M3/00—Conversion of DC power input into DC power output
  • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
  • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
  • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
  • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
  • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
  • H02M3/338—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement
  • H02M3/3385—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only in a self-oscillating arrangement with automatic control of output voltage or current

Definitions

• the present invention relates to the field of voltage converters of the low-voltage switching power supply type.
• the invention applies more particularly to isolated power supplies, that is to say having no common point between the input voltage (for example, the AC supply network) and the regulated DC output voltage.
• the insulation is obtained by means of a transformer whose primary winding is associated with a switch and whose secondary winding is associated with a diode and a capacitor supplying the output voltage.
• the invention relates more particularly to so-called self-oscillating converters in which the switch is controlled by frequency modulation, in contrast to converters whose switch is controlled by pulse width modulation.
• FIG. 1 represents a classic example of a switching power supply of the type to which the present invention applies.
• Two input terminals P, N receive an alternating voltage V ac , for example the mains voltage.
• This voltage V ac is the subject of a rectification, for example a full-wave rectification by means of a bridge 1 of diodes.
• the alternating input terminals of bridge 1 are connected to terminals P and N.
• the rectified output terminals 2, 3 of bridge 1 provide a voltage which is impeded- Really smoothed by means of a capacitor C1 connected between terminals 2 and 3.
• Terminals 2 and 3 constitute the input terminals of the switching power supply proper.
• the converter of FIG. 1 constitutes a converter known as "flybac" or “with energy recovery” in which a transformer 4 with reverse phase points has its primary winding 5 connected in series with a switch 6 between the terminals 2 and 3.
• the phase point of winding 5 is connected to one terminal of switch 6 and the other terminal of winding is connected to terminal 2.
• a secondary winding 7 of transformer 4 is associated with a capacitor C2 at terminals 8, 9 from which the continuous output voltage V or t- is supplied.
• the phase point of winding 7 is connected to terminal 8 by a diode D2, the cathode of diode D2 being connected to terminal 8.
• the other terminal of winding 7 is connected to terminal 9.
• the switch 6 is controlled by a circuit 10 (CTRL), the role of which is to cyclically open and close the switch 6.
• CTRL circuit 10
• the opening of the switch 6 is caused by comparison of the current flowing in this switch when it is closed with respect to a reference value.
• a detector 11 is generally used (for example, a resistor) in series with the switch 6 and the measurement result of which is supplied to the circuit 10.
• the auxiliary winding 8 is in direct phase relation with the secondary winding 7.
• a detection of the end of demagnetization of the winding 8 corresponds, firstly approximation, upon detection of the end of demagnetization of the winding 7 ..
• the phase point of the winding 8 is connected to an input terminal of the circuit 10 while the opposite terminal of the winding is connected to the ground 3.
• the voltage across the terminals of the auxiliary winding 8 is monitored by circuit 10. Demagnetization is considered to be complete when this voltage drops below a predetermined voltage threshold. The switch 6 is then closed. It will be remembered that, as the phase points of the primary and auxiliary windings are reversed, the voltage across the terminals of the auxiliary winding 8 is negative outside these demagnetization periods (when the switch 6 is closed) .
• the auxiliary winding 8 also serves to supply a local supply voltage to the control circuit 10.
• a local supply capacitor C3 is connected to the supply terminals of the circuit 10.
• a positive electrode 12 of the capacitor is connected, by a diode D3, to the phase point of winding 8, the anode of diode D3 being connected to the phase point.
• the other electrode of capacitor C3 is connected to ground 3.
• FIG. 2 represents a classic example of a circuit 10 for self-oscillating control of a voltage converter.
• the switch 6 generally consists of a MOS transistor whose drain is connected to the primary winding 5 and whose source is connected, by a resistor Rll, to ground 3.
• the resistor Rll plays the role of a converter current-voltage to an input terminal of a first comparator 13 of the circuit 10.
• the role of comparator 13 is to compare the current in switch 6 with respect to a reference value VRJ_ supplied to the other terminal of comparator 13.
• the reference voltage V ⁇ j of comparator 13 is chosen as a function of the output voltage or ⁇ - desired and of the transformation ratio between the primary and secondary windings.
• the voltage V Rj _ conditions the power of the converter which is proportional to the value of the inductances of the transformer 4 and to the square of the current in the primary winding 5 when the switch 6 is closed.
• the output of comparator 13 is sent to the reset input (R) of a flip-flop RS 15 or the like whose non-inverted output Q provides the control signal from switch 6.
• the output of flip-flop Q is generally applied to the gate of transistor 6 via an amplifier 16 (driver).
• the input S for setting the flip-flop 15 is connected to the output of a second comparator 17 whose function is to detect the end of demagnetization of the auxiliary winding 8.
• An input of the comparator 17 receives a voltage reference VR V of an element 18.
• V RV is chosen to correspond to a voltage threshold below which it is considered that demagnetization has ended. Ideally, the voltage V ⁇ y is zero.
• the other input of the comparator 17 is connected, by a diode D4, to the phase point of the winding 8, the anode of the diode D4 being connected to this phase point.
• the output of comparator 13 switches to l high state when the current in the switch 6, multiplied by the value of the resistance R11, exceeds the voltage R j, while the comparator 17 switches to the low state when the voltage across the terminals of the auxiliary winding 8 (neglecting the voltage drop in the diode D4) becomes lower than the voltage VRV-
• this state has priority at the level of flip-flop 15 which provides a level outlet bottom. This causes the opening of the switch 6, therefore a start of demagnetization.
• a self-oscillating circuit Compared to a switching power supply operating in pulse width modulation (PWM), a self-oscillating circuit has the advantage of low cost. In particular, it is not necessary to provide an oscillator generating a sawtooth signal, the width of the pulses of which is modulated.
• PWM pulse width modulation
• the inputs of flip-flop 15 can be associated with front-shaping circuits (trigger).
• a delay element may be provided at the output of the comparator 17 according to the sensitivity of the latter. A low sensitivity is then compensated for by increasing the detection threshold Vpy and by delaying the output signal.
• Self-oscillating control circuit converters are also known which allow regulation of the output voltage.
• these converters 1 impose a measurement of the secondary voltage of the transformer and, by way consequently, a galvanic isolation element for transmitting the measured value to the control circuit. This considerably increases the cost and constitutes an often unacceptable disadvantage of this type of converter.
• a converter with a self-oscillating control circuit is therefore conventionally incompatible with regulation of the output voltage V out , while preserving a low cost.
• the present invention aims to propose a new converter type switching power supply with self-oscillating control circuit which allows regulation of the output voltage without the need to resort to a galvanic isolation element between the secondary of the transformer and the control circuit.
• the invention aims more particularly to propose a solution which respects the conventional RS rocker structure or equivalent of a self-oscillating converter.
• the invention also aims to propose a solution which is compatible with protection of the converter against a short circuit in the secondary, always without galvanic isolation element.
• the invention further aims to propose a solution whose control circuit can be integrated.
• the present invention provides a voltage converter comprising a transformer, a primary winding of which is connected in series with a switch for cutting a supply voltage, and a secondary winding of which is associated with a capacitor providing a continuous low voltage, and a self-oscillating switch control circuit comprising means for detecting the end of demagnetization of an auxiliary winding of the transformer in order to cause the closing of the switch and means for detecting the current in the switch in the closed state in order to cause it to open when this current reaches a setpoint, and means for making the setpoint variable as a function of the voltage across the terminals of the auxiliary winding.
• the converter comprises means for reducing the conduction time of the switch when the voltage across the terminals of the auxiliary winding deviates from a range of predetermined values.
• said setpoint is reduced when the voltage across the auxiliary winding leaves said range of values.
• the converter comprises means for inhibiting the means for making the setpoint variable during a start-up phase where the voltage delivered by the converter has not yet reached a minimum regulation value.
• the converter comprises a variable voltage generator supplying the setpoint on an input of a first comparator, the other input of which receives a voltage which is a function of the current in the switching switch, the output of the first comparator triggering the opening of the switch.
• the variable voltage generator includes a first analog error amplifier receiving a measured voltage proportional to the voltage across the auxiliary winding and providing a voltage proportional to the positive difference between the voltage measured and a first predetermined threshold, and a second analog error amplifier receiving said measured voltage and supplying a voltage proportional to the positive difference between a second predetermined threshold and the measured voltage, the second threshold being less than the first threshold.
• the results of the two error amplifiers are summed to provide a control signal from a variable voltage source.
• variable voltage source delivers, in the absence of correction by one of the error amplifiers, a predetermined nominal voltage.
• the converter comprises means for inhibiting the operation of the second error amplifier as long as a local supply voltage of the control circuit has not reached a predetermined value.
• the analog error amplifiers are hysteresis amplifiers.
• FIGS. 1 and 2 which have been described previously are intended to expose the state of the art and the problem posed;
• FIG. 3 represents an embodiment of a converter according to the invention;
• FIG. 4 very schematically shows an embodiment of a variable reference generator of a converter according to the present invention;
• FIGS. 5A to 5G illustrate, in the form of timing diagrams, the operation of a converter according to the invention.
• a characteristic of the present invention is to use an image of the voltage across the terminals of the secondary winding supplying the output voltage to modulate the reference voltage of the comparator conditioning the duration of conduction of the switching switch. In other words, provision is made to make the reference voltage used to bring about the opening of the switch for cutting the supply voltage variable, and to make this variation a function of the output voltage. Regulation of this output voltage is thus obtained.
• Another characteristic of the present invention is to provide voltage regulation
• the voltage across the terminals of the auxiliary winding is monitored with respect to two thresholds framing a desired nominal voltage and it is ensured that the voltage remains between these two thresholds.
• a first threshold corresponds to a maximum voltage value in normal operation.
• a second threshold (lower than the first) indicates an excessive current draw (collapse of the output voltage). Between the two thresholds, the switching frequency is not modified and the variable reference voltage is maintained at a nominal value.
• Another characteristic of the present invention is to obtain the image of the output voltage without using additional galvanic isolation means. For this, advantage is taken of the existence of the auxiliary winding used for local supply of the control circuit of the switching switch.
• FIG. 3 represents an embodiment of a voltage converter according to the present invention. This figure is to be compared to that of FIG. 2 and illustrates the modifications to be made compared to the conventional converter.
• a switching switch 6 for example, an N-channel MOS transistor
• a secondary winding 7 of the transformer 4 charges a capacitor C2 via a diode D2, the output voltage V out being taken from the terminals of capacitor C2. Windings 5 and 7 have reversed phase points.
• the switch 6 is controlled by the output (Q) of a flip-flop RS 15 (in practice, by passing through an amplifier 16) of a control circuit 30.
• the input to set (a) of the flip-flop 15 is controlled by a comparator 17 detecting demagnetization across an auxiliary winding 8 of the transformer.
• a diode D4 connects the phase point of the winding 8 to the negative terminal of the comparator 17.
• the positive terminal of the comparator 17 receives a fixed voltage reference V ⁇ y, supplied by an element 18.
• a capacitor C3 supplies a voltage V ⁇ local supply of the control circuit 30.
• the electrode 12 of the capacitor C3 is connected, by a diode D3 in series with a resistor R7, to the phase point of the winding 8 and, by resistor RI, to the terminal positive 2 of application of the voltage to be cut.
• this voltage corresponds to a rectified voltage supplied by a diode bridge 1 supplied by an alternating voltage V ac .
• a capacitor C1 connects the rectified output terminal 2 of the bridge 1 to a ground terminal 3 to smooth the rectified voltage.
• the reset input R of the flip-flop 15 is connected to the output of a comparator 13, the positive input of which receives a current measurement in the switch 6.
• the positive input of the comparator 13 is connected to a terminal 12 of input of circuit 30 which measures a voltage as a function of current in switch 6.
• a resistor R11 is used, in a conventional manner, as current-voltage converter.
• the negative input of comparator 13 receives a variable reference voltage V ⁇ supplied by a generator 20.
• the essential modification to be made to the circuit of FIG. 2 for the implementation of the invention is to replace the generator (14, FIG. 2) of fixed reference voltage, input of the RS flip-flop reset comparator, by generator 20.
• the function of the generator 20 is to decrease the reference voltage V ⁇ when the output voltage of the converter deviates from a predetermined regulation range. This causes a reduction in the conduction time of the switch 6, hence a reduction in the energy stored by the transformer. This results in an increase in the frequency of the self-oscillating system and, consequently, a decrease in the energy supplied to the secondary of the transformer. If the output voltage increases, this means a drop in the current drawn by the load and the regulation operated by the invention minimizes consumption while avoiding the appearance of harmful overvoltages at the level of the load. If the output voltage decreases, this means that there is an overload, or even a short circuit. It is therefore necessary to protect the converter by reducing this current.
• the generator 20 of the invention must however be compatible with the operation of the circuit at start-up where the output voltage is initially zero.
• the invention provides for inhibiting regulation when the system is started.
• a nominal voltage NO is generated for the generator 20 and this value is not modified as long as the output voltage has not reached a minimum regulation value.
• a voltage V A -r j is taken proportional to the voltage across the terminals of the auxiliary winding 8.
• This voltage ⁇ JJX is taken from an input terminal 31 of the circuit 30.
• terminal 31 is connected to a first electrode of a capacitor C5, the other electrode of which is connected to ground.
• Terminal 31 is also connected to point middle of a resistive voltage divider bridge.
• This bridge consists of a resistor R5 in series with a resistor R6 between ground 3 and the cathode of a diode D5, the anode of which is connected to the anode of diode D3.
• a resistor R7 connects the anodes of the diodes D3 and D5 to the phase point of the winding 8.
• the voltage V AT j ⁇ is a function of the transformation ratio between the auxiliary 8 and secondary windings 7 as well as the values of the resistors R5 , R6 and R7.
• the voltage v AT j ⁇ corresponds to the voltage across the winding 8, affected by the proportionality coefficient fixed by the resistive bridge consisting on the one hand of resistors R7 and R6 in series and on the other hand resistance R5.
• the resistor RI associated with the capacitor C3 is used, as before, to start the circuit.
• the diode D3 serves to protect the measurement circuit of the starting circuit (RI, C3) in order to prevent the voltage V A ⁇ j ⁇ from being influenced by the voltage at the primary of the transformer.
• the capacitor C5 is used to absorb variations in the voltage ⁇ r.
• the circuit 30 has been shown in the form of an integrated circuit delimited by a dotted line grouping together all of the regulation functions of the invention.
• the switching frequency of a converter according to the invention depends on the duration during which the switch 6 is closed and on the demagnetization duration of the auxiliary winding (therefore of the secondary winding).
• the closing time of the switch 6 depends on the input voltage (voltage across the capacitor C1), the amplitude of the current in the switch 6 and the inductance of the primary winding 5 of the transformer.
• the amplitude of the current in the switch 6 corresponds to the variable reference voltage VRR divided by the resistance R11.
• the duration of opening of transistor 6 depends on the duration of demagnetization of the secondary and auxiliary windings. bonded and, therefore, essentially the coupling of the transformer 4.
• FIG. 4 represents an embodiment of a variable voltage reference generator 20 according to the present invention. Such a generator is intended to be installed in a voltage converter of the type illustrated in the figure.
• the generator 20 comprises a first error amplifier 21 (E / A), a first input (positive) of which receives a voltage V & ⁇ proportional to the voltage across the terminals of the auxiliary winding of the transformer.
• a negative input of the error amplifier 21 receives a fixed reference voltage VR Q supplied by a voltage generation element 23.
• a second error amplifier 22 receives, on its negative input, the voltage TAUX and, on its positive input, a fixed reference voltage V R0 ! supplied by a voltage generation element 24.
• the voltage references of the error amplifiers 21 and 22 are different and condition the regulation range, that is to say a range of output voltages where the switching frequency is not modified.
• the outputs of amplifiers 21 and 22 are combined in an adder 25, the output of which controls a variable voltage source 28.
• the source 28 provides the variable voltage reference V-gg.
• the source 28 is dimensioned so that, at rest, it provides a predetermined nominal voltage reference V ⁇ - Q M which is a function of the nominal output voltage for which the converter is designed.
• the error amplifier 22 is activated by means of an ENA signal supplied by a comparator 26 (COMP).
• a positive input of the comparator receives the local supply voltage V ⁇ of the control circuit (not shown in FIG. 4).
• a negative input of comparator 26 receives a fixed voltage reference VR ⁇ , corresponding to a level below which it is considered that the circuit is in a starting phase.
• the comparator 26 is used to invalidate the error amplifier 22 as long as the local supply voltage is not sufficient, therefore in particular during the start-up period of the converter.
• the signal ENA masking the results of the error amplifier 22 is obtained by comparing the voltage V ⁇ across the terminals of the capacitor C3 with respect to the threshold Vjyj supplied by an element 27.
• Error amplifiers 21 and 22 are analog amplifiers which therefore supply a signal which is a function of the amplitude of the difference between their respective inputs.
• the comparator 26 is a digital comparator providing a high or low output state depending on whether the voltage V ⁇ is higher or lower than the reference V ⁇ .
• the amplifier 21 delivers the difference between the voltage V ⁇ T JX and the reference VR Q
• the amplifier 22 delivers the difference between the reference RQ I and the voltage VAUX- ' Only positive differences are taken into account. In other words, the output voltage is regulated so as to be brought back into a regulation range where it is considered to correspond to its nominal value.
• VRO, R Q ⁇ / jyi are supplied by conventional means.
• a circuit 32 (SUPPLY, FIG. 2) supplying the supply voltages of the various elements of the circuit 30 from a reference of type voltage "bandgap".
• the constitution of such a circuit is conventional and is not the subject of the present invention.
• FIGS. 5A to 5G illustrate, in the form of timing diagrams, an example of operation of a converter equipped with the variable voltage generator 20 of FIG. 4 in four operating phases.
• FIG. 5A shows an example of the shape of the voltage ⁇ .
• FIG. 5B represents the signal ENA supplied by the comparator 26.
• FIG. 5C represents the voltage VAUX-
• FIG. 5D represents the voltage V 2 ⁇ at the output of the amplifier 21.
• FIG. 5E represents the voltage V22 at the output of 1 ' amplifier 22.
• FIG. 5F represents the voltage V25 at the output of the summator 25.
• FIG. 5G represents the voltage Vj ⁇ of variable reference.
• a first phase A illustrated in FIGS. 5 corresponds to the start-up phase of the converter.
• the voltage V Q - ⁇ is zero, the capacitor C3 (providing local power) being fully discharged.
• V M JJ ⁇ instant tl
• the control circuit is not supplied and no voltage reference can be generated. It follows that the output signal of the various error amplifiers is zero, as is the control signal 25 of the voltage RR. From time tl, we begin to be able to detect the voltage which begins to grow as the supply voltage is cut.
• the variable reference voltage Vj ⁇ begins to increase from time tl following the growth of the local supply voltage.
• the error amplifiers 21 and 22 are hysteresis amplifiers. Consequently, the threshold v R0 will be reached by the auxiliary voltage ⁇ ⁇ j before the voltage v dd reaches its limit value, and the threshold VR Q 1 will preferably be reached before the voltage Vdd reaches the value Vpjj.
• the outputs of the error amplifiers 21 and 22 remain zero as does the output of the summator 25.
• the output voltage VRR of the generator then remains at its nominal level jqQM 'fixed by the dimensions of the various components (phase B).
• the choice of the values R g and RQ I depends on the precision desired for the regulation.
• phase C The nominal phase B continues until an instant t4 when it is assumed the appearance of an output overvoltage (phase C).
• an overvoltage corresponds, for example, to a reduction in the current drawn by the load supplied by the converter.
• the voltage s j x starts to increase at the same time as the output voltage.
• the voltage 2 ⁇ begins to grow following the increase in the voltage ⁇ jj x- This results in a signal corresponding to the level of the output of the summator 25.
• the signal 25 then causes a decrease in the reference voltage VRR in an inversely proportional manner. This decrease in the variable reference voltage acts on the appearance of opening edges of the cutting switch 6.
• Phase D corresponds to current regulation.
• Another advantage of the present invention is that it does not require any measurement at the transformer secondary, nor galvanic isolation means to transfer a measurement result.
• the invention uses the voltage V ⁇ T X to vary the reference of a comparator using information linked to the current in the switch.
• the present invention is susceptible of various variants and modifications which will appear to those skilled in the art.
• the hysteresis of the analog error amplifiers 21 and 22 will be adapted as a function of the desired and / or necessary stability for the control signal of the variable voltage source 28.
• the values of the resistors and of the capacitors conditioning the voltages and operating frequency of the self-oscillating converter of the invention depend on the application and are within the reach of those skilled in the art from the functional indications given above.
• the switching frequency will be several kHz (preferably, greater than 20 kHz to be non-audible), therefore clearly greater than the frequency of the voltage Vac if it is a sector.
• a converter of the invention is intended for output voltages between a few volts and a few tens of volts for powers ranging from a few Watts to a few tens of Watts.
• the implementation of the invention does not exclude the use of elements for shaping the switching fronts (trigger) as well as a delay element at the output of the demagnetization comparator, as is the case. case for a conventional converter.

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