DE19538575A1 - Inductive proximity sensor with high switching speed - Google Patents

Abstract

The distance between sensing coil L1 and object is measured by the pulse width proportional to the phase-shift Ph1 between the coil current and voltage. An analogue comparator K1 measures the coil voltage relative to a reference dc voltage and switches a logic gate when the coil voltage becomes larger or smaller than the reference. A second comparator K2 measures the voltage across a reference impedance Zref relative to a second dc voltage and switches a gate. The output from K1 and the inverted K2 output are fed into a logic gate which generates a pulse sequence F of pulse width proportional to the phase-shift at the oscillator frequency. A Schmitt trigger can be added to introduce hysteresis in the switch point.

Description

The invention relates to a device for measuring the complex impedance of a lossy coil whose impedance depends on a non-electrical measured variable is dependent, and an inductive sensor or Eddy current sensor.

A coil dependent on a non-electrical measured variable - hereinafter referred to as Sen Sorspule or measuring coil called - is used in operational measurement, process monitoring and sensors in a variety of ways as an inductive distance sensor or inductive Proximity switch for distance measurement and as an eddy current sensor for destruction free material testing. The non-electrical parameters can be from Ab was a measurement object - hereinafter referred to as the control flag - or the physika mical properties of the test object, such as electrical conductivity σ, the Per meability µ or derived quantities.

If such a, mostly metallic, object to be measured enters the alternating electromagnetic field the coil of an inductive sensor, the complex impedance of the coil changed by the non-electrical measurands. The non-electrical measurands often affect only one component of the complex at a suitable frequency Impedance, for example, affects the electrical conductivity of the test object Loss resistance of the coil or the permeability of the object to be measured Kitchen sink. However, the coil is always lossy without a measurement object, i. H. an inductive In addition to the basic inductance, the sensor has additional ohmic losses.

The electrical measuring circuits should if possible only the proportion of the impedance or detect a quantity derived from the impedance by the non-electrical Measured variable is dependent. A measuring circuit for evaluating the change in com plex impedance or a derived quantity is inherent in its behavior depending on the coil. The quality Q or the loss factor D and the tempera the behavior of the coil determine the sufficiently high sensitivity Properties of the measurement. In other words, the measurement properties become whole largely determined by the losses of the coil. By suitable choice of measuring circuit is attempted, only the part of the complex impedance dependent on the measured variable or measure a quantity derived from it, for example the effective resistance or the inductance or the quality or the loss factor. There are numerous measuring known circuits to solve this problem, but have few circuits has proven itself in practice. Mostly, the evaluation circuits are Improvement of the measuring properties directly adapted to the sensor coil.

The behavior of electrical and electronic circuits is unique depending on the components used. The quality and the temperature behavior Resistors and capacitors are controlled with modern technologies bar; the properties of modern semiconductor circuits are also characterized by ge suitable choice of circuit can be determined; however, only a few circuits are known that improve the quality and temperature response of a lossy coil.  

The most important non-contact sensors for process control and plant transfer are inductive distance or proximity sensors or proximity switches and Proximity initiators. An inductive proximity sensor contains a coil with a directional high-frequency electromagnetic field. Mostly han is used for this Standard cylindrical single or half-shell cores made of ferromagnetic Ferrite material. This creates a rotationally symmetrical pattern on the exposed legs metric single-shell cores a preferred direction of the electromagnetic high frequency quenzfeld. An electric or magnetic becomes in this directional high-frequency field conductive or ferromagnetic material - a so-called control flag - brought, occurs a damping of the magnetic and electrical components of the Hochfre quenzfeldes and thus a damping of the coil.

A commercially available inductive proximity sensor contains a high-frequency oscillator with an LC resonant circuit. The coil of this LC resonant circuit generates the directed one high frequency electromagnetic field. Due to the damping described above electromagnetic radio frequency field takes the amplitude of the radio frequency Vibrations of the oscillator. The amplitude of the oscillator radio frequency oscillation can act as a measure of the attenuation of the directed electromagnetic high frequency field are used, that is, as a measure of the distance of a control flag.

The amplitude of the oscillator high-frequency signal is used to evaluate the damping generally rectified and filtered with a low pass filter. This same directional and filtered radio frequency amplitude signal is a DC voltage or DC signal and can either directly into an analog signal to display the Distance of a control flag or with the help of a subsequent evaluation circuit can be converted into a switching signal, the switching signal at a defined distance of the control flag changes its switching state. Particularly inductive Proximity switches that work according to the function described last serve in numerous embodiments and in large numbers for system control and Plant monitoring. As a circuit for generating the high-frequency vibrations a Meissner oscillator circuit is very often used.

The distance that can be achieved and used depends on an inductive proximity sensor area, d. H. the distance of the control lug from the "active" surface of the coil, in depend essentially on the size and properties of the high-frequency coil. The Damping by the control flag causes a change in the quality Q of the coil. At Approaching the control flag, the quality Q is in the form of a maximum value Q₀ an S curve reduced to a minimum value. The switching point or the meas It is best to lay rich in the steepest part of the S-curve, i. a. is that on Turning point. If you want to enlarge the measuring range, you also have to use the shallower ones Use parts of the S-curve. The measurement is in a flatter part with a much greater uncertainty, since a certain evaluable quality level here means a large change in the distance s of the control flag.

The influence of the ambient temperature on the relative quality Q / Q₀ of a coil shows that the good decreases with increasing temperature. The influence of temperature on the Coil quality limits the usable distance measuring range with inductive proximity sensors because the change in quality due to the influence of temperature in  an intended temperature working range can be larger than that by a Tax flag effect change.

In many applications and in many embodiments, distance measurement is used rich selected so low that no special measures to a Temperaturkom pensation are necessary. In critical applications, temperature is used dependent resistors, for example thermistors or PTC thermistors, around the temperature to compensate for the switching distance. However, this additional effort leads only satisfactory results in a limited temperature range. A it cannot significantly increase the usable distance measuring range can be achieved.

A lossy coil is described by the complex impedance Z. An AC voltage U applied to the coil leads the AC current I flowing through the coil by the phase angle ϕ _UI , which is less than 90 ° (degrees) by the loss angle δ. The behavior of a lossy sensor coil of an inductive proximity sensor is described by an equivalent circuit in which the field loss resistance R _F through an ohmic parallel resistor in parallel with a reactance X _L through a pure inductance L and in series with this parallel connection through the winding resistance R _{CU of} the coil an ohmic series resistor is taken into account, as shown, for example, in German Offenlegungsschrift OS 38 14 131 in FIG. 2a. Represented in this equivalent circuit known per se

• - the field loss resistance R _{F is} the active losses of the alternating electromagnetic field due to eddy currents and magnetic reversal losses in the coil core and in the test object (control flag),
• - The reactance X _L the inductive losses of the measuring coil and the electromagnetic alternating field due to the permeability µ of the test object (control flag) and the dielectric losses due to the capacitance of the measuring coil and
• - The winding resistance R _CU, the direct current losses through the copper resistance of the coil wire and the alternating current losses through the skin effect in the coil wire.

The quality of a coil is given by the ratio of the reactance Im ( Z ) and the loss resistance Re ( Z ), where Z is the complex impedance (impedance) of the coil.

The influence of the ambient temperature on the impedance Z and on the quality Q of a coil is essentially caused by the temperature dependence of the loss resistances. The temperature coefficient α of the DC resistance of the copper winding of the coil is known to be approximately 3.95 · 10 ^-3 / K. The temperature responses of the other loss resistors are usually smaller, but their size depends on the type of coil.

The quality and thus the temperature response of a coil is a composite Size that is strong in different designs and even from specimen to specimen  Is subject to fluctuations. Therefore a compensation or a reduction of the temperature response at least separately for each type of coil will.

As a measure of the distance of a measurement object (control flag) from the sensor coil, the measured variables complex impedance Z or, alternatively, quality Q or impedance Z of the measurement coil are measured. The quality is often measured with an LC resonant circuit or the impedance is measured in the non-destructive material test. The disadvantage of these measurements is the dependency on the temperature and the material of the test object and thus the reduced accuracy of the measured variable. In quality measurement with an LC resonant circuit, this resonant circuit is excited to resonate vibrations in an oscillator, the oscillation amplitude of the oscillator being a measure of the quality.

The impedance Z can be determined from the simultaneous effective value measurement of the voltage U _eff applied to the measuring coil and the current I _eff flowing through the measuring coil.

The impedance Z contains the active component R (loss resistance) and the frequency-dependent reactive component X _L = ωL. The impedance Z results from the complex impedance Z = R + jωL.

If the effective resistance R is negligible, the reactive resistance X _L results from the current and voltage measurement.

In order to obtain the inductance L from the measurements, the frequency f = ω / 2π of the measuring voltage must also be known. To measure the complex impedance Z that is actually to be detected, two separate measuring processes must be used either

• - The effective resistance R as the real part Re ( Z ) and the reactance X _L as the imaginary part Im ( Z ) of the complex impedance or
• - The impedance Z and the phase angle ϕ _UI between the current I flowing through the measuring coil and the voltage U applied to the measuring coil are measured. In most applications, only one of the measured variables shown above is measured, and only the one that responds as selectively as possible to the distance to be measured between the control lug and the measuring coil and that is influenced as little as possible by disturbance variables such as temperature and electromagnetic radiation, so that the accuracy and resolution is as high as possible.

In German Offenlegungsschrift 35 13 403, a procedure is specified according to which is the temperature coefficient of the winding copper resistance of the voice circuit coil used to compensate for the temperature coefficient of the quality of the resonant circuit is a voltage proportional to the copper resistance of the voice circuit coil is applied to the resonant circuit with a second coil. This temperature comm However, compensation can only be realized under very special conditions. The Kupferwi the level of the second coil must be significantly greater than the copper resistance of the Voice coil, while the inductance of both coils must be the same. This In practice, conditions are very difficult to meet.

In German Offenlegungsschrift 38 14 131 a procedure and a Vor Direction specified in which the effective resistance of the sensor coil as field loss resistance stand is measured directly with a power measurement. This field loss resistance the sensor coil, caused by the loss of the electromagnetic field due to the stray field and due to electromagnetic damping by Control flag, when choosing a suitable frequency depends on the distance s Tax flag. When using a four-wire circuit, the measurement is in one wide temperature range independent of temperature, as this is the temperature dependent Winding resistance of the coil (also called copper resistance) and the lei The resistance of the four leads is not included in the measurement. By phasing correct multiplication of the coil alternating current and the induced coil alternation voltage and subsequent low-pass filtering becomes the active power in the coil measured, which is inversely proportional to the field loss resistance.

In a similar way, the phase can be multiplied by 90 ° phase-shifted coil alternating current and the induced coil alternating voltage voltage and subsequent low-pass filtering measure the reactive power in the coil also inversely proportional to the coil reactance or the coil inductance act. For easy separation of the primary and secondary side, i.e. H. of the coil current and the induced coil voltage, the induced coil voltage is measured using a second winding decoupled. In the simplest case, the two windings are wrapped identically and bifilar.

German Offenlegungsschrift 43 28 097 describes a device for measuring the Impedance of passive sensors (inductive, capacitive and ohmic sensors) with double-locked phase locked loop (PLL) is described, wherein in one feedback line a measuring phase shifter as a low-pass or High-pass filter is arranged with the sensor component. The frequency analogs Signal evaluation of such a phase locked loop has a very high measuring sensitivity possibility. Furthermore, the frequency-analog output signal can be done in a simple manner further process and convert it into a digital signal.

In all previously known inductive proximity switches, the switching speed is d. H. the cut-off frequency for switching operations, through low-pass filtering for generation the distance-dependent output signal (usually a DC voltage) and especially  by the energy stored in the resonant circuit and in the electromagnetic field determined and thus significantly lowered. The cutoff frequency of the measurable switching device gears is caused by electrical reloading and by inertia of the stationary Ren, alternating electromagnetic field of the sensor coil determined. The frequency behavior Overall, the behavior shows the switching frequency of an inductive proximity switch ten of a low pass filter, d. H. the lowest frequency limit of the measuring chain determines in essentially its entire frequency response. The low pass filter mentioned above serves in addition to the generation of the distance-dependent output signal i. a. also the interferer pressure of internal and external interference signals. The cut-off frequency of the switch is usually gears at around 1 kHz, only with considerable electronic means is a limit to achieve a frequency of 10 kHz, but only with a lower interference suppression or a poorer signal-to-noise ratio.

Furthermore, the temperature response of the impedance of the sensor coil, in particular the winding resistance R _CU , limits the measuring sensitivity here, so that only those measuring methods should be improved that take into account or compensate for the temperature response of the sensor coil.

Furthermore, the analog output signal or the switching point of the material Control flag dependent, whereby the measuring accuracy of the inductive proximity sensor is deteriorating. This dependency is determined either by a material actuator or through complex compensation measures with auxiliary coils on the secondary side a transformer or by a differential method with a mathematically-stale Technical signal processing to compensate for the material actuator eliminated. in the German utility model G 94 12 765.4, this problem is addressed using a dif rentialverfahren by measuring the voltage induced in the sensor coil and the Resonance frequency of an LC resonant circuit formed by the sensor coil and by solved a mathematical-technical link between these two signals. Here too the switching speed of the inductive proximity sensor because of the low-pass filtering of the two signals and the energy stored in the resonant circuit reduced.

It can be seen that correction methods are known for some influencing and disturbance variables are. However, no method and no device is known to date which are based on the Procedure itself the influences and disturbances of the measurement described above reduce or avoid entirely.

The invention is therefore based on the object of the detectable cutoff frequency an inductive proximity sensor or eddy current sensor or the maximum measurable Switching speed of an inductive proximity switch or proximity initiator to increase significantly without deteriorating its measuring properties, so that the measurement accuracy and susceptibility to interference are maintained or improved.  

The invention solves the problem primarily according to claim 1, according to which on a sensor coil of an inductive proximity sensor, the phase shift between applied AC voltage U or induced AC voltage U _ind and by flowing AC current I directly during each half cycle or a multiple of each half cycle of current and Voltage is measured with electronic means. The solution is based on the knowledge that the complex impedance Z or a derived measured variable of the sensor coil, here preferably the phase shift ϕ _UI between current and voltage, can be measured as often and as immediately as possible after a possible change, if the impedance Z without one Delay is determined by additional filters. Each low-pass and band-pass filter with its cutoff or resonance frequency f₀ delays the signal between its input and output by the delay time T₀ = 1 / f₀.

The use of digital technology is becoming increasingly prevalent in measurement technology their high signal processing speed up to cut-off frequencies of a few GHz and because of their greater immunity to interference and easier signal processing compared to analog technology. The processing takes place in digital measurement technology Processing of the signals in a value-discrete and time-discrete manner. The necessary quantization of the For the above reasons, measurement signals should occur at the beginning of the electrode, if possible that the entire measuring device with simple and inexpensive digital switching elements and assemblies can be built. In the simplest case, the value of an analog signal to be measured (current or voltage) as a binary signal be given, d. i.e. whether the signal is larger or smaller than a comparison signal. This task can be accomplished for a voltage with a voltage comparator. For example, the value zero can serve as the reference voltage, so that at a pure symmetrical AC voltage with no DC component of zero crossing use as a switching condition for the binary output signal of the voltage comparator can be detected. The switching condition can be finite in the same way Comparative DC voltage are not equal to zero. This is preferred for a AC voltage with a DC voltage component applied. The same procedure as described above can in the same way with an analog AC signal be performed.

In an inductive proximity sensor according to the invention, the measurement of the complex impedance Z or, alternatively, the measurement of the phase shift ϕ _UI at a zero crossing of the measuring coil alternating voltage U and the measuring coil alternating current I or when these alternating signals are exceeded or undershot compared to a comparison direct signal is carried out . Voltage signals are preferably used so that only the voltage comparators described above can be used. For this purpose, the AC signal I is converted with a current-voltage converter, preferably a pure reference impedance Z _ref , for example an ohmic resistor, a low-loss capacitor or a low-loss coil, or a mixed combination of these components into a proportional AC voltage U _I. When using a capacitor or a coil in the reference impedance Z _ref , an additional phase shift ϕ _I between the measuring coil alternating current I and the alternating voltage U _{I is} generated, which must be taken into account with the correct sign in the phase shift ϕ _UI .

The phase shift ϕ _UI between current and voltage of the sensor coil results in

and is measured as the time interval T _UI between the comparator switching point t _{U of} the sensor coil voltage U and the comparator switching point t _I of the sensor coil current I , the frequency f or the period T of the alternating current or alternating voltage signal having to be taken into account.

To determine the phase shift ϕ _UI , the time interval T _{UI is} measured based on the period T of the sensor coil alternating signal. By switching the two output signals of the two comparators by means of digital gates, flip-flops, memory elements or flip-flops or combinations of these digital logic elements, a pulse train with a pulse repetition frequency or pulse rate is obtained which is derived from the frequency f of the measuring coil AC voltage and is therefore identical or twice the size. Digital logic elements are also known which contain such comparators as an input stage. In the simplest case, such an analog comparator can be an operational amplifier that is not fed back. The duty cycle of the pulse train thus corresponds to the phase shift ϕ _UI . The measurement of the duty cycle can be converted into an analog or digital signal using analog and digital electrical or electronic means (not described in more detail here) and evaluated mathematically and in terms of circuitry, no low-pass filtering being necessary.

The complex impedance Z is measured, for example, by means of the above-described measurement of the phase shift ϕ _UI or the apparent power Z. The apparent power Z can be carried out, for example, by means of an effective value measurement of the measuring coil current I _eff and the measuring coil voltage U _eff .

So is the effective resistance

and the reactance

the sensor coil is determined, the calculation of the quantities R and X _L can be carried out by means of a mathematical circuit combination with known electrical engineering means. The complex impedance Z of the sensor coil of an inductive proximity sensor is composed as described above from the parallel connection of the field loss resistor R _F and the pure reactance X _L and the winding resistance R 1 as a series resistor. The largest part of the active component of the complex impedance Z is the strongly temperature-dependent winding resistance R _CU , which is referred to below as R₁.

It is therefore the object of the present invention to provide a measuring method and a measuring device in which the temperature dependency of the coil is no longer included in the measurement of the distance of the control flag, so that, for example, in the application as a proximity sensor, the distance or as a proximity switch, the switching distance is very great is measured stably over a wide temperature range. The device according to the invention is characterized in that measuring means are provided to measure the field loss resistance R _F and the pure measuring coil reactance X _L directly. This device according to the invention is based on the knowledge that this field loss resistance R _F and the pure coil reactance X _{L are} dependent on the distance s of the control flag, while the winding resistance R 1 is independent of the distance s of the control flag.

The field loss resistance R _F and the pure reactance X _{L of} the sensor coil cannot be measured directly from the measuring coil voltage U and from the measuring coil current I, since the measuring coil voltage U is measured too large by the voltage drop across the winding resistance R 1. The invention is therefore based on the further finding that the voltage drop across the parallel connection of R _F and X _L , the so-called induced voltage U _ind , is to be measured. The induced measuring coil voltage U _ind (now referred to as U _{1, ind} = U ₁) can only be measured by means of an auxiliary coil L₂ as U _{2, ind} = U ₂, which is attached directly to the measuring coil L₁ and is directly magnetically coupled to it, so that the coupling factor k is as close as possible to one. Corresponding to the known transmission of alternating voltages by means of a transformer or transformer, the induced voltage of the primary side U _{1, ind} = U ₁ to the induced voltage of the secondary side U _{2, ind} = U ₂ is like the number of windings of the primary side N 1 to that of the secondary side N ₂ .

The secondary voltage U ₂ is therefore directly proportional to the inductive measuring coil primary voltage U ₁, which cannot be measured directly, the winding ratio N 1 / N 2 being arbitrary, but must be known. The winding resistance R₂ of the secondary auxiliary winding L₂ does not go into a voltage measurement if this measurement is carried out with high resistance, ie, the measuring current is very small, or the voltage U ₂ is measured without power; in this case the induced voltage U ₂ is equal to the voltage at the coil connections. The advantage of the powerless measurement of the induced voltage is that the temperature-dependent winding resistance R₂ no longer influences the measurement.

In a preferred embodiment, the secondary voltage U ₂ is made equal to the primary voltage U ₁ by making the number of turns on the primary side N₁ and N₂ equal on the secondary side. This case can be realized in a simple manner by winding both windings together in a bifilar manner, so that the coupling losses are low and the pure inductance of the coil L 1 on the primary side becomes almost the same as the inductance of the coil L 2 on the secondary side.

The measurement of the measuring coil current I which, for example, through the coil L 1 on the Primary side flows, preferably takes place by means of a voltage measurement in the above described way with an ohmic resistor as a current-voltage converter.

In a further embodiment of the invention, the measuring coil forms the inductance an LC series resonant circuit or LC parallel resonant circuit, which is in one known oscillator excited at its resonance frequency f₀ to natural vibrations becomes. The advantage of this excitation of the measuring coil of an inductive proximity sensor is the minimum external energy requirement of an oscillator that operates in its resonance will, d. that is, the power consumption of an oscillator resonates when excited Minimum.

In a further embodiment of the invention, an inductive approximation switch or proximity initiator a mathematical circuit-technical link made with digital electronic means that a binary output signal of Switch or initiator is generated in such a way that the state of the binary Output signal changes only when the control flag has a fixed switching distance falls below or exceeds, depending on the design of the switch. The binary output signal of the inductive proximity switch should indicate whether there is a Control flag further away from or compared to a specified switching distance is closer to the active sensor surface of the measuring coil of the proximity switch.  

The devices according to the invention will be explained in more detail with reference to drawings will:

Fig. 1a shows a lossy coil with the impedance Z ;

Fig. 1b shows the equivalent circuit diagram of a lossy coil with the inductance L, with the field loss resistor R _F due to the damping and losses of the directional electromagnetic high-frequency field and with the wire resistance R₁;

Fig. 2a shows a lossy coil with two magnetically coupled windings;

FIG. 2b shows the equivalent circuit of a lossy coil with two magnetically coupled windings;

Fig. 3 shows the basic method for measuring the phase ϕ _UI regardless of the wire resistance R₁;

Fig. 4 shows the dependence of U _L , I _L and quantities derived therefrom on the distance s of a control vane;

Fig. 5a shows an embodiment of an inductive proximity sensor with a simple measuring coil and two voltage;

FIG. 5b shows the pulse diagram of the inductive proximity sensor according to FIG. 5a;

Fig. 6 shows an extended embodiment of an inductive proximity sensor with a voltage-coupled auxiliary coil according to Figure 2 and excited by means of an oscillator.

FIG. 7 shows an embodiment of an inductive proximity switch for measuring the phase ϕ _UI according to FIG. 3 with a phase shifter and a phase comparator;

FIG. 8 shows an extended exemplary embodiment according to FIG. 7 with a Schmitt trigger output.

An analysis of a lossy coil with a winding according to FIG. 1 shows that the phase shift ϕ _{UI is} determined by the wire resistance R₁ of the lossy coil in the materials and frequencies in question for an inductive proximity switch.

In practice, it can be assumed that the winding or wire resistance R₁ is significantly smaller than the field loss resistance R _F , so that the relationship for the phase shift ϕ _{UI is} simplified.

The use of a coil with two magnetically coupled windings according to Fig. 2 allows access to the phase ϕ _UI practically without - as shown above - influence of the wire resistors R₁ and R₂. With the coil, which is connected as the first winding with the inductance L 1 between the terminals A and B, a second winding is wound with the inductor L 2 bifilar, which is led out between the terminals A 'and B'. As a result, both windings have the same inductance L₁ = L₂ = L, the combined effect of the mutual inductance

is lifted and the connections A 'and B' allow one of the wire twists stands R₁ and R₂ practically uninfluenced measurement. As winding wire can for both windings together stranded or for both windings separately more solid Enamelled copper wire can be used. The connections A and A 'and B and B 'denote those connections that have the same voltage polarity when applied have an alternating voltage or on which the windings in the same Are connected.

In principle, different windings can also be used, then they are Secondary side parameters considering the turns ratio and the To convert the coupling factor to the primary side. But this bill has none Influence on the phase relationship to be evaluated.

To measure the phase ϕ _UI , an alternating current source with the current I is connected to the terminals A and B of the coil according to FIG. 3. In the second winding, the voltage U ₂ is coupled, which can be measured directly at the terminals A 'and B' in a powerless, high-resistance voltage measurement with I ₂ = 0. If the current I is unknown, the voltage U _i = R₁ · I (at I ₂ = 0) present between the terminals A and A 'and proportional to the coil current I can be evaluated. If both voltages U and U _{i are} measured without power, or if the current I is evaluated directly and the voltage U is measured without power, the following results for the phase:

Thus advantageously neither the amplitude of the excitation current or voltage, nor the temperature-dependent wire resistances in the measurement.

In Fig. 5, an inductive proximity sensor with a simple measuring coil of the com plex impedance Z of FIG. 1 is shown. The measurement of the zero crossing of the voltage drop U on the sensor coil or the measurement of the time of exceeding and falling below the first comparison voltage U _{V, U} by the alternating voltage U is carried out with the voltage comparator K 1. At the output of K₁ there is a pulse train with the pulse repetition frequency f of the AC voltage U , the voltage for the positive half-wave of the AC voltage U or if the comparison voltage U _{V, U is} exceeded _, "logic one" and for the negative half-wave or if the voltage falls below Has comparison voltage U _{V, U} "logic zero" or vice versa.

The measurement of the zero crossing of the coil current I as a voltage drop U _I at the reference resistor Z _ref or the measurement of the time when the second reference voltage U _{V, I is} exceeded and undershot by the AC voltage U _I by means of the second voltage comparator K 2 also results in a pulse train with Pulse repetition frequency f, which is however phase-shifted by the phase shift ϕ _UI ∞. With the help of a logical AND gate UG, which logically links the output signal of K₁ and the inverted output signal of K₂, you get a pulse sequence F with the pulse repetition frequency f or the period T = 1 / f and the pulse width tip according to the equation 6 is directly proportional to the phase shift ϕ _UI ∞. With further digital modules, not shown here, such as a clock generator, counter, gate and flip-flops, the phase angle ϕ _UI ∞ or the pulse width T _{UI can} be measured digitally by means of known measuring arrangements.

According to FIG. 4, the phase φ ∞ _UI is in the absence of the control vane or very large distance s → ∞ almost exactly 90 degrees. With decreasing distance s, the phase ϕ _UI also decreases. In the application of a proximity switch according to the invention, the phase ϕ _{UI is} compared with a reference value ,₁, below which a switching process is triggered. If a stable switching point is to be achieved despite a possible drifting frequency f of the excitation, the reference phase ϕ 1 must be generated independently of frequency, e.g. B. with an all-pass. However, it can be assumed that the frequency is stable, so that the reference phase ϕ₁ with a more simple arrangement, e.g. B. a low-pass filter or a delay element. For example, it is possible to generate the effect of the phase angle ϕ₁ by means of the reference impedance Z _ref by a phase shift ϕ _I between the measuring coil current I and the AC voltage U _I.

The sizes R _F and L are both different depending on the distance s and material of the control flag, as well as on the frequency f of the excitation. If the frequency f is selected correctly, a greatly reduced material dependency can be achieved. There is also the possibility of using the measuring coil as a frequency-determining part of the excitation, e.g. B. together with a capacitor C in the parallel resonant circuit of an oscillator so as to reduce the energy requirement. Using the oscillation condition of such a parallel resonant circuit, the following then follows for the phase:

With

The most significant advantage of the method according to the invention is that which can be achieved thereby high switching speed. At each zero crossing, the phase difference can be be true, e.g. B. at a frequency f = 50 kHz can be about every 10 microseconds Switching process are triggered. In contrast, conventional inductive  Proximity sensors, which evaluate the amplitude of a parallel resonant circuit, are the first the energy stored in the resonant circuit is reduced. The time arises constant:

If you take z. For example, if the numerical values of L = 160 µH and R = 0.3 Ω of a coil typically used in inductive proximity sensors, the time constant τ _{LC is} approximately 270 µs. The time constant solely by the energy in the resonant circuit is thus many times greater than the achievable switching time of the proximity switch according to the invention.

Due to the high speed and the purely digital output signal, in increased interference suppression by evaluating several in a very simple manner Zero crossings take place.

In Fig. 6, an inductive proximity sensor is shown with a measuring coil L₁, which has an auxiliary coil L₂ for decoupling the induced measuring coil voltage to be measured U ₁ = U ₂. A capacitor C is connected together with the measuring coil L 1 and a resistor R to the frequency-determining parallel resonant circuit of the oscillator OSZ. The resistor R is used for current-voltage conversion, since the wire resistor R 1 that can also be used on the primary side (not shown here in the drawing) often has a value that is too low for reliable evaluation.

A reference phase ϕ₁ can be generated with an analog all-pass Ph₁ as a phase shifter. The signs of the resulting voltage signals are converted into digital signals with the two analog comparators K 1 and K 2. The two output signals of the comparators K₁ and K₂ are - as described above in the embodiment according to FIG. 5 - evaluated by means of a logical AND gate UG and thus a pulse sequence with the frequency f = 1 / T is generated.

An embodiment of a proximity switch according to the invention is shown in FIG. 7. A capacitor C is connected together with the measuring coil L 1 and a resistor R to the frequency-determining parallel resonant circuit of the oscillator OSZ. The resistor R is used for current-voltage conversion, since the wire resistor R 1 that can also be used on the primary side (not shown here in the drawing) often has a value that is too low for reliable evaluation.

The reference phase ϕ₁ is generated with an analog all-pass Ph₁ as a phase shifter. The signs of the resulting voltage signals are analog with the two Comparators K₁ and K₂ implemented in digital signals. The comparator K₁ controls the Clock input of the edge-controlled D memory element or D flip-flop D₁. This is used as a digital phase comparator; it saves that at the time the clock edge existing state, i.e. the sign of the current, to the next Clock edge. This state is led to the output of the switching signal S. The The function of a D flip-flop is, for example, in E. Schrüfer, Electrical Measurement Technology, 5th  Edition shown on page 330.

For ease of use, it is desirable that the proximity switch not switch back and forth continuously when the control lug is at the sensing range. A simple improvement shows the embodiment of an inductive proximity switch with switching hysteresis according to Fig. 8. The additional devices of a second phase shifter Ph₂, a D flip-flop D₂ and an RS flip-flop RS cause the behavior of a Schmitt trigger. The phase shifter Ph₂ delays the sign signal of the current by a small hysteresis phase ϕ₂. The resulting signal is performed on the flip-flop D₂, the function of which is identical to the flip-flop D₁. The state-controlled flip-flop RS causes only states that are outside the hysteresis range determined by Ph₂ to cause a change in the switching signal S.

Instead of the gate UG, either a NAND gate or an OR gate or a NOR gate can be used without changing the function and the arrangement of the circuit according to FIGS. 5 and 6.

If either an exclusive-OR gate or antivalence gate or an equivalence gate is used instead of the gate UG in FIGS. 5 and 6, a pulse sequence F with twice the frequency 2 f or with half the period T / 2 = 1 is produced / 2f of the measuring coil alternating current L, in which case the outputs of the two comparators K 1 and K 2 are connected directly to the two inputs of these gates.

Without a change of functions can in the circuits of FIGS. 5 to 8, the two outputs of the comparators K₁ and K₂ are interchanged.

Likewise, the phase shifter Ph₁ can also be inserted between the reference resistor Z _ref and the input of the comparator K₂ without the basic behavior of the circuits according to FIGS. 5 to 8 changing.

Furthermore, in Fig. 8 the output of comparator K₂ directly with the D inputs of the two flip-flops D₁ and D₂ and at the same time the output of comparator K₁ with the clock input of D flip-flop D₂ and the output of the phase shifter Ph₂ with the clock input of the D -Flipflops D₁ be connected without changing anything in the basic behavior of the circuit.

The rela in the circuits of Figs. 7 and 8 specified memory members hung as flip-flops represent only one embodiment of this invention. The D flip-flop, for example, a clocked RS flip-flop in which the input signal on the set input and the inverted input signal the reset input is routed. Alternatively, the use of transparent or state-controlled D flip-flops is possible. Other versions of digital flip-flops, flip-flops or memory elements can also be used if the functions of the circuits according to the invention presented above are achieved.

Another embodiment of the proximity switch according to the invention is Speed measurement of high-speed shafts and disks as speed sensors. For this purpose, the movement of a cam mounted on a single-running shaft or the slot provided there is scanned by the fixed speed sensor.

Claims (28)

1. Inductive proximity sensor for measuring the distance of an electrically conductive or permeable measurement object as a control flag, wherein a sensor coil of the proximity sensor generates an electromagnetic high-frequency field by the sensor coil being fed with an alternating current I or with an alternating voltage U of frequency f, and this Field is influenced by the measurement object, so that the impedance Z of the coil is changed and so that this change in Z serves as a measure of the distance of the control lug and is measured by electronic means ,

• - That electronic means are available to generate a pulse train, the frequency f is equal to the frequency or twice the frequency of the alternating current I or the alternating voltage U and the pulse width T _UI is proportional to the phase shift ϕ _UI , which is between the coil AC voltage U and the coil AC current I is measured directly during each half-cycle or a multiple of each half-cycle of the current and the voltage,
• - That a current / voltage converter transforms the alternating current I into an AC voltage U ₁ to I proportio and
• - That the pulse width T _{UI is} generated by comparing the AC voltage U and U _I with a comparison DC voltage U _{V, U} and U _{V, I} by the time start of the pulse width T _UI when the AC voltage U or U _I exceeds or falls below the reference DC voltage U _{V, U} or U _{V, I} and the time end when falling below or exceeding U _I or U compared to U _{V, I} or U _{V, U is} determined.

2. Inductive proximity sensor according to claim 1, characterized in

• that the alternating voltage U at the sensor coil of the proximity sensor at the impedance Z is generated by the measuring coil alternating current I as a voltage drop,
• that the AC voltage U _I at the reference AC resistance with the impedance Z _{ref is} generated by the AC current I as a voltage drop,
• - That a first analog comparator K 1 generates a signal change of its output signal K 1 from "logic zero" to "logic one" or vice versa when the measuring coil AC voltage U is exceeded or undershot compared to the comparison voltage U _{V, U of} a first DC voltage source,
• - That a second analog comparator K₂ when the reference AC voltage U _I exceeds or falls below the reference voltage U _{V, I} a second DC voltage source generates a signal change in its output signal K₂ from "logic zero" to "logic one" or vice versa and
• - That the output signal of K₁ and the inverted output signal of K₂ or the inverted output signal of K₁ and the non-inverted output signal of K₂ with an AND gate UG a pulse train F with the frequency f = 1 / T of the measuring coil alternating current I and the pulse width T. _UI is, the pulse width being directly proportional to the phase shift ϕ _UI between the coil current I and the coil voltage U.

3. Inductive proximity sensor according to claim 2, characterized in that the Gate UG is a NAND gate. 4. Inductive proximity sensor according to claim 2, characterized in that the Gate UG is an OR gate. 5. Inductive proximity sensor according to claim 2, characterized in that the Gate UG is a NOR gate. 6. Inductive proximity sensor according to claim 2, characterized in that the Gate UG is an exclusive OR gate or antivalence gate, by the outputs of the two comparators K₁ and K₂ directly with the two Inputs of the gate UG are connected so that a pulse train with the double frequency 2f of the measuring coil current arises. 7. Inductive proximity sensor according to claim 2, characterized in that the Gate UG is an equivalence gate by the outputs of the two com parators K₁ and K₂ verbun directly to the two inputs of the gate UG be so that a pulse train with twice the frequency 2f of the measurement coil current arises. 8. Inductive proximity sensor according to claim 2 or 3 or 4 or 5 or 6 or 7, characterized in that the reference AC resistance Z _{ref is} an ohmic resistance. 9. Inductive proximity sensor according to claim 2 or 3 or 4 or 5 or 6 or 7, characterized in that the reference AC resistance Z _{ref is} a low-loss capacitor with the capacitance C _ref , so that the phase shift ϕ _UI by the additional phase shift ϕ _{I is} reduced by approximately 90 ° (degrees). 10. Inductive proximity sensor according to claim 2 or 3 or 4 or 5 or 6 or 7, characterized in that the reference AC resistance Z _{ref is} a low-loss coil with the inductance L _ref , so that the phase shift ϕ _UI by the additional phase shift ϕ _I from about 90 degrees. 11. Inductive proximity sensor according to claim 2 or 3 or 4 or 5 or 6 or 7, characterized; that the reference AC resistance Z _ref consists of a combination of an ohmic resistor, a low-loss capacitor or a low-loss coil, so that the phase shift ϕ _{UI is changed} by the additional phase shift ϕ _I between the measuring coil alternating current I and the alternating voltage U. 12. Inductive proximity sensor according to claim 2 or 3 or 4 or 5 or 6 or 7, characterized in that the comparative DC voltage U _{V, U is} zero, so that the comparator K₁ at a zero crossing of the measuring coil AC voltage U changes its output signal. 13. Inductive proximity sensor according to claim 2 or 3 or 4 or 5 or 6 or 7, characterized in that the comparison direct voltage U _{V, I is} zero, so that the comparator K₂ at a zero crossing of the reference AC voltage U _{I changes} its output signal. 14. Inductive proximity sensor according to claim 2 or 3 or 4 or 5 or 6 or 7, characterized in that an oscillatory resonant circuit is formed from the measuring coil with the impedance Z and from the reference AC resistance Z _ref by the series circuit of Z. and Z _{ref is} supplemented by an AC resistor Z _p connected in parallel. 15. Inductive proximity sensor according to claim 14 and either 8 or 9 or 10, characterized in that the AC resistor Z _{p connected in} parallel is a capacitor with the capacitance C. 16. Inductive proximity sensor according to claim 2 or 3 or 4 or 5 or 6 or 7, characterized in that an oscillatory resonant resonant circuit is formed from the measuring coil with the impedance Z and from the reference AC resistance Z _ref by the series connection of Z. and Z _{ref is} supplemented by an AC resistor Z _r connected in series. 17. Inductive proximity sensor according to claim 16 and either 8 or 9 or 10, characterized in that the series-connected AC resistor Z _{r is} a capacitor with the capacitance C. 18. Inductive proximity sensor according to claim 14 or 15 or 16 or 17, characterized characterized in that the resonant circuit and an electronic amplifier form the components of an oscillatory oscillator by the Reso resonance circuit is the feedback of the oscillator and thus the resonance frequency f of the oscillator. 19. Inductive proximity sensor according to claim 18, characterized in that the Oscillator has a sinusoidal output signal.   20. Inductive proximity sensor according to claim 2 or 3 or 4 or 5 or 6 or 7, characterized in that

• - That the measuring coil consists of two magnetically coupled windings, the first winding with the connections A and B, with the number of turns N₁ and with the inductance L₁ by means of the measuring coil alternating current I generates an electromagnetic high-frequency field, and
• - That the induced in the first winding AC voltage U ₁, which is the voltage drop across the inner parallel circuit of inductor L₁ and field loss resistance R _F due to the current I , by means of the second winding with the connections A 'and B', with the number of turns N₂ and measured with the inductance L₂ as AC voltage U ₂, so that the winding resistance R₁ of the first winding does not enter the AC voltage U ₂.

21. Inductive proximity sensor according to claim 20, characterized in that the AC voltage U ₂ is measured at the second winding with high resistance, ie the measuring current I ₂ is very small and thus almost zero, so that the AC voltage U ₂ is almost independent of the winding resistance R₂ of the second winding. 22. Inductive proximity sensor according to claim 20 or 21, characterized in that the two windings of the measuring coil are wound bifilarly, so that the number of turns N₁ is equal to the number of turns N₂, the two windings have the same inductance L₁ equal to L₂ and thus the voltage U ₂ on the second winding is almost the same size as the induced voltage U ₁ of the first winding. 23. Inductive proximity sensor according to claims 1 and 21 or 22, characterized in that the measuring coil alternating current I is measured as a voltage drop U _i at the winding resistance R₁ of the first winding by the voltage voltage U _i between the connections of the two windings with the same span voltage polarity, for example A and A ', is measured and the other two connections of the two windings, for example B and B' are at the same voltage potential. 24. Inductive proximity sensor according to claim 2 or 3 or 4 or 5 or 6 or 7, characterized in that the measuring coil AC voltage U , which is led from the measuring coil to the input of the comparator K₁, shifted by a phase shifter Ph₁ by a reference phase ϕ₁ becomes. 25. Inductive proximity sensor according to claim 20 or 21 or 22 or 23, characterized in that the induced alternating voltage U ₂ or U '₂, which is guided by the second winding with the connections A' and B 'to the input of the comparator K₁, by means of a phase shifter Ph₁ is shifted by a reference phase ϕ₁. 26. Inductive proximity switch or proximity initiator according to claim 2 to 25, characterized in

• - That the output of the comparator on the clock input of the flankenge controlled D memory element or D flip-flop D₁ is performed,
• - That the output of the comparator K₂ on the state input, as D input of the D memory element D₁ is performed, so that the D memory element has the function of a digital phase comparator and the output of the D memory element carries the switching signal S, and
• - That the switching point or the change in the state of the switching signal S by means of the reference phase phase₁ of the phase shifter Ph₁ or by means of the phase of Z _{ref is} adjustable.

27. Inductive proximity switch or proximity switch with switching hysteresis according to claim 26, characterized in

• - That either the reset input or the set input of a state-controlled RS flip-flop or RS flip-flop RS is connected to the output of the D flip-flop D 1,
• - That either the set input or the reset input of the state-controlled RS flip-flop RS is connected to the output of a second edge-controlled D memory element or D flip-flop D₂,
• - That the two clock inputs of the two D flip-flops D₁ and D₂ are connected to the same output of the comparator K₁,
• - That the state input or D input of the D flip-flop D₁ is connected directly to the output of the comparator K₂ and
• - That the state input or D input of the D flip-flop D₂ is connected via a second phase shifter Ph₂ to the output of the comparator K₂ ver, the input signal of the D input of the D flip-flop D₂ being shifted by the hysteresis phase ϕ₂, so that for the output signal or the switching signal S of the RS flip-flop RS gives the behavior of a Schmitt trigger, ie the output signal S only changes when the change in the phase shift ϕ _UI between the two output signals from K₁ and K₂ is greater than ϕ₂ .

28. Inductive proximity switch according to claim 2 to 27, characterized in that the inductive proximity sensor or proximity switch as Tachometer is used.