CN209389940U - Constant Current Control Circuit for LLC Resonant Converter - Google Patents

Abstract

This application discloses a kind of constant-current control circuits for LLC resonant converter.The LLC resonant converter includes the first bipolar junction transistor and the second bipolar junction transistor to be worked using self-oscillation mode, the constant-current control circuit includes: switch element, for being shorted the driving current of at least one first bipolar junction transistor and second bipolar junction transistor；And drive module, including exporting current calculation module, for calculate the difference of resonance current signal and the first transformer magnetizing current signal absolute value average value as thermal compensation signal, and the on state of the switch element according to compensation signal control, to realize current constant control.In the complicated circuit for constituting charge pump PFC module and LLC resonant transformation combined application, which still can be improved constant-current control accuracy.

Description

Constant Current Control Circuit for LLC Resonant Converter

technical field

The utility model relates to the technical field of power supplies, in particular to a constant current control circuit for an LLC resonant converter.

Background technique

The LED driving circuit is used to provide a direct current output current to the LED lamp, so that the LED lamp is lit to be used as an illumination source. Key performance parameters of LED driver circuits include power factor (PF) and output current ripple. The power factor represents the ratio of active power to reactive power. The output current ripple characterizes the AC component of the DC output current. For example, the AC component is a power frequency component, which will cause stroboscopic flickering of the LED lamp, which not only affects the lighting effect, but also affects the life of the LED lamp. The high power factor of the LED drive circuit can improve the utilization rate of electric energy, and the low output current ripple can reduce flicker.

In order to take into account high power factor and low output current ripple, the LED drive circuit can adopt a variety of cascaded circuit schemes, including: single-stage flyback primary side control constant current system architecture and the first type of ripple elimination circuit cascaded solution; the second type cascaded scheme composed of boost topology and flyback primary-side control constant current topology; the third type cascaded scheme composed of boost topology and resonant half-bridge LLC structure; charge pump PFC module and resonant half-bridge The fourth type of cascading scheme composed of bridge LLC structure.

All of the above four types of circuit solutions can achieve high power factor (PF) and low output current ripple (no flicker) at the same time. However, the disadvantage of the first type of cascade solution is that the ripple elimination circuit has a great influence on the system efficiency, especially when the resonant output voltage is relatively low. The disadvantage of the second type of cascading solution is that the system of the two-level solution is more complicated, the system cost is higher, and EMI debugging is more difficult and the efficiency is not high. The efficiency of the third type cascading scheme and the fourth type cascading scheme is higher than that of the second type cascading scheme, but the system is more complicated and the cost is higher.

In the fourth type of cascade solution, the resonant switching converter is a power converter that uses a switching tube to obtain a square wave voltage and uses a resonant circuit for resonance to realize energy transmission. The LLC resonant converter has high power density and fewer electronic components, and at the same time has a smooth current waveform, which is beneficial to improve electromagnetic interference, and can realize zero voltage switching (Zero Voltage Switching) of the switching tube in the entire operating range. , ZVS) and zero current switching (Zero Current Switching, ZCS), help to obtain extremely high efficiency. Further, adding a combination of a current-mode charge pump passive PFC and a voltage-mode charge pump passive PFC on the LLC half-bridge drive can obtain a high power factor (PF) and a very low total harmonic distortion (THD). Therefore, the fourth type of cascade scheme has obvious advantages in terms of circuit efficiency.

Further, it is expected that both improvement of circuit efficiency and reduction of circuit cost can be achieved in the fourth type of cascading scheme.

Utility model content

In view of the above problems, the present application provides a constant current control circuit for an LLC resonant converter, wherein the compensation signal obtained by the constant current control circuit includes a resonant current signal and a first transformer excitation current signal, thereby improving the accuracy of constant current control.

According to one aspect of the present invention, a constant current control circuit for an LLC resonant converter is provided, the LLC resonant converter includes a first transformer, a first bipolar transistor and a second bipolar transistor, the The first bipolar transistor and the second bipolar transistor work in a self-oscillating mode, so that the resonance current and the excitation current flow through the primary winding of the first transformer, and the constant current control circuit includes: a switching element , for short-circuiting the driving current of at least one of the first bipolar transistor and the second bipolar transistor; and a driving module, including an output current calculation module, used for calculating the resonant current signal and the excitation of the first transformer The average value of the absolute value of the current signal difference is used as a compensation signal, and the conduction state of the switching element is controlled according to the compensation signal to realize the control of the resonant frequency, so as to realize the constant current control.

Preferably, the LLC resonant converter further includes a second transformer, the second transformer has a load winding, and a first drive winding and a second drive winding coupled to the load winding, the load winding of the second transformer connected to the resonant circuit to obtain a resonant current, the same-named end of the first driving winding and the different-named end of the second driving winding are respectively connected to the first bipolar transistor and the second bipolar transistor base, thereby providing a corresponding driving current generated according to the induction current of the resonant current.

Preferably, the switching element connects the same-named end and the different-named end of the first driving winding to each other when short-circuiting the driving current.

Preferably, the second transformer further includes a control winding, and the switching element connects the same-named end and the different-named end of the control winding to each other when the driving current is short-circuited.

Preferably, the switching element includes: a first transistor and a second transistor, the first transistor is connected between the opposite terminal of the first drive winding and the ground terminal, and the second transistor is connected to the second transistor. Between the same terminal of a driving winding and a ground terminal connected to an intermediate node of the first bipolar transistor and the second bipolar transistor.

Preferably, it also includes: a first operational amplifier and a second operational amplifier, respectively connected to the control terminals of the first transistor and the second transistor, wherein the drive module provides a turn-on signal, and the first operational amplifier and the The second operational amplifier provides an off signal, and the switch control signals of the first transistor and the second transistor are superposition signals of the on signal and the off signal.

Preferably, the respective non-inverting input terminals of the first operational amplifier and the second operational amplifier receive a negative potential reference voltage, and the inverting input terminals are connected to the respective output terminals to realize the same-named terminal of the corresponding winding of the second transformer and a negative voltage clamp on the opposite side.

Preferably, the switch element includes: a first transistor and a second transistor, connected in reverse series between the same-name terminal and the ground terminal of the corresponding winding of the second transformer, and the different-name terminal of the corresponding winding of the second transformer terminal and the ground terminal are connected to an intermediate node of the first bipolar transistor and the second bipolar transistor.

Preferably, the driving module is connected to the control terminals of the first transistor and the second transistor to provide switch control signals.

Preferably, the first transformer includes a primary winding and a secondary winding, the primary winding is used as a part of a resonant tank, and the secondary winding is coupled with the primary winding to provide a resonant output voltage, wherein the The output current calculation module obtains the compensation signal according to the current sampling signal of the resonance current and the voltage feedback signal of the resonance output voltage.

Preferably, the output current calculation module includes: a third operational amplifier and a third transistor connected to the output terminal for generating the first current; a fourth operational amplifier and a fourth transistor connected to the output terminal for generating the first current two currents; a plurality of current mirrors for subtracting the first current and the second current to generate an equivalent charging current; and a capacitor for integrating the equivalent charging current to generate the compensation signal, Wherein, the non-inverting input terminals of the third operational amplifier and the fourth operational amplifier receive the first reference voltage and the second reference voltage respectively, and the inverting input of one of the third operational amplifier and the fourth operational amplifier One terminal receives the current sampling signal, and the other inverting input terminal is grounded.

Preferably, the output current calculation module further includes: a first switch for selectively grounding the inverting input terminal of the third operational amplifier or receiving the current sampling signal; a second switch for connecting the The inverting input terminal of the fourth operational amplifier is selectively grounded or receives the current sampling signal; the comparator compares the voltage feedback signal with a third reference voltage, thereby generating the first switch and the second Two switch control signals.

Preferably, the second reference voltage is greater than the first reference voltage.

Preferably, the driving module further includes: an oscillator, generating a clock signal according to the ramp signal and the compensation signal; and a logic module, generating an opening signal or a switch control signal according to the clock signal.

According to another aspect of the present utility model, a constant current control method for an LLC resonant converter is provided, the LLC resonant converter includes a first transformer, a first bipolar transistor and a second bipolar transistor, the The first bipolar transistor and the second bipolar transistor work in a self-excited oscillation mode, so that the resonance current and the excitation current flow through the primary winding of the first transformer, and the constant current control method includes: calculating The average value of the absolute value of the difference between the resonant current signal and the first transformer excitation current signal is obtained to obtain a compensation signal; the conduction state of the switching element is controlled according to the compensation signal so as to realize the control of the resonant frequency to realize constant current control, wherein , short-circuiting the driving current of at least one of the first bipolar transistor and the second bipolar transistor when the switch element is turned on.

Preferably, the step of obtaining the compensation signal includes: comparing the current sampling signal of the resonant current signal with the first reference voltage to generate the first current; using the second reference voltage to generate the second current; combining the first current with the obtained and subtracting the second current to generate an equivalent charging current; and integrating the equivalent charging current to generate the compensation signal, wherein the second reference voltage is greater than the first reference voltage.

Preferably, the method further includes switching the path of the current sampling signal according to the resonant output voltage signal, so as to obtain an average value of absolute values of differences between the resonant current signal and the first transformer excitation current signal.

Preferably, the resonant output voltage signal is compared with a third reference voltage to obtain a path switching signal of the circuit sampling signal.

Preferably, the method further includes performing negative voltage clamping on the control terminals of the first bipolar transistor and the second bipolar transistor.

According to the constant current control circuit of the embodiment of the utility model, the driving current of the first bipolar transistor and the second bipolar transistor is short-circuited by using a switching element, so that the first bipolar transistor and the second bipolar transistor The switching period of the type transistor follows the switch control signal, thereby realizing the control of the resonant frequency. The compensation signal obtained by the constant current control circuit represents the average value of the absolute value of the difference between the resonant current and the excitation current of the first transformer, and the frequency of the switch control signal is controlled according to the negative feedback of the average value, so that the primary side of the first transformer can The constant current control of the output current on the secondary side of the first transformer is realized. In the complex circuit formed by combining the charge pump PFC module and the LLC resonant converter, the constant current control circuit can still improve the constant current control precision.

In a preferred embodiment, the control circuit in the constant current control circuit can directly control the drive current of the base of the first bipolar transistor, and use the coupling between the first drive winding and the second drive winding to indirectly control the second The drive current for the base of a bipolar transistor. The constant current control circuit does not need to provide an additional control circuit for the second bipolar transistor, so that the circuit structure of the control circuit can be further simplified and the circuit cost can be reduced.

In a preferred embodiment, the control circuit includes a first transistor and a second transistor connected between the opposite end and the same end of the first drive winding, for short-circuiting the first drive winding to control the first bipolar transistor base drive current. The middle node of the first bipolar transistor and the second bipolar transistor serves as the ground terminal (floating ground) of the control circuit. The first transistor and the second transistor are used as switching elements for short-circuiting the first driving winding, so there is no need for actual grounding. The control circuit does not need to use a power supply circuit to generate the control current required by the control winding, so the power consumption of the circuit and the circuit cost can be reduced.

Description of drawings

Through the following description of the embodiments of the utility model with reference to the accompanying drawings, the above-mentioned and other purposes, features and advantages of the utility model will be more clear, in the accompanying drawings:

FIG. 1 shows a schematic circuit diagram of a power supply device according to the prior art.

Fig. 2 shows a schematic circuit diagram of the LED driving circuit according to the first embodiment of the present invention.

Fig. 3 shows a schematic circuit diagram of an LED driving circuit according to a second embodiment of the present invention.

FIG. 4 shows a schematic circuit diagram of the control circuit in the LED driving circuit shown in FIG. 3 .

FIG. 5 shows a working waveform diagram of the LED driving circuit shown in FIG. 3 .

6a to 6c show the equivalent circuit diagrams of the LED driving circuit shown in FIG. 3 in the first stage.

7a to 7b show the equivalent circuit diagrams of the LED driving circuit shown in FIG. 3 in the second stage.

8a to 8c show the equivalent circuit diagrams of the LED driving circuit shown in FIG. 3 in the third stage.

9a to 9b show the equivalent circuit diagrams of the LED driving circuit shown in FIG. 3 in the fourth stage.

Fig. 10 shows a schematic circuit diagram of the control circuit in the LED driving circuit according to the third embodiment of the present invention.

FIG. 11 shows the working waveform diagram of the control circuit shown in FIG. 10 .

FIG. 12 shows a detailed circuit block diagram of the control circuit shown in FIG. 4 .

FIG. 13 shows a schematic circuit diagram of an output current calculation module in the control circuit shown in FIG. 12 .

FIG. 14 shows a schematic diagram of the principle of current regulation by the control circuit shown in FIG. 12 .

Detailed ways

Various embodiments of the present invention will be described in more detail below with reference to the accompanying drawings. In the various drawings, the same elements are denoted by the same or similar reference numerals. For the sake of clarity, various parts in the drawings have not been drawn to scale.

FIG. 1 shows a schematic circuit diagram of a power supply device according to the prior art. The power supply device 100 includes a rectifier bridge DB, a filter capacitor Ce, a charge pump PFC module 110 , and a resonant converter 120 . The rectifier bridge DB is used to convert the AC input voltage AC into a rectified input voltage. The charge pump PFC module 110 adopts the resonant output voltage and resonant current obtained from the resonant tank, and superimposes them on the input end of the LLC resonant converter 120 to implement power factor correction. The filter capacitor Ce converts the rectified input voltage into a smooth DC input voltage. The resonant converter 120 converts the DC input voltage into a resonant output voltage to supply power to the load LD.

The charge pump PFC module 110 includes diodes DX1 and DX2, diodes Di1 and Di2, boost capacitors Ci1 and Ci2. The current source charge pump module includes a boost capacitor Ci2 and a diode Di2, and uses the resonant current generated by the resonant circuit composed of the resonant inductor Lr and the resonant capacitor Cr as the current source. The voltage source charge pump module includes a boost capacitor Ci1 and a diode Di1, and uses the terminal voltage of the resonant capacitor Cr as a voltage source.

The resonant converter 120 includes a control circuit 121, switching elements M1 and M2, a coupling capacitor Cc, a resonant inductor Lr, and a resonant capacitor Cr. The control circuit 121 controls the conduction state of the switching elements M1 and M2 to generate a square wave voltage. The square wave voltage is input into the resonant tank to generate resonance. The terminal voltage of the resonant capacitor Cr supplies power to the load.

In the power supply device 100, the current source charge pump uses the high-frequency current loop generated by the on and off of the switch element to obtain the electric energy of the AC input voltage, and the voltage source charge pump uses the high-frequency current loop generated by the switch element on and off. The node voltage gains power from the AC input voltage, thereby boosting the voltage and current of the DC input voltage. Switching elements M1 and M2 generate high-frequency voltage and current by switching on and off states. Since the resonant circuit composed of the resonant inductor Lr and the resonant capacitor Cr acts as the load of the switching elements M1 and M2, the high frequency output current is the resonant current at the resonant frequency.

In the LED driving circuit, the switching elements M1 and M2 used in the resonant converter are metal oxide semiconductor field effect transistors (MOSFETs) as switching tubes. Although the MOSFET has excellent switching performance, a complex control circuit is required to provide a control signal for the switch tube. Therefore, the use of the MOSFET as the switch tube leads to an increase in the cost of the LED driving circuit.

Fig. 2 shows a schematic circuit diagram of the LED driving circuit according to the first embodiment of the present invention. The power supply device 200 includes a rectifier bridge DB, a filter capacitor Cht, a charge pump PFC module 210 , and an LLC resonant converter 220 .

The rectifier bridge DB is used to convert the AC input voltage AC into a rectified input voltage.

The charge pump PFC module 210 includes a diode D1 and a boost capacitor Cboost. The diode D1 is connected between the positive output terminal of the rectifier bridge DB and the positive input terminal of the LLC resonant converter 220 to form a unidirectional conductive path from the rectifier bridge DB to the LLC resonant converter 220 . The cathode of the diode D1 is connected to the first end of the resonant capacitor Cr in the LLC resonant converter 220 . The boost capacitor Cboost is connected between the positive output terminal and the negative output terminal of the rectifier bridge DB. The charge pump PFC module 210 uses the resonant current obtained from the resonant tank to extract current from the rectified input voltage to implement power factor correction, and provide current to the filter capacitor Cht to implement the boost function.

The filter capacitor Cht is connected between the output terminal of the charge pump PFC module 210 and the negative output terminal of the rectifier bridge DB. The filter capacitor Cht converts the rectified input voltage into a smooth DC input voltage.

The LLC resonant converter 220 includes a first transformer T1, a second transformer T2, bipolar transistors Q1 and Q2, diodes D2 and D3, a capacitor Cmid, a resonant capacitor Cr and a resonant inductor Lr. Diodes D2 and D3 are connected in antiparallel to bipolar transistors Q1 and Q2, respectively, and capacitor Cmid is connected in parallel to bipolar transistor Q2.

On the primary side of the first transformer T1, the primary winding Lp of the first transformer T1, the resonant capacitor Cr and the resonant inductor Lr form a resonant circuit. Between the positive and negative input terminals of the LLC resonant converter 220, bipolar transistors Q1 and Q2 are connected in series, and the intermediate node of the two is connected to the resonant tank. In the resonant circuit, the sampling resistor Rs is connected in series with the primary winding Lp, so that a sampling signal used to characterize the inductor current flowing through the primary winding Lp can be obtained. The second transformer T2 comprises four windings around the same core, namely the load winding W1, the drive windings W2 and W3, the control winding W4. In the resonant tank, the load winding W1 is connected in series with the primary winding Lp. Meanwhile, drive windings W2 and W3 are coupled to the bases of bipolar transistors Q1 and Q2 respectively, but in opposite directions. That is, the dot end of the drive winding W2 is connected to the base of the bipolar transistor Q1 , and the dot end of the drive winding W3 is connected to the base of the bipolar transistor Q2 . These windings are used to provide the necessary current to drive the bases of bipolar transistors Q1 and Q2 to realize self-oscillating drive (SOC, Self-Oscillating Converter). Under the control of the self-oscillating driving signal, the bipolar transistors Q1 and Q2 are alternately turned on and off to convert the DC input voltage into a square wave voltage. The square wave voltage is input into the resonant tank to generate a resonant current at the resonant frequency. Therefore, through the resonant circuit, electric energy is transmitted from the primary side of the first transformer T1 to the secondary side of the first transformer T1.

On the secondary side of the first transformer T1, diodes D4 and D5 form a rectification circuit. Both ends of the secondary winding are respectively connected to the anodes of diodes D4 and D5, and the middle tap of the secondary winding is grounded. The output capacitor C1 is connected between the cathodes of the diodes D4 and D5 and ground, providing a DC resonant output voltage across them.

The LLC resonant converter 220 also includes a constant current control circuit 221 . The constant current control circuit 221 obtains the current sampling signal CS of the resonant current from the sampling resistor Rs of the resonant converter 220 , and obtains the voltage feedback signal FB of the resonant output voltage from the auxiliary winding Lf of the first transformer T1 of the resonant converter 220 . The constant current control circuit 221 includes driving terminals DR1 and DR2 respectively connected to the different terminal and the same terminal of the control winding W4 of the second transformer T2. The constant current control circuit 221 controls the connection relationship between the driving terminals DR1 and DR2, thereby controlling the resonant frequency, thereby controlling the resonant current.

During operation, the LLC resonant converter 220 converts the DC input voltage into a resonant output voltage to power the LED load. The switching commutation of the bipolar transistors Q1 and Q2 in the LLC resonant converter 220 is naturally occurring at the inherent SOC oscillation frequency. However, when the LLC resonant converter 220 is working, its switching frequency needs to be adjusted, which is generally higher than the inherent SOC oscillation frequency.

The LED drive circuit according to this embodiment adopts the cascaded solution of the charge pump PFC module and the LLC resonant converter to realize AC-DC voltage conversion and supply power to the LED load, thereby obtaining a very high power factor (PF) and a very low total voltage. Harmonic Distortion (THD). In the LLC resonant converter, the bipolar transistor is used as the switch tube, and the self-excited oscillation is used to control the on and off state of the switch tube, and to control the short circuit of the control winding and release the short circuit state at an appropriate time to control the switch tube alternately. conduction, thereby controlling the resonant frequency, thereby realizing constant current control, simplifying the control circuit and reducing circuit cost.

Fig. 3 shows a schematic circuit diagram of an LED driving circuit according to a second embodiment of the present invention. The power supply device 300 includes a rectifier bridge DB, a filter capacitor Cht, a charge pump PFC module 310 , and an LLC resonant converter 320 .

The rectifier bridge DB is used to convert the AC input voltage AC into a rectified input voltage.

The charge pump PFC module 310 includes a diode D1 and a boost capacitor Cboost. The diode D1 is connected between the positive output terminal of the rectifier bridge DB and the positive input terminal of the LLC resonant converter 320 to form a unidirectional conductive path from the rectifier bridge DB to the LLC resonant converter 320 . The anode of the diode D1 is connected to the first end of the resonant capacitor Cr in the LLC resonant converter 320 . The boost capacitor Cboost is connected between the positive output terminal and the negative output terminal of the rectifier bridge DB. The charge pump PFC module 310 uses the resonant current obtained from the resonant tank to extract current from the rectified input voltage to implement power factor correction, and provide current to the filter capacitor Cht to implement the boost function.

The filter capacitor Cht is connected between the output terminal of the charge pump PFC module 310 and the negative output terminal of the rectifier bridge DB. The filter capacitor Cht converts the rectified input voltage into a smooth DC input voltage.

The LLC resonant converter 320 includes a first transformer T1, a second transformer T2, bipolar transistors Q1 and Q2, diodes D2 and D3, a capacitor Cmid, a resonant capacitor Cr and a resonant inductor Lr. Diodes D2 and D3 are connected in antiparallel to bipolar transistors Q1 and Q2, respectively, and capacitor Cmid is connected in parallel to bipolar transistor Q2.

On the primary side of the first transformer T1, the primary winding Lp of the first transformer T1, the resonant capacitor Cr and the resonant inductor Lr form a resonant circuit. Between the positive and negative input terminals of the LLC resonant converter 320, bipolar transistors Q1 and Q2 are connected in series, and the intermediate node of the two is connected to the resonant tank. In the resonant circuit, the sampling resistor Rs is connected in series with the primary winding Lp, so that a sampling signal used to characterize the inductor current flowing through the primary winding Lp can be obtained. The second transformer T2 comprises three windings around the same core, namely the load winding W1, the drive windings W2 and W3. In the resonant tank, the load winding W1 is connected in series with the primary winding Lp. Meanwhile, drive windings W2 and W3 are coupled to the bases of bipolar transistors Q1 and Q2 respectively, but in opposite directions. That is, the dot end of the drive winding W2 is connected to the base of the bipolar transistor Q1, and the dot end of the drive winding W3 is connected to the base of the bipolar transistor Q2. These windings are used to provide necessary current to drive the bases of the bipolar transistors Q1 and Q2 to realize self-oscillating drive (SOC, Self-Oscillating Converter). Under the control of the self-oscillating driving signal, the bipolar transistors Q1 and Q2 are alternately turned on and off to convert the DC input voltage into a square wave voltage. The square wave voltage is input into the resonant tank to generate a resonant current at the resonant frequency. Therefore, through the resonant circuit, electric energy is transmitted from the primary side of the first transformer T1 to the secondary side of the first transformer T1.

On the secondary side of the first transformer T1, diodes D4 and D5 form a rectification circuit. Both ends of the secondary winding are respectively connected to the anodes of diodes D4 and D5, and the middle tap of the secondary winding is grounded. The output capacitor C1 is connected between the cathodes of the diodes D4 and D5 and ground, providing a DC resonant output voltage across them.

The LLC resonant converter 320 also includes a constant current control circuit 321 . The constant current control circuit 221 obtains the current sampling signal CS of the resonant current from the sampling resistor Rs of the resonant converter 220 , and obtains the voltage feedback signal FB of the resonant output voltage from the auxiliary winding Lf of the first transformer T1 of the resonant converter 220 . The constant current control circuit 321 includes driving terminals DR1 and DR2 respectively connected to the different terminal and the same terminal of the driving winding W2 of the second transformer T2, and a ground terminal GND connected to the middle node of the bipolar transistors Q1 and Q2. The constant current control circuit 321 controls the connection relationship between the driving terminals DR1 and DR2 and the ground terminal GND, thereby controlling the resonant frequency, thereby controlling the resonant current.

During operation, the LLC resonant converter 320 converts the DC input voltage into a resonant output voltage to power the LED load. The switching commutation of bipolar transistors Q1 and Q2 in LLC resonant converter 320 is naturally occurring at the inherent SOC oscillation frequency. However, when the LLC resonant converter 320 is working, its switching frequency needs to be adjusted, which is generally higher than the inherent SOC oscillation frequency.

The LED drive circuit according to this embodiment adopts the cascaded solution of the charge pump PFC module and the LLC resonant converter to realize AC-DC voltage conversion and supply power to the LED load, thereby obtaining a very high power factor (PF) and a very low total voltage. Harmonic Distortion (THD). In the LLC resonant converter, a bipolar transistor is used as a switch tube, and self-excited oscillation is used to control the on and off states of the switch tube, and to control the short circuit of at least one drive winding and release the short circuit state at an appropriate time to control the switch The tubes are turned on alternately to control the resonant frequency, thereby realizing constant current control, simplifying the control circuit and reducing circuit cost.

FIG. 4 shows a schematic circuit diagram of the control circuit in the LED driving circuit shown in FIG. 3 .

The constant current control circuit 321 includes transistors M1 and M2 , operational amplifiers U1 and U2 , and a driving module 3211 . In this embodiment, transistors M1 and M2 are, for example, MOSFETs. Further, the first terminal and the second terminal of the transistor M1 are respectively connected between the driving terminal DR1 and the ground terminal GND, and the first terminal and the second terminal of the transistor M2 are respectively connected between the driving terminal DR2 and the ground terminal GND.

The driving module 3211 obtains the current sampling signal CS of the resonant current from the sampling resistor Rs of the resonant converter 320, obtains the voltage feedback signal FB of the resonant output voltage from the auxiliary winding Lf of the first transformer T1 of the resonant converter 320, and according to the current sampling The signal CS and the voltage feedback signal FB generate the compensation signal Vcomp. The driving module 3211 is connected to the control terminals of the transistors M1 and M2, and is used to provide the turn-on signals VG1 and VG2 to the transistors M1 and M2 respectively, and the operational amplifiers U1 and U2 provide the turn-off signals to the transistors M1 and M2. Therefore, the transistors M1 and M2 The switch control signal is the superposition signal of the turn-on signal and the turn-off signal. The non-inverting input terminal of the operational amplifier U1 receives a negative potential reference voltage -Vref, preferably -0.1V, and the inverting input terminal is connected to the output terminal. Further, the output terminal of the operational amplifier U1 is connected to the control terminal of the transistor M1. In addition to being able to control M1 to be turned off, the amplifier U1 controls M1 to be in an amplified state when negative voltage appears at the DR1 terminal to ensure that the voltage at the DR1 terminal is not lower than -0.1V. The noninverting input terminal of the operational amplifier U2 receives a negative potential reference voltage, preferably -0.1V, and the inverting input terminal is connected to the output terminal. Further, the output terminal of the operational amplifier U2 is connected to the control terminal of the transistor M2. The operational amplifier U2 In addition to being able to control M2 to turn off, when a negative voltage appears at the DR2 terminal, the operational amplifier U2 controls M2 to be in an amplified state to ensure that the voltage at the DR2 terminal is not lower than -0.1V.

FIG. 5 shows a working waveform diagram of the LED driving circuit shown in FIG. 3 . The figure shows the relationship between the resonance current sampling signal CS, the voltage feedback signal FB, the clock signal CLK obtained by the driving module 3211 and the exciting current CT1 of the first transformer T1 and the exciting current CT2 of the second transformer T2 over time.

The resonant current sampling signal CS intersects the excitation current CT2 of the second transformer T2 at points A, B, and C. The clock signal CLK has two states of high and low (1, 0) levels, and the resonant current sampling signal CS also has two states of positive and negative (>0, <0) levels. There are four different combinations of two states. state, resulting in different circuit stages.

In the low-level state of the clock signal CLK, the driving module 3211 in the constant current control circuit 321 generates the first turn-on signal VG1, so that the transistor M1 is turned on, and the transistor M2 is controlled by the operational amplifier U2, and there are two states: one is The shutdown state; the second is the negative voltage clamping state. In the high-level state of the clock signal CLK, the driving module 3211 in the constant current control circuit 321 generates the second turn-on signal VG2 to make the transistor M2 turn on, and the transistor M1 is controlled by the operational amplifier U1, and there will be two states: one is The shutdown state; the second is the negative voltage clamping state.

During the low-level time period of the clock signal CLK, the current sampling signal CS of the resonant current is converted from a negative current to a positive current. During the high level period of the clock signal CLK, the current sampling signal CS of the resonant current is converted from positive current to negative current.

Therefore, the first stage of the LED driving circuit corresponds to the time period t0 to t1 in the figure, the second stage corresponds to the time period t1 to t2 in the figure, the third stage corresponds to the time period t2 to t3 in the figure, and the fourth The phases correspond to the time period t3 to t4 in the figure.

6a to 6c show the equivalent circuit diagrams of the LED driving circuit shown in FIG. 3 in the first stage. As shown in the figure, the clock signal CLK is at low level, and the resonant current is the first stage of the circuit when the current is negative. In the first stage, the transistor M1 in the constant current control circuit 321 is turned on, and the negative voltage of the transistor M2 is clamped. The current path of the LED driving circuit 300 changes due to the change of the difference between the resonant current sampling signal CS and the excitation current CT2 of the second transformer T2, and the change of the charging state of the capacitor Cmid.

The rectifier bridge DB includes four diodes D11 to D14 forming a bridge circuit to provide a rectified input voltage between the positive output terminal and the negative output terminal of the rectifier bridge DB.

At time t0, bipolar transistors Q1 and Q2 are both off. The AC input voltage charges the capacitor Cmid via the resonant tank. During charging of the capacitor Cmid, the terminal voltage Vmid of the capacitor Cmid gradually rises. The resonant current flows reversely through the primary winding Lp of the first transformer T1 and the load winding W1 of the second transformer T2, that is, flows from the opposite end to the same end in the corresponding winding, and can be obtained according to the resonant current CS and the second transformer T2 The difference value of the exciting current CT2 judges that the currents of the driving windings W2 and W3 flow from the end of the same name to the end of the same name. Since the voltage difference between the two ends of the driving winding W2 is only 0.1V, it can be judged that there is no current flowing inside the driving winding W3 .

As shown in Figure 6a, the resonant current path in the LED drive circuit 300 is: from the positive output terminal of the rectifier bridge DB, through the resonant capacitor Cr, the resonant inductor Lr, the primary winding Lp of the first transformer T1, and the second transformer T2 The load winding W1, the sampling resistor Rs, and the capacitor Cmid return to the negative output terminal of the rectifier bridge DB. In addition, since the transistor M2 in the constant current control circuit 321 clamps the negative voltage, its function is similar to the voltage source connected between the driving terminal DR2 of the constant current control circuit 321 and the ground terminal GND, therefore, the driving of the bipolar transistor Q1 The current path is: from the drive terminal DR2 of the constant current control circuit 321, through the drive winding W2 of the second transformer T2 and the drive terminal DR1 of the constant current control circuit 321, and return to the ground terminal GND of the constant current control circuit 321, forming a current loop. The drive current path of bipolar transistor Q2 is disconnected.

Then, when the voltage Vmid is greater than the terminal voltage of the filter capacitor Cht, the resonant current path changes. At this time, the base-collector junction of the bipolar transistor Q1 freewheels, thereby operating in reverse conduction. The resonant current no longer charges the capacitor Cmid, but charges the filter capacitor Cht via the bipolar transistor Q1.

As shown in Figure 6b, the resonant current path in the LED drive circuit 300 is: from the positive output terminal of the rectifier bridge DB, through the resonant capacitor Cr, the resonant inductor Lr, the primary winding Lp of the first transformer T1, and the second transformer T2 The load winding W1, the sampling resistor Rs, the bipolar transistor Q1, and the filter capacitor Cht return to the negative output terminal of the rectifier bridge DB. In addition, the driving current path of the bipolar transistor Q1 remains unchanged, and the driving current path of the bipolar transistor Q2 is disconnected.

Then, after point A in Figure 5, when the difference between the resonant current CS and the excitation current of the second transformer T2 changes from negative to positive, the current flow direction of the drive windings W2 and W3 of the second transformer T2 also changes, from The internal different terminal flows to the same terminal. At this time, the transistor M2 inside the constant current control circuit 321 changes from the negative voltage clamping state to the off state, and a part of the resonant current flows reversely through the drive winding W2 of the second transformer T2. , that is, flows from the opposite end to the same end in the corresponding winding, and flows through the base-collector junction of the bipolar transistor Q1, so that the bipolar transistor Q1 is completely reversed. Another part of the resonant current charges the filter capacitor Cht via the bipolar transistor Q1.

As shown in Figure 6c, the resonant current path in the LED drive circuit 300 is: from the positive output terminal of the rectifier bridge DB, through the resonant capacitor Cr, the resonant inductor Lr, the primary winding Lp of the first transformer T1, and the second transformer T2 The load winding W1, the sampling resistor Rs, the bipolar transistor Q1, and the filter capacitor Cht return to the negative output terminal of the rectifier bridge DB. In addition, the driving current path of the bipolar transistor Q1 passes through the driving winding W2 of the second transformer T2 and the base-collector junction of the bipolar transistor Q1, and the driving current path of the bipolar transistor Q2 is disconnected.

At time t1, the negative current phase of the resonance current ends, and the first phase of the LED driving circuit 300 ends.

7a to 7b show the equivalent circuit diagrams of the LED driving circuit shown in FIG. 3 in the second stage. As shown in the figure, the clock signal CLK is at low level, and the resonant current is the second stage of the circuit when the current is positive. In the second stage, the transistor M1 in the constant current control circuit 321 is turned on and the transistor M2 is turned off. The current path of the LED driving circuit 300 changes due to the difference between the resonant current sampling signal CS and the excitation current CT2 of the second transformer T2.

At time t1, the resonant current CS is forward, and the drive winding W2 of the second transformer T2 obtains a reverse drive current, so that the bipolar transistor Q1 is turned on. The bipolar transistor Q2 remains in an off state. The resonant current charges the boost capacitor Cboost via diode D1. During charging of the capacitor Cboost, the terminal voltage Vboost of the boost capacitor Cboost gradually increases. The resonant current flows forward through the primary winding Lp of the first transformer T1 and the load winding W1 of the second transformer T2, that is, flows from the end of the same name to the end of the same name in the interior of the corresponding winding, which can be obtained according to the resonant current CS and the second transformer T2 The difference of the excitation current CT2 judges that the currents of the drive windings W2 and W3 flow from the end with the same name to the end with the same name.

As shown in Figure 7a, the resonant current path in the LED driving circuit 300 is: from the first end of the resonant inductor Lr, through the resonant capacitor Cr, the boost capacitor Cboost, the filter capacitor Cht, the bipolar transistor Q1, and the sampling resistor Rs , the load winding W1 of the second transformer T2, the primary winding Lp of the first transformer T1, and return to the second end of the resonant inductor Lr. In addition, since the driving winding W2 of the second transformer T2 obtains the reverse driving current, the driving current path of the bipolar transistor Q1 is: from the terminal with the same name of the driving winding W2 of the second transformer T2, via the bipolar transistor Q1 The base-emitter junction of the second transformer T2 is returned to the opposite end of the drive winding W2. The drive current path of bipolar transistor Q2 is disconnected.

Then, when the terminal voltage Vboost of the boost capacitor Cboost is higher than the terminal voltage Vht of the smoothing capacitor Cht, the resonant current path changes. At this time, the diode D1 is turned on. The resonant current no longer charges the boost capacitor Cboost, but flows to the collector of the bipolar transistor Q1 via the diode D1.

As shown in Figure 7b, the resonant current path in the LED driving circuit 300 is: from the first end of the resonant inductor Lr, through the resonant capacitor Cr, the diode D1, the bipolar transistor Q1, the sampling resistor Rs, and the second transformer T2 The load winding W1, the primary winding Lp of the first transformer T1, returns to the second terminal of the resonant inductor Lr. In addition, since the driving winding W2 of the second transformer T2 obtains the reverse driving current, the driving current path of the bipolar transistor Q1 remains unchanged. The drive current path of bipolar transistor Q2 is disconnected.

At time t2, the low-level phase of the clock signal CLK ends, and the second phase of the LED driving circuit 300 ends.

8a to 8c show the equivalent circuit diagrams of the LED driving circuit shown in FIG. 3 in the third stage. As shown in the figure, the clock signal CLK is at a high level, and the resonant current CS is a positive current, which is the third stage of the circuit. In the third stage, the negative voltage of the transistor M1 in the constant current control circuit 321 is clamped, and the transistor M2 is turned on. The turn-on of the transistor M2 makes the base-emitter of the bipolar transistor Q1 short-circuited, therefore, the bipolar transistor Q1 is always in a cut-off state. The current path of the LED driving circuit 300 changes due to the change of the difference between the current sampling signal CS and the excitation current CT2 of the second transformer T2, and the change of the charging state of the capacitor Cmid.

At time t2, the clock signal CLK is switched from low level to high level, and the bipolar transistors Q1 and Q2 are both off. The capacitor Cmid is discharged via the resonant tank. During the discharge of the capacitor Cmid, the terminal voltage Vmid of the capacitor Cmid gradually decreases. The resonant current flows forward through the primary winding Lp of the first transformer T1 and the load winding W1 of the second transformer T2, that is, flows from the end of the same name to the end of the same name in the interior of the corresponding winding, which can be obtained according to the resonant current CS and the second transformer T2 The difference value of the exciting current CT2 judges that the currents of the driving windings W2 and W3 flow from the end of the same name to the end of the same name. Since the voltage difference between the two ends of W2 is only 0.1V, it can be judged that there is no current flowing inside W3.

As shown in Figure 8a, the resonant current path in the LED driving circuit 300 is: from the first end of the resonant inductor Lr, through the resonant capacitor Cr, diode D1, filter capacitor Cht, capacitor Cmid, sampling resistor Rs, and the second transformer T2 The load winding W1 of the first transformer T1 and the primary winding Lp of the first transformer T1 are returned to the second terminal of the resonant inductor Lr. In addition, since the transistor M1 in the constant current control circuit 321 clamps the negative voltage, the function is similar to the voltage source connected between the driving terminal DR1 and the ground terminal GND of the constant current control circuit 321, therefore, the driving of the bipolar transistor Q1 The current path is: from the driving terminal DR1 of the constant current control circuit 321, through the driving winding W2 of the second transformer T2 and the driving terminal DR2 of the constant current control circuit 321, back to the ground terminal GND of the constant current control circuit 321, forming a current loop. The drive current path of bipolar transistor Q2 is disconnected.

Then, when the voltage Vmid is lower than the voltage of the ground terminal GND of the constant current control circuit 321 , the driving current path of the bipolar transistor Q2 changes. At this time, the base-collector junction of the bipolar transistor Q2 freewheels, thereby operating in reverse conduction. The capacitor Cmid is no longer discharged via the resonant tank.

As shown in Fig. 8b, a part of the resonant current flows forwardly through the driving winding W3 of the second transformer T2, that is, flows from the same-named end to the opposite-named end inside the corresponding winding, and flows through the base set of the bipolar transistor Q2 The electrode junction makes the bipolar transistor Q2 fully reversed conduction. Another part of the resonant current flows to the resonant tank via the bipolar transistor Q2. In addition, the driving current path of the bipolar transistor Q1 remains unchanged.

Then, after point B in Fig. 5, when the difference between the resonant current and the excitation current of the second transformer T2 changes from positive to negative, the current flow direction of the driving windings W2 and W3 of the second transformer T2 also changes, at this time , flows from the same-named end to the different-named end, and the transistor M1 inside the constant current control circuit 321 changes from the negative voltage clamping state to the off state.

As shown in Figure 8c, the resonant current path in the LED driving circuit 300 is: from the first end of the resonant inductor Lr, through the resonant capacitor Cr, diode D1, filter capacitor Cht, transistor Q2, sampling resistor Rs, and the second transformer T2 The load winding W1 of the first transformer T1 and the primary winding Lp of the first transformer T1 are returned to the second terminal of the resonant inductor Lr. In addition, the driving current path of the bipolar transistor Q1 is disconnected, and the driving current path of the bipolar transistor Q2 passes through the base-collector junction of the bipolar transistor Q2.

At time t3, the positive current phase of the resonance current ends, and the third phase of the LED driving circuit 300 ends.

9a to 9b show the equivalent circuit diagrams of the LED driving circuit shown in FIG. 3 in the fourth stage. As shown in the figure, the clock signal CLK is high level, and the resonant current is the fourth stage of the circuit when the current is negative. In the fourth stage, the transistor M1 in the constant current control circuit 321 is turned off, and the transistor M2 is turned on. The turn-on of the transistor M2 makes the base-emitter of the bipolar transistor Q1 short-circuited, therefore, the bipolar transistor Q1 is always in a cut-off state. The current path of the LED driving circuit 300 changes due to the difference between the current sampling signal CS and the excitation current CT2 of the second transformer.

At time t3, the resonant current is reversed, and the driving winding W3 of the second transformer T2 obtains a positive driving current, so that the bipolar transistor Q2 is turned on. The bipolar transistor Q1 is kept in an off state. The boost capacitor Cboost is discharged through the resonant tank, and the terminal voltage Vboost of the boost capacitor Cboost gradually decreases. The resonant current flows reversely through the primary winding Lp of the first transformer T1 and the load winding W1 of the second transformer T2, that is, flows from the opposite end to the same end in the corresponding winding, and can be obtained according to the resonant current CS and the second transformer T2 The difference value of the exciting current CT2 judges that the currents of the drive windings W2 and W3 flow from the end with the same name to the end with the same name.

As shown in Figure 9a, the resonant current path in the LED driving circuit 300 is: from the first end of the boost capacitor Cboost, through the resonant capacitor Cr, the resonant inductor Lr, the primary winding Lp of the first transformer T1, the second transformer The load winding W1 of T2, the sampling resistor Rs, and the bipolar transistor Q2 return to the second end of the boost capacitor Cboost. In addition, because the driving winding W3 of the second transformer T2 obtains the forward driving current, the driving current path of the bipolar transistor Q2 passes through the base-emitter junction of the bipolar transistor Q2, and the bipolar transistor Q2 conducts forward, and the bipolar transistor Q2 The driving current path of the polar transistor Q1 is disconnected.

Then, when the terminal voltage Vboost of the boost capacitor Cboost is lower than the AC input voltage, the resonant current path changes. The AC input voltage supplies power to the resonant tank.

As shown in Figure 9b, the resonant current path in the LED driving circuit 300 is: from the positive output terminal of the rectifier bridge DB, through the resonant capacitor Cr, the resonant inductor Lr, the primary winding Lp of the first transformer T1, and the second transformer T2 The load winding W1, the sampling resistor Rs, and the bipolar transistor Q2 return to the negative output terminal of the rectifier bridge DB. In addition, because the driving winding W3 of the second transformer T obtains the forward driving current, the driving current path of the bipolar transistor Q2 passes through the base-emitter junction of the bipolar transistor Q2, and the bipolar transistor Q2 conducts forward, and the bipolar transistor Q2 The driving current path of the polar transistor Q1 is disconnected.

At time t4, the high-level phase of the clock signal CLK ends, and the fourth phase of the LED driving circuit 300 ends.

Fig. 10 shows a schematic circuit diagram of the constant current control circuit in the LED driving circuit according to the third embodiment of the present invention. The difference between the LED driving circuit according to the third embodiment of the present invention and the second embodiment lies in the circuit structure of the constant current control circuit, and other aspects are the same as the second embodiment, and the differences between the two are mainly described below.

The constant current control circuit 421 includes transistors M1 and M2 and a driving module 4211 . In this embodiment, transistors M1 and M2 are, for example, MOSFETs. Further, the driving terminal DR1 of the constant current control circuit 421 is directly connected to the ground terminal GND, and the transistors M1 and M2 are connected in reverse series to form a top switch, which is connected between the driving terminal DR2 of the constant current control circuit 421 and the ground terminal GND. between. That is, the first terminal of the transistor M1 is connected to the driving terminal DR2 of the constant current control circuit 421 , the first terminal of the transistor M2 is connected to the ground terminal GND of the constant current control circuit 421 , and the second terminals of the transistors M1 and M2 are connected to each other.

The driving module 4211 obtains the current sampling signal CS of the resonant current from the sampling resistor Rs of the resonant converter 220, obtains the voltage feedback signal FB of the resonant output voltage from the additional winding Lf of the first transformer T1 of the resonant converter 220, and according to the current sampling Signal CS and voltage feedback signal FB generate switching control signals for transistors M1 and M2. The driving module 4211 is connected to the control terminals of the transistors M1 and M2, and is used to provide the same switch control signal VG to the transistors M1 and M2.

According to the LED drive circuit of this embodiment, one drive terminal in the control circuit is short-circuited to the ground terminal, and the transistors M1 and M2 are connected in reverse series between the other drive terminal and the ground terminal as a top switch, so that the control circuit can be omitted The negative voltage clamping module (eg, operational amplifier) in the circuit, thus simplifying the circuit structure and reducing the circuit cost.

FIG. 11 shows the working waveform diagram of the control circuit shown in FIG. 10 . The figure shows the relationship between the current sampling signal CS obtained by the driving module 4211, the voltage feedback signal FB, the clock signal CLK and the excitation current CT1 of the first transformer T1 and the excitation current CT2 of the second transformer T2 over time.

The current sampling signal CS intersects the excitation current CT2 of the second transformer T2 at points A, B, and C. The clock signal CLK has two state levels of high and low (1, 0), and the current sampling signal CS also has two state voltages of positive and negative (>0, <0), and there are four different states in two combinations. , resulting in different circuit stages.

On the rising or falling edge of the clock signal CLK, the drive module 4211 in the constant current control circuit 421 generates a switch control signal VG to turn on the transistors M1 and M2, and connect the drive terminal DR2 of the constant current control circuit 421 to the ground terminal GND. between short. On the rising edge or falling edge of the voltage feedback signal FB, the drive module 4211 in the constant current control circuit 421 generates a switch control signal VG, so that the transistors M1 and M2 are turned off, and the drive terminal DR2 of the constant current control circuit 421 is connected to the ground terminal GND. disconnected intermittently.

Therefore, the first stage of the LED driving circuit corresponds to the time period t0 to t1 in the figure, the second stage corresponds to the time period t1 to t2 in the figure, the third stage corresponds to the time period t2 to t3 in the figure, and the fourth The phases correspond to the time period t3 to t4 in the figure.

FIG. 12 shows a detailed circuit block diagram of the constant current control circuit 321 shown in FIG. 4 . The constant current control circuit 321 is, for example, a single packaged chip. Referring to FIG. 4 , the constant current control circuit 321 includes transistors M1 and M2 , operational amplifiers U1 and U2 , and a driving module 3211 .

The driving module 3211 obtains the current sampling signal CS of the resonant current from the sampling resistor Rs of the resonant converter 320, obtains the voltage feedback signal FB of the DC output voltage from the additional winding Lf of the first transformer T1 of the resonant converter 320, and sends the signal to the transistor M1 and M2 provide turn-on signals VG1 and VG2, respectively.

Further, as shown in FIG. 12 , the driving module 3211 of the constant current control circuit 321 includes an output current calculation module 11, a peak current limiting protection module 12, an oscillator 13, a logic module 14, a driving stage 15, a capacitor C12, and a constant current source I11.

The output current calculation module 11 generates a compensation signal Vcomp according to the voltage feedback signal FB and the resonant current sampling signal CS.

The constant current source I11 and the capacitor C12 are connected in series between the power supply terminal and the ground, and a ramp signal is generated at the middle node of the two. The two input terminals of the oscillator 13 respectively receive the ramp signal and the compensation signal Vcomp, and generate the clock signal CLK according to the two. The logic module 14 generates enable signals VG1 and VG2 according to the clock signal CLK.

In the constant current control circuit 321, the frequency of the switch control signal is related to the average value of the absolute value of the difference between the resonant current and the excitation current CT1 of the first transformer T1, that is, the frequency of the switch control signal is controlled according to the negative feedback of the average value , so that the constant output current control of the secondary side of the first transformer T1 can be realized on the primary side of the first transformer T1.

Preferably, the constant current control circuit 321 may also include multiple protection modules, including the clamp module 16 at the voltage feedback terminal, the open circuit protection module 17 , the short circuit protection module 18 , the clamp module 19 and the undervoltage lockout module 22 at the power supply terminal. In addition, the constant current control circuit 321 may further include an overvoltage protection module 20 and an overtemperature protection module 21 .

FIG. 13 shows a schematic circuit diagram of the output current calculation module in the constant current control circuit 321 shown in FIG. 12 , and FIG. 14 shows a schematic diagram of the principle of output current calculation of the control circuit shown in FIG. 12 .

As shown in FIG. 13 , the output current calculation module 11 includes operational amplifiers AMP1 and AMP2 , a comparator COMP1 , switches K11 and K12 , transistors M11 to M16 , resistors R11 and R12 , and a capacitor C11 .

The non-inverting input terminal of the operational amplifier AMP1 receives the reference voltage VREF1 , the inverting input terminal selectively receives the current sampling signal CS or ground, and the output terminal is connected to the gate of the transistor M11 . Further, the source of the transistor M11 is connected to the inverting input terminal of the operational amplifier AMP1. Transistors M13 and M14 form a first current mirror, transistors M17 and M18 form a second current mirror, and the first current mirror and the second current mirror are coupled to each other. The transistor M11 is connected in series with the transistor M13, so that the current flowing through the transistor M11 is coupled through a current mirror to generate a first current I11 flowing through the transistor M18.

The non-inverting input terminal of the operational amplifier AMP2 receives the reference voltage VREF2 , the inverting input terminal selectively receives the current sampling signal CS or ground, and the output terminal is connected to the gate of the transistor M12 . Further, the source of the transistor M12 is connected to the inverting input terminal of the operational amplifier AMP2. Transistors M15 and M16 form a third current mirror. The transistor M12 is connected in series with the transistor M15, so that the current flowing through the transistor M12 is coupled through a current mirror to generate a second current I12 flowing through the transistor M16.

Transistors M16 and M18 are connected in series such that the second current mirror and the third current mirror are connected in series. Further, the capacitor C11 is connected between the middle node of the transistors M16 and M18 and the ground, so as to provide the compensation signal Vcomp at both ends of the capacitor C11.

The non-inverting input terminal and the inverting input terminal of the comparator COMP1 receive the voltage feedback signal FB and the reference voltage VREF3 respectively. Switches K11 and K12 are single-pole double-throw switches, respectively. The output terminal of the comparator COMP1 is connected to the control terminals of the switches K11 and K12, so that the switches K11 and K12 switch simultaneously, thereby connecting the inverting input terminals of the operational amplifiers AMP1 and AMP2 to ground via the resistor R11 or connected via the resistor R12 in a complementary manner. To the current sampling signal CS.

For further details, in the following analysis, the magnification of the current mirror is assumed to be 1, and the above-mentioned reference voltages VREF1, VREF2 and VREF3 are set to 0.85, 0.95V and 0.2V, respectively. However, the present invention is not limited thereto, and the reference voltage VREF2 is greater than VREF1 and can be any appropriate value respectively. Further, the reference voltage signal Vcscc=VREF2-VREF1 is defined.

The first stage T1: the voltage feedback signal FB is greater than the reference voltage VREF3

In the first stage, the switch control signal generated by the comparator COMP1 switches the switches K11 and K12 to the A terminal respectively. The inverting input terminal of the operational amplifier AMP2 receives the current sampling signal CS via the resistor R12, and the inverting terminal of the operational amplifier AMP1 is grounded via the resistor R11. When the current sampling signal CS is zero, the second current I12=VREF2/R generated by the operational amplifier AMP2 at the intermediate node, where R represents the resistance values of the resistors R11 and R12. The first current I11=VREF1/R generated by the operational amplifier AMP1 at the intermediate node, where R represents the resistance values of the resistors R11 and R12, therefore, the current received by the capacitor C11 IDIFF=I12-I11=(VREF2-VREF1)/R= Vcscc/R>0, charge the capacitor C11. When the current sampling signal CS is less than zero (denoted as CS1, representing the part where the CS current is less than 0 in the first stage), the second current I12 generated by the operational amplifier AMP2 at the intermediate node = VREF2/R+|CS1|*Rs/R>0 . The first current I11=VREF1/R generated by the operational amplifier AMP1 at the middle node, wherein, R represents the resistance values of the resistors R11 and R12. Therefore, the current IDIFF=I12-I11=(VREF2-VREF1)/R+|CS1|*Rs/R=Vcscc/R+|CS1|*Rs/R>0 received by the capacitor C11 charges the capacitor C11.

When the current sampling signal CS is greater than zero (denoted as CS2, representing the part where the CS current is greater than 0 in the first stage), the second current I12=VREF2/R−CS2*Rs/R generated by the operational amplifier AMP2 at the intermediate node. I11=VREF1/R generated by the operational amplifier AMP1 at the middle node, where R represents the resistance value of the resistors R11 and R12. Therefore, the current received by capacitor C11 IDIFF=I12-I11=(VREF2-VREF1)/R-CS2*Rs/R=Vcscc/R-CS2*Rs/R When Vcscc/R>CS2*Rs/R, capacitor C11 The received current IDIFF=Vcscc/R−CS2*Rs/R>0 charges the capacitor C11. When Vcscc/R<CS2*Rs/R, the current IDIFF=Vcscc/R−CS2*Rs/R<0 received by the capacitor C11 discharges the capacitor C11. Therefore, in the first stage: the average value of the current IDIFF received by capacitor C11 is equal to:

Since the reference voltage signal is a DC voltage, the

is the area of S(c) in Figure 14

is the area of S(b) in Figure 14

According to geometrical knowledge, the area of the region S(a)=S(b)-S(c). And S(a) is the area surrounded by the resonant current CS and the first transformer excitation current CT1, therefore:

So formula (1) becomes:

Define the equivalent average charging current of capacitor C11 as Isource; the equivalent average discharging current is Isink

In the first stage, the equivalent average charging current Isource = Vcscc/R.

In the first stage, the equivalent average discharge current Isink is equal to the average value of the absolute value of the difference between the resonant current and the excitation current CT1 of the first transformer T1:

Wherein, Rs represents the resistance value of the sampling resistor Rs, R represents the resistance values of the resistors R11 and R12, CS represents the current sampling signal related to the resonance current, and CT1 represents the current sampling signal related to the excitation current.

The second stage T2: the voltage feedback signal FB is less than the reference voltage VREF3

In the second stage, the switch control signal generated by the comparator COMP1 switches the switches K11 and K12 to the B terminal respectively. The inverting input terminal of the operational amplifier AMP2 is grounded via the resistor R11 , and the inverting terminal of the operational amplifier AMP1 receives the current sampling signal CS via the resistor R12 .

When the current sampling signal CS is zero, the second current I12=VREF2/R generated by the operational amplifier AMP2 at the intermediate node, where R represents the resistance values of the resistors R11 and R12. The first current I11=VREF1/R generated by the operational amplifier AMP1 at the intermediate node, where R represents the resistance values of the resistors R11 and R12, therefore, the current IDIFF received by the capacitor C11=I12-I11=(VREF2-VREF1)/R= Vcscc/R>0, charge the capacitor C11.

When the current sampling signal CS is greater than zero (denoted as CS3, representing the portion where the CS current is greater than 0 in the second stage), the second current I12=VREF2/R generated by the operational amplifier AMP2 at the intermediate node. I11=VREF1-CS3*Rs/R generated by the operational amplifier AMP1 at the intermediate node, where R represents the resistance values of resistors R11 and R12. Therefore, the current IDIFF=I12-I11=(VREF2-VREF1)/R+|CS|*Rs/R=Vcscc/R+CS3*Rs/R>0 received by the capacitor C11 charges the capacitor C11.

When the current sampling signal CS is less than zero (denoted as CS4, representing the portion where the CS current is less than 0 in the second stage), the second current I12=VREF2/R generated by the operational amplifier AMP2 at the intermediate node. The first current I11 generated by the operational amplifier AMP1 at the intermediate node=VREF1/R+|CS4|*Rs/R>0, where R represents the resistance values of the resistors R11 and R12. Therefore, the current received by capacitor C11 IDIFF=(VREF2-VREF1)/R-|CS4|*Rs/R=Vcscc/R-|CS4|*Rs/R>0 in Vcscc/R>|CS4|*Rs/R When , the current IDIFF =Vcscc/R−|CS4|*Rs/R>0 received by the capacitor C11 charges the capacitor C11. When Vcscc/R<|CS4|*Rs/R, the current IDIFF=Vcscc/R−|CS4|*Rs/R<0 received by the capacitor C11 discharges the capacitor C11. Therefore, in the second stage: the average value of the current IDIFF received by capacitor C11 is equal to:

Since the reference voltage signal is a DC voltage, the

In the same way, you can get

So formula (3) also becomes:

In the second stage, the equivalent average charging current Isource = Vcscc/R.

In the second stage, the equivalent average discharge current Isink is also equal to the average value of the absolute value of the difference between the resonant current and the excitation current CT1 of the first transformer T1:

Among them, Rs represents the resistance value of the sampling resistor Rs, R represents the resistance values of the resistors R11 and R12, CS represents the current sampling signal related to the resonance current, and CT1 represents the current sampling signal related to the excitation current.

In the output current calculation module 11, the function of the capacitor C11 is to integrate the equivalent average charging current Isource and the equivalent average discharging current Isink in the resonant cycle, thereby generating the compensation signal Vcomp. when When it is greater than the reference voltage signal Vcscc, the average value of the equivalent average discharge current Isink is greater than the average value of the equivalent average charge current Isource, and the compensation signal Vcomp decreases, so that the frequency of the switch control signal decreases, thereby reducing

when When it is less than the reference voltage signal Vcscc, the average value of the equivalent average discharge current Isink is smaller than the average value of the equivalent average charge current Isource, and the compensation signal Vcomp increases to increase the frequency of the switch control signal, thereby increasing

when When it is equal to the reference voltage signal Vcscc, the average value of the equivalent average discharge current Isink is equal to the average value of the equivalent average charge current Isource, and the compensation signal Vcomp remains unchanged.

As shown in Figure 14, the output current calculation module 11 obtains the compensation signal according to the average value of the absolute value of the difference between the resonance current signal and the first transformer excitation current signal, and the value of the compensation signal corresponds to the resonance current CS and the first transformer excitation current signal The area surrounded by the current CT1 is S(a)=S(b)-S(c). According to the principle of the LLC half-bridge, the area of the area is proportional to the output current. Specifically, a switch control signal is generated according to the compensation signal to short-circuit the driving current of at least one of the first bipolar transistor and the second bipolar transistor, so as to realize the control of the resonance frequency. In the constant current control feedback loop, the compensation signal Vcomp remains unchanged, that is, the corresponding area remains unchanged, so that the resonant frequency maintains a constant value corresponding to the desired current output, thereby realizing constant current control.

In the above-mentioned embodiments, the LED driving circuit including the charge pump PFC module and the LLC resonant converter is described. It can be understood that based on a similar working principle, the LLC resonant converter can be used alone and still achieve the same technical effect.

In the above-mentioned embodiments, it is described that in the LLC resonant converter, by controlling the short-circuit of the drive winding of the upper bipolar transistor and releasing the short-circuit state at an appropriate time, the switch tubes are controlled to be turned on alternately, so that the bipolar The switching period of the transistor follows the period of the switching control signal inside the circuit, and further controls the frequency of the switching control signal according to the negative feedback of the resonant current, so that the resonant frequency maintains a constant value corresponding to the desired current output, thereby realizing constant current control. However, the present invention is not limited thereto. It can be understood that, based on a similar working principle, the circuit path control of the drive winding of the lower bipolar transistor of the LLC resonant converter can also achieve the same technical effect.

Embodiments according to the present invention are as described above, and these embodiments do not exhaustively describe all details, nor limit the utility model to only the specific embodiments described. Obviously many modifications and variations are possible in light of the above description. This description selects and specifically describes these embodiments in order to better explain the principle and practical application of the utility model, so that those skilled in the art can make good use of the utility model and the modification and use on the basis of the utility model . The invention is to be limited only by the claims and their full scope and equivalents.

Claims (14)

1. A constant current control circuit for an LLC resonant converter, the LLC resonant converter comprising a first transformer, a first bipolar transistor and a second bipolar transistor, the first bipolar transistor and The second bipolar transistor works in a self-excited oscillation mode, so that the resonant current and the excitation current flow through the primary winding of the first transformer, and it is characterized in that the constant current control circuit includes: a switching element for short-circuiting a driving current of at least one of the first bipolar transistor and the second bipolar transistor; and The drive module includes an output current calculation module, which is used to calculate the average value of the absolute value of the difference between the resonance current signal and the first transformer excitation current signal as the compensation signal, and control the conduction state of the switching element according to the compensation signal to realize Resonant frequency control to achieve constant current control. 2. The constant current control circuit according to claim 1, wherein the LLC resonant converter further comprises a second transformer, the second transformer has a load winding, and a first driver coupled to the load winding winding and a second drive winding, the load winding of the second transformer is connected to the resonant tank to obtain a resonant current, the same-name end of the first drive winding and the opposite-name end of the second drive winding are respectively connected to the The bases of the first bipolar transistor and the second bipolar transistor provide corresponding driving currents generated according to the induction current of the resonance current. 3 . The constant current control circuit according to claim 2 , wherein the switching element connects the same-named end and the different-named end of the first driving winding to each other when the driving current is short-circuited. 4. The constant current control circuit according to claim 2, wherein the second transformer further comprises a control winding, and when the switching element short-circuits the drive current, the same terminal and the different terminal of the control winding are connected to each other. The name ends are connected to each other. 5. The constant current control circuit according to claim 3 or 4, wherein the switching element comprises: A first transistor and a second transistor, the first transistor is connected between the opposite end and the ground end of the first driving winding, and the second transistor is connected between the same end and the ground end of the first driving winding Between, the ground terminal is connected to the middle node of the first bipolar transistor and the second bipolar transistor. 6. The constant current control circuit according to claim 5, further comprising: The first operational amplifier and the second operational amplifier are respectively connected to the control terminals of the first transistor and the second transistor, Wherein, the driving module provides a turn-on signal, the first operational amplifier and the second operational amplifier provide a turn-off signal, and the switch control signals of the first transistor and the second transistor are the turn-on signal and the turn-off signal. The superimposition signal of the broken signal. 7. The constant current control circuit according to claim 6, wherein the respective non-inverting input terminals of the first operational amplifier and the second operational amplifier receive a negative potential reference voltage, and the inverting input terminals are connected to the respective output end, so as to realize the negative voltage clamping of the same-named end and the different-named end of the corresponding winding of the second transformer. 8. The constant current control circuit according to claim 3 or 4, wherein the switching element comprises: The first transistor and the second transistor are connected in reverse series between the same-name terminal and the ground terminal of the corresponding winding of the second transformer, and the opposite-name terminal and the ground terminal of the corresponding winding of the second transformer are connected to the intermediate nodes of the first bipolar transistor and the second bipolar transistor. 9. The constant current control circuit according to claim 8, wherein the driving module is connected to the control terminals of the first transistor and the second transistor to provide a switch control signal. 10. The constant current control circuit according to claim 1, wherein the first transformer comprises a primary winding and a secondary winding, the primary winding is part of a resonant circuit, and the secondary winding is connected to the secondary winding The primary winding is coupled to provide a resonant output voltage, Wherein, the output current calculation module obtains the compensation signal according to the current sampling signal of the resonance current and the voltage feedback signal of the resonance output voltage. 11. The constant current control circuit according to claim 10, wherein the output current calculation module comprises: a third operational amplifier and a third transistor connected to the output terminal for generating a first current; A fourth operational amplifier and a fourth transistor connected to the output terminal are used to generate a second current; a plurality of current mirrors for subtracting the first current and the second current to generate an equivalent charging current; and capacitor for integrating the equivalent charging current to generate the compensation signal, Wherein, the non-inverting input terminals of the third operational amplifier and the fourth operational amplifier receive the first reference voltage and the second reference voltage respectively, and the inverting input of one of the third operational amplifier and the fourth operational amplifier One terminal receives the current sampling signal, and the other inverting input terminal is grounded. 12. The constant current control circuit according to claim 11, wherein the output current calculation module further comprises: a first switch, configured to selectively ground the inverting input terminal of the third operational amplifier or receive the current sampling signal; a second switch, configured to selectively ground the inverting input terminal of the fourth operational amplifier or receive the current sampling signal; The comparator compares the voltage feedback signal with a third reference voltage, so as to generate control signals for the first switch and the second switch. 13. The constant current control circuit according to claim 11, wherein the second reference voltage is greater than the first reference voltage. 14. The constant current control circuit according to claim 1, wherein the drive module further comprises: an oscillator for generating a clock signal according to the ramp signal and the compensation signal; and The logic module generates an opening signal or a switch control signal according to the clock signal.