CN204992594U

CN204992594U - Reactive power compensator based on novel many level of modularization topological structure

Info

Publication number: CN204992594U
Application number: CN201520492743.5U
Authority: CN
Inventors: 高晗璎; 李妍美; 魏巍; 王海瑞; 李伟力
Original assignee: Harbin University of Science and Technology
Current assignee: Harbin University of Science and Technology
Priority date: 2015-07-09
Filing date: 2015-07-09
Publication date: 2016-01-20
Anticipated expiration: 2025-07-09

Abstract

The invention relates to a reactive power compensation device based on a novel modular multilevel topology structure, which relates to a high-voltage, high-power reactive power compensation device. The purpose of the utility model is to solve the problems of limited compensation in the field of high voltage and high power and insufficient ability to suppress circulating current in the prior art. The utility model includes a three-phase AC power supply, a resistance-inductive load, an MMC converter, a signal detection circuit, a control circuit and a drive circuit. The MMC converter includes three bridge arms connected in parallel with the same structure, and each bridge arm includes The midpoint of the arm is symmetrical and connected in series with the upper bridge arm and the lower bridge arm. The upper bridge arm includes inductors connected in series, several half-bridge units and an H-bridge unit. The inductor of the upper bridge arm is connected in series with the inductor of the lower bridge arm. The midpoints of the three bridge arms of the inverter are connected in parallel between the three-phase AC power supply and the load through wires. The utility model can not only compensate the reactive power of the power grid, solve the three-phase unbalance problem of the system, but also support the voltage of the power grid and restrain the circulating current.

Description

Reactive power compensation device based on a novel modular multilevel topology

技术领域 technical field

本实用新型涉及一种无功补偿装置，具体涉及基于新型模块化多电平换流器的无功补偿装置，属于高压大功率无功补偿技术领域。 The utility model relates to a reactive power compensation device, in particular to a reactive power compensation device based on a novel modularized multilevel converter, and belongs to the technical field of high voltage and high power reactive power compensation.

背景技术 Background technique

随着电力换流器及非线性负载的大量应用，在电网中产生了大量的无功及谐波。利用STATCOM进行无功补偿和谐波抑制，不仅能够满足电网节能、降损的要求，而且还有利于电网供电质量的改善。 With the extensive application of power converters and nonlinear loads, a large number of reactive power and harmonics are generated in the power grid. The use of STATCOM for reactive power compensation and harmonic suppression can not only meet the requirements of power grid energy saving and loss reduction, but also help improve the quality of power grid power supply.

目前，常见的多电平拓扑结构主要有三种：二极管钳位型、飞跨电容型以及H桥级联型。二极管钳位型和飞跨电容型结构，随着电平数的增加，所需的开关器件和钳位电容数量会大大增加，不利于实现更高电平的变换电路，而且电容电压不易均衡，推广应用受到限制。H桥级联结构，当三相输出的电流不均衡时，桥臂间不能传递有功能量，难以实现三相模块间的电容电压平衡。 At present, there are mainly three common multilevel topologies: diode-clamped, flying-capacitor, and H-bridge cascaded. Diode clamping type and flying capacitor type structure, as the number of levels increases, the number of switching devices and clamping capacitors required will greatly increase, which is not conducive to the realization of higher level conversion circuits, and the capacitor voltage is not easy to balance, Promoted apps are limited. H-bridge cascaded structure, when the three-phase output current is unbalanced, the active energy cannot be transmitted between the bridge arms, and it is difficult to realize the capacitor voltage balance between the three-phase modules.

实用新型内容 Utility model content

本实用新型的目的是为了解决现有技术在高压大功率领域补偿受限以及对环流的抑制能力不足的问题。 The purpose of the utility model is to solve the problems of limited compensation in the field of high voltage and high power and insufficient ability to suppress circulating current in the prior art.

本实用新型的技术方案是：基于新型模块化多电平拓扑结构的无功补偿装置，包括三相交流电源、阻感负载、换流器、控制电路、信号检测电路和驱动电路，所述换流器包括三个结构相同并联连接的桥臂，每个桥臂包括关于桥臂中点对称且串联连接的上桥臂和下桥臂，所述上桥臂包括相互串联的电感、若干半桥单元和一个H桥单元，上桥臂的电感与下桥臂的电感串联连接，换流器三个桥臂的中点通过导线并联接在三相交流电源和阻感负载之间，信号检测电路的输入端分别连接三相交流电源的输出端、阻感负载的输入端、换流器的输出端、换流器三个桥臂、换流器的每个半桥单元和H桥单元，信号检测电路的输出端连接控制电路的输入端，控制电路的输出端通过驱动电路与换流器建立连接。 The technical solution of the utility model is: a reactive power compensation device based on a new modular multi-level topology, including a three-phase AC power supply, a resistive load, a converter, a control circuit, a signal detection circuit and a drive circuit. The inverter includes three bridge arms with the same structure connected in parallel, each bridge arm includes an upper bridge arm and a lower bridge arm connected in series symmetrically with respect to the midpoint of the bridge arm, and the upper bridge arm includes inductors connected in series, several half-bridges Unit and an H-bridge unit, the inductance of the upper bridge arm is connected in series with the inductance of the lower bridge arm, the midpoint of the three bridge arms of the converter is connected in parallel between the three-phase AC power supply and the resistive load through a wire, and the signal detection circuit The input terminals of the three-phase AC power supply, the input terminal of the resistive load, the output terminal of the converter, the three bridge arms of the converter, each half-bridge unit of the converter and the H-bridge unit are respectively connected. The output end of the detection circuit is connected to the input end of the control circuit, and the output end of the control circuit is connected to the converter through the drive circuit.

所述控制电路包括载波移相控制器、第一控制单元和第二控制单元，第一控制单元的输出端和第二控制单元的输出端均与载波移相控制器建立连接，所述第一控制单元包括第一比较器、第二比较器、第三比较器、第四比较器、第五比较器、第一PI控制器、第二PI控制器、第三PI控制器、第一坐标转换器、第二坐标转换器、第一电抗器和第二电抗器，第一比较器、第一PI控制器、第二比较器、第二PI控制器和第三比较器依次串联后接入第一坐标变换器，第四比较器、第三PI控制器和第五比较器依次串联后接入第一坐标转换器，第一坐标转换器的输出端连接载波移相控制器，载波移相控制器的输出端连接驱动电路，所述换流器的输出端连接第二坐标变换器，第二坐标变换器的第一输出端分别连接第二比较器和第二电抗器，第二电抗器的输出端连接第三比较器，第二坐标变换器的另一输出端分别连接第四比较器和第一电抗器，第一电抗器的输出端连接第五比较器，采用两个并行的控制器对换流器的功率模块和全桥模块分别进行控制，提高了系统的控制效率，使系统运行更加稳定，第一控制单元用于控制换流器的功率模块部分，第一控制单元在控制换流器负载电流的同时对换流器的功率模块的电容电压进行平衡控制。 The control circuit includes a carrier phase shift controller, a first control unit and a second control unit, the output terminals of the first control unit and the output terminals of the second control unit are connected to the carrier phase shift controller, and the first The control unit includes a first comparator, a second comparator, a third comparator, a fourth comparator, a fifth comparator, a first PI controller, a second PI controller, a third PI controller, a first coordinate conversion device, the second coordinate converter, the first reactor and the second reactor, the first comparator, the first PI controller, the second comparator, the second PI controller and the third comparator are sequentially connected in series and connected to the first A coordinate converter, the fourth comparator, the third PI controller and the fifth comparator are sequentially connected in series to the first coordinate converter, the output end of the first coordinate converter is connected to the carrier phase shift controller, and the carrier phase shift control The output end of the converter is connected to the drive circuit, the output end of the converter is connected to the second coordinate transformer, the first output end of the second coordinate transformer is respectively connected to the second comparator and the second reactor, the second reactor The output end is connected to the third comparator, the other output end of the second coordinate transformer is respectively connected to the fourth comparator and the first reactor, and the output end of the first reactor is connected to the fifth comparator, using two parallel controllers The power module and the full bridge module of the converter are controlled separately, which improves the control efficiency of the system and makes the system run more stably. The first control unit is used to control the power module part of the converter, and the first control unit controls the converter While controlling the load current of the converter, the capacitor voltage of the power module of the converter is balancedly controlled.

所述第二控制单元包括第六比较器、第七比较器、第八比较器、第九比较器、第一比例控制器、第二比例控制器、第四PI控制器、第五PI控制器和函数模块，第六比较器、第一比例控制器、第七比较器、第四PI控制器和第二比例控制器依次串联后接入第九比较器，第八比较器、第五PI控制器和函数模块依次串联后接入第九比较器，第九比较器的输出端连接所述载波移相控制器，通过控制全桥模块的输出电压来控制环流，即通过采用合理的控制策略，在全桥模块中插入适当的电压可以抑制环流，通过独立的控制单元不仅抑制环流，同时实现了对全桥模块电容电压的均衡控制。 The second control unit includes a sixth comparator, a seventh comparator, an eighth comparator, a ninth comparator, a first proportional controller, a second proportional controller, a fourth PI controller, and a fifth PI controller And the function module, the sixth comparator, the first proportional controller, the seventh comparator, the fourth PI controller and the second proportional controller are sequentially connected in series to the ninth comparator, the eighth comparator, and the fifth PI control The circuit breaker and the function module are connected in series in sequence to the ninth comparator, the output terminal of the ninth comparator is connected to the carrier phase shift controller, and the circulating current is controlled by controlling the output voltage of the full bridge module, that is, by adopting a reasonable control strategy, Inserting an appropriate voltage into the full-bridge module can suppress the circulating current. The independent control unit not only suppresses the circulating current, but also realizes the balanced control of the capacitor voltage of the full-bridge module.

所述控制电路包括DSP模块和FPGA模块，DSP模块的输出端连接FPGA模块，FPGA模块的输出端连接驱动电路，所述第一控制单元和第二控制单元集成在DSP模块内，所述载波移相控制器集成在FPGA模块内，采用DSP+FPGA的控制方式，DSP作为运算和控制部分，FPGA用来产生PWM波，这样大大提高了控制、运算速度，降低了整个装置的响应时间。 The control circuit includes a DSP module and an FPGA module, the output of the DSP module is connected to the FPGA module, the output of the FPGA module is connected to the drive circuit, the first control unit and the second control unit are integrated in the DSP module, and the carrier wave shift The phase controller is integrated in the FPGA module, adopting the control method of DSP+FPGA, DSP is used as the operation and control part, and FPGA is used to generate PWM waves, which greatly improves the control and operation speed and reduces the response time of the whole device.

所述基于新型模块化多电平拓扑结构的无功补偿装置包括过零检测电路，所述过零检测电路包括电压传感器、比较电路和光耦，电压传感器的输入端连接三相交流电源的输出端，电压传感器的输出端连接比较电路的输入端，比较电路的输出端通过光耦接入DSP模块，使无功补偿装置实现了对电网电压的锁相过程。 The reactive power compensation device based on the new modular multi-level topology structure includes a zero-crossing detection circuit, the zero-crossing detection circuit includes a voltage sensor, a comparison circuit and an optocoupler, and the input end of the voltage sensor is connected to the output end of the three-phase AC power supply , the output terminal of the voltage sensor is connected to the input terminal of the comparison circuit, and the output terminal of the comparison circuit is connected to the DSP module through an optocoupler, so that the reactive power compensation device realizes the phase-locking process of the grid voltage.

所述信号检测电路包括电流检测及调理电路，所述电流检测及调理电路包括电流传感器和光耦芯片，电流传感器的输出端连接光耦芯片，减少电流采样环节引起的滞后和提高检测信号的抗干扰能力。 The signal detection circuit includes a current detection and conditioning circuit, the current detection and conditioning circuit includes a current sensor and an optocoupler chip, and the output end of the current sensor is connected to the optocoupler chip to reduce the hysteresis caused by the current sampling link and improve the anti-interference of the detection signal ability.

所述驱动电路包括电平转换电路和驱动芯片，电平转换电路的输入端为驱动电路的输入端，电平转换电路的输出端连接驱动芯片，驱动芯片的输出端为驱动电路的输出端。 The driving circuit includes a level shifting circuit and a driving chip, the input end of the level shifting circuit is the input end of the driving circuit, the output end of the level shifting circuit is connected to the driving chip, and the output end of the driving chip is the output end of the driving circuit.

所述基于新型模块化多电平拓扑结构的无功补偿装置的控制方法，具体包括：采用基于瞬时无功功率理论的i_p-i_q电流检测法对电流进行检测；对半桥单元和全桥单元分别进行控制，得到PWM控制信号，定义半桥单元为功率模块，全桥单元为全桥模块，所述功率模块的控制包括对功率模块的电容均压控制和基于前馈解耦的电压电流双闭环控制，全桥模块的控制包括对全桥模块电容电压均衡和整个无功补偿装置的环流的抑制控制。 The control method of the reactive power compensation device based on the new modular multi-level topology specifically includes: detecting the current by using the i _p -i _q current detection method based on the instantaneous reactive power theory; The bridge units are controlled separately to obtain the PWM control signal, and the half-bridge unit is defined as a power module, and the full-bridge unit is a full-bridge module. The control of the power module includes the capacitor voltage equalization control of the power module and the voltage based on feedforward decoupling Current double closed-loop control, the control of the full bridge module includes the suppression control of the capacitor voltage balance of the full bridge module and the circulation of the entire reactive power compensation device.

所述电容均压控制包括相间电压平衡控制和独立电压平衡控制，所述相间电压平衡控制为：电压外环是使每相2N个功率模块电容电压的平均值与功率模块电容电压的给定值U_ref作差，经PI调节后与该相的环流作差，经电流环比例调节后作为平均电压控制的指令信号，定义为u_Aj，其中，j＝a,b,c，所述与环流i_cir的表达式分别为： The capacitor voltage equalization control includes phase-to-phase voltage balance control and independent voltage balance control, and the phase-to-phase voltage balance control is: the voltage outer loop is the average value of the capacitor voltages of 2N power modules in each phase Make a difference with the given value U _ref of the capacitor voltage of the power module, make a difference with the circulating current of this phase after PI adjustment, and use it as the command signal for average voltage control after the current loop proportional adjustment, defined as u _Aj , where j=a ,b,c, said and the expressions of the circulation i _cir are:

${\overset{&OverBar; &OverBar;}{U u}}_{c c} = = \frac{11}{22 N N} {Σ Σ}_{j j = = 11}^{22 N N} {U u}_{c c j j}$

${i i}_{c c i i r r} = = \frac{11}{22} (({i i}_{P P} + + {i i}_{N N}))$

独立电容电压平衡控制为：检测每相上、下桥臂功率模块的电容电压U_cji，将其与功率模块的电容电压的给定值U_ref相比较，经比例调节，再与相应的桥臂电流相乘，经过符号函数的修正，得到每相上、下桥臂独立电压平衡控制的指令u_BPj和u_BNj。采用电容电压分级控制策略，确保各模块电容电压均衡，实现了良好的无功补偿、稳压、均压以及环流抑制效果。 The independent capacitor voltage balance control is as follows: detect the capacitor voltage U _cji of the upper and lower bridge arm power modules of each phase, compare it with the given value U _ref of the capacitor voltage of the power module, adjust the ratio, and then compare it with the corresponding bridge arm The currents are multiplied, and after the correction of the sign function, the instructions u _BPj and u _BNj for the independent voltage balance control of the upper and lower bridge arms of each phase are obtained. The capacitance voltage classification control strategy is adopted to ensure that the capacitance voltage of each module is balanced, and to achieve good reactive power compensation, voltage stabilization, voltage equalization and circulation suppression effects.

所述基于前馈解耦的电压电流双闭环控制的前馈解耦控制策略为： The feedforward decoupling control strategy of the voltage and current double closed-loop control based on feedforward decoupling is:

$[\begin{matrix} {x x}_{11} \\ {x x}_{22} \end{matrix}] = = [\begin{matrix} {K K}_{11} (({i i}_{d d}^{* *} - - {i i}_{d d})) + + \frac{{K K}_{11}}{{T T}_{11}} &Integral; &Integral; (({i i}_{d d}^{* *} - - {i i}_{d d})) d d t t \\ {K K}_{22} (({i i}_{q q}^{* *} - - {i i}_{q q})) + + \frac{{K K}_{22}}{{T T}_{22}} &Integral; &Integral; (({i i}_{q q}^{* *} - - {i i}_{q q})) d d t t \end{matrix}]$

其中，x₁，x₂为中间变量，K₁、K₂为比例系数，T₁，T₂为积分系数，i_d和i_q分别为换流器输出的dq坐标轴的电流分量，i_d ^*和i_q ^*分别为负载中的有功电流分量和无功电流分量； Among them, x ₁ and x ₂ are intermediate variables, K ₁ and K ₂ are proportional coefficients, T ₁ and T ₂ are integral coefficients, i _d and i _q are the current components of the dq coordinate axes output by the converter, and i _d ^* and i _q ^* are active current components and reactive current components in the load respectively;

所述电压电流双闭环控制包括：将给定直流电压U_ref与换流器直流侧电容电压进行比较，经电压PI调节，其输出作为有功电流的给定值i_d ^*，负载电流经坐标变换、取反处理得到无功电流给定值i_q ^*；将换流器输出的三相电流i_abc经坐标变换得到i_d和i_q，将i_d和i_q分别与i_d ^*、i_q ^*进行比较，经电流环PI调节，得到期望的输出电压V_cd、V_cq；对V_cd、V_cq进行逆变换得到静止坐标系下的三相调制波，将该三相调制波与载波移相控制器产生的三角载波比较后得到PWM控制信号。 The voltage and current double closed-loop control includes: comparing the given DC voltage U _ref with the capacitor voltage on the DC side of the converter, adjusting the voltage PI, and outputting it as the given value _id ^* of the active current, and transforming the load current through coordinate transformation , Negative processing to get the given value of reactive current i _q ^* ; Transform the three-phase current i _abc output by the converter to get i _d and i _q through coordinate transformation, and compare i _d and i _q with i _d ^* and i _q respectively ^* For comparison, the desired output voltages V _cd and V _cq are obtained through the adjustment of the current loop PI; the inverse transformation of V _cd and V _cq is carried out to obtain the three-phase modulation wave in the static coordinate system, and the three-phase modulation wave and the carrier wave are shifted The PWM control signal is obtained after comparing the triangular carrier generated by the phase controller.

所述全桥模块电容电压均衡过程包括：每一相上、下桥臂全桥模块电容电压的给定值V_H,ref与全桥模块的实际电容电压进行比较，经PI控制器，其输出乘以该桥臂电流的符号函数后，将生成的全桥子模块电压给定值V_r,ref与三角载波比较后得到PWM波，其中，r＝P,N，驱动全桥模块中相应的功率开关管，对全桥子模块的电容进行充放电控制，实现全桥模块电容电压的均衡； The full-bridge module capacitor voltage equalization process includes: the given value V _H,ref of the full-bridge module capacitor voltage of each phase upper and lower bridge arms is compared with the actual capacitor voltage of the full-bridge module, and through the PI controller, its output After multiplying by the sign function of the bridge arm current, compare the generated full-bridge sub-module voltage given value V _r,ref with the triangular carrier to obtain a PWM wave, where r=P,N, drive the corresponding in the full-bridge module The power switch tube controls the charge and discharge of the capacitors of the full-bridge sub-modules, and realizes the balance of the capacitor voltage of the full-bridge modules;

所述环流抑制过程包括：将每一相环流分别与环流的参考值进行比较，此时环流的参考值i_cir,ref＝i_dc/3，得到的结果通过一个比例控制器形成一个全桥模块的电压调整信号，将这个电压调整信号平均分成2份，分别加在该相上、下桥臂全桥模块的电压信号上。针对三相间的环流问题，提出了一种增加H桥模块的新型拓扑结构，通过独立的控制单元，实现对环流的有效抑制。 The circulation suppression process includes: comparing the circulation current of each phase with the reference value of the circulation current, at this time, the reference value of the circulation current i _cir,ref = i _dc /3, and the obtained result forms a full-bridge module through a proportional controller Divide the voltage adjustment signal into two parts on average, and add them to the voltage signals of the upper and lower bridge arm full-bridge modules of the phase respectively. Aiming at the circulation problem among the three phases, a new topology structure with H-bridge modules is proposed, and the circulation current can be effectively suppressed through an independent control unit.

本实用新型与现有技术相比具有以下效果：本实用新型不但能补偿电网无功，解决系统三相不平衡问题，还有支撑电网电压、抑制环流的作用，所述信号检测单元检测电网三相电压、负载侧三相电流、换流器输出的三相反馈电流、直流侧各模块的电容电压以及三相的桥臂电流，然后，将检测到反馈量在控制电路中进行运算和调节，得到PWM控制信号；最后，将控制信号进行功放以驱动换流器的各个模块中功率开关管，使换流器输出相应的补偿电流，实现无功补偿。本实用新型设计的新型拓扑中每相上、下桥臂各加入了1个H桥单元，这样整个三相系统中的H桥单元共有6个。换流器由半桥单元和H桥单元构成，其中半桥单元模块用于控制桥臂中的基波负载电流，而H桥单元作为电压校正模块用于抑制环流。采用基于瞬时无功功率理论的i_p-i_q电流检测法，该方法具有较好的实时性，能准确的检测出电网中的有功电流和无功电流的大小，提高系统的稳定性。本实用新型的换流器相对于传统的多电平换流器具有明显的优势，能够实现对无功功率、谐波以及不平衡的综合补偿，其输出为多电平，接近于正弦波，谐波含量小，选择适当的控制算法可减小功率管的开关频率，降低开关损耗。 Compared with the prior art, the utility model has the following effects: the utility model can not only compensate the reactive power of the grid, solve the three-phase unbalance problem of the system, but also support the grid voltage and suppress the circulation. Phase voltage, three-phase current on the load side, three-phase feedback current output by the converter, capacitor voltage of each module on the DC side, and three-phase bridge arm current, and then calculate and adjust the detected feedback amount in the control circuit, The PWM control signal is obtained; finally, the control signal is amplified to drive the power switch tubes in each module of the converter, so that the converter outputs a corresponding compensation current to realize reactive power compensation. In the new topology designed by the utility model, one H-bridge unit is added to the upper and lower bridge arms of each phase, so that there are 6 H-bridge units in the whole three-phase system. The converter is composed of a half-bridge unit and an H-bridge unit. The half-bridge unit module is used to control the fundamental load current in the bridge arm, and the H-bridge unit is used as a voltage correction module to suppress the circulating current. The i _p -i _q current detection method based on the theory of instantaneous reactive power is adopted. This method has better real-time performance and can accurately detect the magnitude of active current and reactive current in the power grid, improving the stability of the system. Compared with the traditional multi-level converter, the converter of the utility model has obvious advantages, and can realize comprehensive compensation for reactive power, harmonics and unbalance, and its output is multi-level, close to sine wave, The harmonic content is small, and choosing an appropriate control algorithm can reduce the switching frequency of the power tube and reduce the switching loss.

附图说明 Description of drawings

图1、系统整体框图； Figure 1. The overall block diagram of the system;

图2、新型MMC换流器拓扑结构电路图； Figure 2. The topology circuit diagram of the new MMC converter;

图3、STATCOM的等效控制模型示意图； Figure 3. Schematic diagram of the equivalent control model of STATCOM;

图4、相间电压平衡控制示意图； Figure 4. Schematic diagram of phase-to-phase voltage balance control;

图5、独立电压平衡控制示意图； Figure 5. Schematic diagram of independent voltage balance control;

图6、环流抑制器工作原理示意图； Figure 6. Schematic diagram of the working principle of the circulation suppressor;

图7、模块的工作状态示意图，其中(a)为投入状态，(b)为切除状态，(c)为闭锁状态； Figure 7. Schematic diagram of the working state of the module, where (a) is the input state, (b) is the cut state, and (c) is the locked state;

图8、新型三相MMC拓扑结构的等效模型示意图； Figure 8. Schematic diagram of the equivalent model of the new three-phase MMC topology;

图9、无功电流检测框图； Figure 9. Reactive current detection block diagram;

图10、有功、无功电流控制框图； Figure 10. Active and reactive current control block diagram;

图11、前馈解耦等效控制框图； Figure 11. Feedforward decoupling equivalent control block diagram;

图12、STATCOM电压电流双闭环的控制框图； Figure 12. The control block diagram of STATCOM voltage and current double closed loop;

图13、STATCOM系统总控制框图； Figure 13, STATCOM system general control block diagram;

图14、控制器2的工作原理图； Fig. 14, the working principle diagram of controller 2;

图15、过零检测电路图； Figure 15. Zero-crossing detection circuit diagram;

图16、电流检测及其调理电路电路图； Figure 16. Current detection and its conditioning circuit circuit diagram;

图17、隔离驱动电路电路图； Figure 17. Circuit diagram of the isolation drive circuit;

图18、主程序流程图； Figure 18, the main program flow chart;

图19、捕获中断流程图； Figure 19. Capture interrupt flow chart;

图20、T1中断子程序流程图； Figure 20, T1 interrupt subroutine flow chart;

图21、补偿前a相电压、电流波形图； Figure 21. Phase a voltage and current waveforms before compensation;

图22、补偿后a相电压、电流波形图； Figure 22. Phase A voltage and current waveforms after compensation;

图23、补偿前a相电流波形图； Figure 23. Waveform diagram of phase a current before compensation;

图24、补偿后a相电流波形图； Figure 24. Waveform diagram of phase a current after compensation;

图25、平衡控制后a相模块电容电压波形； Figure 25. Capacitor voltage waveform of phase a module after balance control;

图26、抑制前的三相环流波形； Figure 26. Three-phase circulating current waveform before suppression;

图27、抑制后的三相环流波形。 Figure 27. Three-phase circulating current waveform after suppression.

具体实施方式 detailed description

结合附图说明本实用新型的具体实施方式，本实用新型的基于新型模块化多电平拓扑结构的无功补偿装置，包括三相交流电源、阻感负载、MMC换流器、控制电路、信号检测电路和驱动电路，所述MMC换流器包括三个结构相同并联连接的桥臂，每个桥臂包括关于桥臂中点对称且串联连接的上桥臂和下桥臂，所述上桥臂包括相互串联的电感、若干半桥单元和一个H桥单元，上桥臂的电感与下桥臂的电感串联连接，MMC换流器三个桥臂的中点通过导线并联接在三相交流电源和阻感负载之间，信号检测电路的输入端分别连接三相交流电源的输出端、阻感负载的输入端、MMC换流器的输出端、MMC换流器三个桥臂、MMC换流器的每个半桥单元和H桥单元，信号检测电路的输出端连接控制电路的输入端，控制电路的输出端通过驱动电路与MMC换流器建立连接，半桥单元模块用于控制桥臂中的负载基波电流，H桥单元作为电压校正模块用于抑制环流，信号检测电路，检测电网三相电压、负载侧三相电流、MMC输出的三相反馈电流、直流侧各子模块的电容电压以及三相的桥臂电流；然后，将检测到反馈量在控制单元中进行运算和调节，得到PWM控制信号；最后，将控制信号进行功放以驱动MMC子模块中功率开关管，使换流器输出相应的补偿电流，实现无功补偿。 The specific embodiment of the utility model is illustrated in conjunction with the accompanying drawings. The reactive power compensation device based on the novel modular multi-level topology of the utility model includes a three-phase AC power supply, a resistive load, an MMC converter, a control circuit, a signal A detection circuit and a drive circuit, the MMC converter includes three bridge arms with the same structure connected in parallel, each bridge arm includes an upper bridge arm and a lower bridge arm connected in series symmetrically with respect to the midpoint of the bridge arm, the upper bridge arm The arm includes inductors connected in series, several half-bridge units and an H-bridge unit. The inductor of the upper bridge arm is connected in series with the inductor of the lower bridge arm. The midpoints of the three bridge arms of the MMC converter are connected in parallel to the three-phase AC through wires. Between the power supply and the resistive load, the input terminal of the signal detection circuit is respectively connected to the output terminal of the three-phase AC power supply, the input terminal of the resistive load, the output terminal of the MMC converter, the three bridge arms of the MMC converter, and the MMC converter. Each half-bridge unit and H-bridge unit of the converter, the output terminal of the signal detection circuit is connected to the input terminal of the control circuit, the output terminal of the control circuit is connected with the MMC converter through the drive circuit, and the half-bridge unit module is used to control the bridge The load fundamental wave current in the arm, the H bridge unit is used as a voltage correction module to suppress the circulating current, the signal detection circuit detects the three-phase voltage of the grid, the three-phase current of the load side, the three-phase feedback current output by the MMC, and the sub-modules of the DC side The capacitor voltage and the three-phase bridge arm current; then, the detected feedback is calculated and adjusted in the control unit to obtain the PWM control signal; finally, the control signal is amplified to drive the power switch in the MMC sub-module, so that the switching The rectifier outputs the corresponding compensation current to realize reactive power compensation.

本实施方式的模块化多电平MMC换流器的拓扑结构具有公共直流母线，三相间能量能够相互流动，在电网畸变的情况下，能够实现无功功率、谐波以及不平衡的综合补偿，高度模块化，易于冗余设计，而且输出为多电平，接近于正弦波，谐波含量小。 The topology of the modular multilevel MMC converter in this embodiment has a common DC bus, and the energy between the three phases can flow mutually. In the case of grid distortion, comprehensive compensation for reactive power, harmonics and unbalance can be realized. Highly modular, easy for redundant design, and the output is multi-level, close to sine wave, with small harmonic content.

将半桥模块称为功率模块，H桥单元称为全桥模块。如图2所示，SM为半桥模块，每个半桥模块由两个带有反向续流二极管的IGBT和1个储能电容C组成，每个半桥模块有两种开关状态，控制每一相中模块的通断，能使模块的输出电压通过叠加形成2N+1个电平的输出电压。 The half-bridge module is called a power module, and the H-bridge unit is called a full-bridge module. As shown in Figure 2, SM is a half-bridge module, each half-bridge module is composed of two IGBTs with reverse freewheeling diodes and an energy storage capacitor C, each half-bridge module has two switching states, the control The on-off of the modules in each phase can make the output voltages of the modules superimposed to form an output voltage of 2N+1 levels.

MMC换流器的功率模块工作状态如图7所示，图中箭头表明电流的流向。所示功率模块共有三种工作状态： The working state of the power module of the MMC converter is shown in Figure 7, and the arrows in the figure indicate the flow direction of the current. The power modules shown have three operating states:

1)T₁(D₁)开通、T₂(D₂)关断为投入状态，如图7中图(a)所示； 1) T ₁ (D ₁ ) is turned on and T ₂ (D ₂ ) is turned off to be in the input state, as shown in (a) in Figure 7;

2)T₁(D₁)关断、T₂(D₂)开通为切除状态，如图7中图(b)所示； 2) When T ₁ (D ₁ ) is off and T ₂ (D ₂ ) is on, it is in the cut-off state, as shown in (b) in Figure 7;

3)T₁和T₂均关断为闭锁状态，如图7中图(c)所示； 3) T ₁ and T ₂ are both turned off and in a locked state, as shown in (c) in Figure 7;

设s_i为MMC换流器的功率模块的开关函数，可表示为 Let _si be the switching function of the power module of the MMC converter, which can be expressed as

则每个功率模块的等效输出电压u_o可表示为 Then the equivalent output voltage u _o of each power module can be expressed as

u_o＝s_iV_d(2) u _o =s _i V _d (2)

HB为H桥单元，有3种开关状态，所示MMC全桥模块的开关函数s_i，s_i∈{-1,0,1}，s_i的取值决定了H桥单元输出电压的极性。 HB is an H-bridge unit, which has three switching states. The switching function s _i of the MMC full-bridge module shown is s _i ∈ {-1,0,1}, and the value of s _i determines the pole of the output voltage of the H-bridge unit. sex.

新型三相MMC换流器功率拓扑结构的等效模型如图8所示，其中，直流侧电流为i_dc，三相上桥臂电流分别为i_aP、i_bP、i_cP，三相下桥臂电流分别为i_aN、i_bN、i_cN，三相输出电流分别为i_a、i_b、i_c。 The equivalent model of the new three-phase MMC converter power topology is shown in Fig. 8, where the DC side current is i _dc , the three-phase upper bridge arm currents are i _aP , i _bP , and i _cP respectively, and the three-phase lower bridge arm currents are i aP , i bP , and i cP The arm currents are i _aN , i _bN , and i _cN respectively, and the three-phase output currents are respectively i _a , i _b , and i _c .

以a相为例，根据KCL电路理论可知，a相输出电流可表示为 Taking phase a as an example, according to KCL circuit theory, the output current of phase a can be expressed as

i_a＝i_aP-i_aN(3) i _a =i _aP -i _aN (3)

设a相桥臂的环流为i_cir,a，因上、下桥臂的电路结构相同，则有 Let the circulating current of the bridge arm of phase a be i _cir,a , because the circuit structure of the upper and lower bridge arms is the same, then

${j j}_{a a P P} = = {i i}_{c c i i r r,, a a} + + \frac{{i i}_{a a}}{22} - - - - - - ((44))$

${i i}_{a a N N} = = {i i}_{c c i i r r,, a a} - - \frac{{i i}_{a a}}{22} - - - - - - ((55))$

将式(5)和(6)相加，得到 Adding equations (5) and (6), we get

${i i}_{c c i i r r,, a a} = = \frac{11}{22} (({i i}_{a a P P} + + {i i}_{a a N N})) - - - - - - ((66))$

三相MMC直流母线的电流i_dc为a、b、c三相环流之和，即 The current i _dc of the three-phase MMC DC bus is the sum of the three-phase circulating currents of a, b, and c, that is

i_dc＝i_cir,a+i_cir,b+i_cir,c(7) i _dc =i _cir,a +i _cir,b +i _cir,c (7)

由于三相对称，三相环流i_cir,j可表示为 Due to the three-phase symmetry, the three-phase circulating current i _cir,j can be expressed as

${i i}_{c c i i r r,, j j} = = \frac{{i i}_{d d c c}}{33} + + {i i}_{z z j j}^{* *} - - - - - - ((88))$

式中，i_zj ^*是环流中的二倍频负序交流分量，其中，j＝a,b,c，结合式(4)、(5)与(8)可得： In the formula, i _zj ^* is the double-frequency negative-sequence AC component in the circulating current, where j=a, b, c, combined with formulas (4), (5) and (8), it can be obtained:

${i i}_{a a P P} = = \frac{{i i}_{d d c c}}{33} + + \frac{{i i}_{a a}}{22} + + {i i}_{z z j j}^{* *} - - - - - - ((99))$

${i i}_{a a N N} = = \frac{{i i}_{d d c c}}{33} - - \frac{{i i}_{a a}}{22} + + {i i}_{z z j j}^{* *} - - - - - - ((1010))$

综合式(9)与(10)，可得a相环流的二倍频负序交流成分为 Combining formulas (9) and (10), it can be obtained that the double frequency negative-sequence AC component of a-phase circulation is

${i i}_{z z j j}^{* *} = = \frac{11}{22} (({i i}_{a a P P} + + {i i}_{a a N N})) - - \frac{{i i}_{d d c c}}{33} - - - - - - ((1111))$

MMC换流器功率模块可等效为可控的电压源V_jr，其中，j＝a,b,c；r＝P,N，则a相上桥输出电压V_aP和下桥臂的输出电压V_aN可表示为 The power module of the MMC converter can be equivalent to a controllable voltage source V _jr , where, j=a, b, c; r=P, N, then the output voltage V _aP of the upper bridge of phase a and the output voltage of the lower bridge arm V _aN can be expressed as

${V V}_{a a P P} = = {Σ Σ}_{i i = = 11}^{N N} {S S}_{i i} {V V}_{d d} - - - - - - ((1212))$

${V V}_{a a N N} = = {Σ Σ}_{i i = = N N + + 11}^{22 N N} {S S}_{i i} {V V}_{d d} - - - - - - ((1313))$

在图8的等效模型中，全桥模块的输出电压为V_H,jr，以直流母线电压的中点为参考，MMC系统输出的三相电压为V_j，每个桥臂的等效电阻为R_e，根据KVL电路理论，可以得到 In the equivalent model in Figure 8, the output voltage of the full-bridge module is V _H,jr , taking the midpoint of the DC bus voltage as a reference, the three-phase voltage output by the MMC system is V _j , and the equivalent resistance of each bridge arm is R _e , according to the KVL circuit theory, we can get

$\frac{{U u}_{d d}}{22} - - {V V}_{a a P P} - - {V V}_{H h,, a a P P} - - {V V}_{a a} = = {R R}_{e e} {i i}_{a a P P} + + L L \frac{{di di}_{a a P P}}{d d t t} - - - - - - ((1414))$

$\frac{{U u}_{d d}}{22} - - {V V}_{a a N N} - - {V V}_{H h,, a a N N} + + {V V}_{a a} = = {R R}_{e e} {i i}_{a a N N} + + L L \frac{{di di}_{a a N N}}{d d t t} - - - - - - ((1515))$

将式(14)和(15)相加，再结合式(4)至(6)，可得： Adding formulas (14) and (15), and combining formulas (4) to (6), we can get:

$22 L L \frac{{di di}_{c c i i r r,, a a}}{d d t t} + + 22 {R R}_{e e} {i i}_{c c i i r r,, a a} = = (({U u}_{d d} - - {V V}_{a a P P} - - {V V}_{a a N N})) - - (({V V}_{H h,, a a P P} + + {V V}_{H h,, a a N N})) - - - - - - ((1616))$

由式(16)可以看出，可以通过控制全桥模块的输出电压(V_H,aP+V_H,aN)大小与电压差(U_d-V_aP-V_aN)相等从而达到消除环流的目的。 It can be seen from formula (16) that the purpose of eliminating circulating current can be achieved by controlling the output voltage (V _H,aP +V _H,aN ) of the full-bridge module to be equal to the voltage difference (U _d -V _aP -V _aN ) .

将式(15)减去式(14)，可得 Subtract formula (14) from formula (15) to get

${V V}_{a a} = = \frac{11}{22} (({V V}_{a a N N} - - {V V}_{a a P P})) + + \frac{11}{22} (({V V}_{H h,, a a N N} - - {V V}_{H h,, a a P P})) + + \frac{{R R}_{e e}}{22} (({i i}_{a a N N} - - {i i}_{a a P P})) + + \frac{L L}{22} ((\frac{{di di}_{a a N N}}{d d t t} - - \frac{{di di}_{a a P P}}{d d t t})) - - - - - - ((1717))$

由式(17)可见，全桥模块对MMC的输出电压几乎没有影响，原因有二：首先，全桥模块的输出电压等级相对于MMC系统的输出电压来说是很小；其次，控制全桥模块插入每一相上、下桥臂的电压是相等的，因此式(17)中(V_H,aP-V_H,aN)/2一项可近似看成0，并不影响系统的输出电压。 It can be seen from formula (17) that the full-bridge module has almost no influence on the output voltage of the MMC for two reasons: first, the output voltage level of the full-bridge module is relatively small compared to the output voltage of the MMC system; secondly, the control of the full-bridge The voltage of the upper and lower bridge arms of each phase when the module is inserted is equal, so the term (V _H,aP -V _H,aN )/2 in formula (17) can be approximately regarded as 0, which does not affect the output voltage of the system .

所述控制电路包括载波移相控制器、第一控制单元和第二控制单元，第一控制单元的输出端和第二控制单元的输出端均与载波移相控制器建立连接，所述第一控制单元包括第一比较器1、第二比较器3、第三比较器5、第四比较器7、第五比较器9、第一PI控制器2、第二PI控制器4、第三PI控制器8、第一坐标转换器6、第二坐标转换器12、第一电抗器10和第二电抗器11，第一比较器1、第一PI控制器2、第二比较器3、第二PI控制器4和第三比较器5依次串联后接入第一坐标变换器，第四比较器7、第三PI控制器8和第五比较器9依次串联后接入第一坐标转换器6，第一坐标转换器6的输出端连接载波移相控制器，载波移相控制器的输出端连接驱动电路，所述MMC换流器的输出端连接第二坐标变换器，第二坐标变换器的第一输出端分别连接第二比较器3和第二电抗器11，第二电抗器11的输出端连接第三比较器5，第二坐标变换器的另一输出端分别连接第四比较器7和第一电抗器10，第一电抗器10的输出端连接第五比较器9。 The control circuit includes a carrier phase shift controller, a first control unit and a second control unit, the output terminals of the first control unit and the output terminals of the second control unit are connected to the carrier phase shift controller, and the first The control unit includes a first comparator 1, a second comparator 3, a third comparator 5, a fourth comparator 7, a fifth comparator 9, a first PI controller 2, a second PI controller 4, a third PI Controller 8, first coordinate converter 6, second coordinate converter 12, first reactor 10 and second reactor 11, first comparator 1, first PI controller 2, second comparator 3, second The second PI controller 4 and the third comparator 5 are serially connected to the first coordinate converter, and the fourth comparator 7, the third PI controller 8 and the fifth comparator 9 are serially connected to the first coordinate converter. 6. The output end of the first coordinate converter 6 is connected to the carrier phase shift controller, the output end of the carrier phase shift controller is connected to the driving circuit, the output end of the MMC converter is connected to the second coordinate converter, and the second coordinate conversion The first output end of the transformer is respectively connected to the second comparator 3 and the second reactor 11, the output end of the second reactor 11 is connected to the third comparator 5, and the other output end of the second coordinate transformer is respectively connected to the fourth comparator 7 and the first reactor 10, the output terminal of the first reactor 10 is connected to the fifth comparator 9.

所述第二控制单元包括第六比较器13、第七比较器15、第八比较器18、第九比较器21、第一比例控制器14、第二比例控制器17、第四PI控制器16、第五PI控制器19和函数模块20，第六比较器13、第一比例控制器14、第七比较器15、第四PI控制器16和第二比例控制器17依次串联后接入第九比较器21，第八比较器18、第五PI控制器19和函数模块20依次串联后接入第九比较器21，第九比较器21的输出端连接所述载波移相控制器。 The second control unit includes a sixth comparator 13, a seventh comparator 15, an eighth comparator 18, a ninth comparator 21, a first proportional controller 14, a second proportional controller 17, a fourth PI controller 16. The fifth PI controller 19 and the function module 20, the sixth comparator 13, the first proportional controller 14, the seventh comparator 15, the fourth PI controller 16 and the second proportional controller 17 are sequentially connected in series The ninth comparator 21, the eighth comparator 18, the fifth PI controller 19 and the function module 20 are sequentially connected in series to the ninth comparator 21, and the output terminal of the ninth comparator 21 is connected to the carrier phase shift controller.

本实施例的调制技术采用载波移相控制策略。 The modulation technique in this embodiment adopts a carrier phase shift control strategy.

所述控制电路包括DSP模块和FPGA模块，DSP模块的输出端连接FPGA模块，FPGA模块的输出端连接驱动电路，所述第一控制单元和第二控制单元集成在DSP模块内，所述载波移相控制器集成在FPGA模块内，本实施例的DSP控制模块以TI公司的TMS320F2812为核心，DSP模块和FPGA模块配合实现负载侧电流采样、补偿电流采样、电容电压采样、桥臂电流采样、电流电压双闭环控制、CPS-SPWM波生成等功能，辅助电路由开关电源、保护电路等组成。 The control circuit includes a DSP module and an FPGA module, the output of the DSP module is connected to the FPGA module, the output of the FPGA module is connected to the drive circuit, the first control unit and the second control unit are integrated in the DSP module, and the carrier wave shift The phase controller is integrated in the FPGA module. The DSP control module of this embodiment takes TI's TMS320F2812 as the core, and the DSP module and the FPGA module cooperate to realize load-side current sampling, compensation current sampling, capacitor voltage sampling, bridge arm current sampling, current Voltage double closed-loop control, CPS-SPWM wave generation and other functions, the auxiliary circuit is composed of switching power supply, protection circuit and so on.

所述基于新型模块化多电平拓扑结构的无功补偿装置包括过零检测电路，所述过零检测电路包括电压传感器22、比较电路23和光耦24，电压传感器22的输入端连接三相交流电源的输出端，电压传感器22的输出端连接比较电路23的输入端，比较电路23的输出端通过光耦24接入DSP模块。本实施方式的采用电压霍尔模块CHV-25P把A相电压幅值降为5V左右，然后通过比较电路23产生一个与电网同频的方波信号，其上升沿与a相正向电压的过零点重合，两个上升沿的时间差即为电网a相电压的周期。 The reactive power compensation device based on the new modular multilevel topology structure includes a zero-crossing detection circuit, and the zero-crossing detection circuit includes a voltage sensor 22, a comparison circuit 23 and an optocoupler 24, and the input terminal of the voltage sensor 22 is connected to a three-phase AC The output terminal of the power supply and the output terminal of the voltage sensor 22 are connected to the input terminal of the comparison circuit 23 , and the output terminal of the comparison circuit 23 is connected to the DSP module through the optocoupler 24 . In this embodiment, the voltage Hall module CHV-25P is used to reduce the voltage amplitude of phase A to about 5V, and then a square wave signal with the same frequency as the power grid is generated through the comparison circuit 23. When the zero point coincides, the time difference between the two rising edges is the period of the phase a voltage of the power grid.

基于新型模块化多电平拓扑结构的无功补偿装置包括电流检测及调理电路，所述电流检测及调理电路包括电流传感器25、光电隔离放大器26和运算放大器27，电流传感器25的输出端连接光电隔离放大器26，光电隔离放大器26的输出端连接运算放大器27，运算放大器27的输出端为电流检测及调理电路的输出端，本实用新型采用高速的电流霍尔模块CHB-25NP实现三相电流检测，并利用光电隔离放大器26进行隔离，以提高检测信号的抗干扰能力，本实施方式采用的光电隔离放大器的型号为HCPL7840，运算放大器27的型号为LF358，运算放大器27给光耦HCPL7840的输出电压加上2V的偏置。 The reactive power compensation device based on the new modular multi-level topology structure includes a current detection and conditioning circuit, the current detection and conditioning circuit includes a current sensor 25, a photoelectric isolation amplifier 26 and an operational amplifier 27, and the output terminal of the current sensor 25 is connected to a photoelectric The isolation amplifier 26, the output end of the photoelectric isolation amplifier 26 is connected to the operational amplifier 27, the output end of the operational amplifier 27 is the output end of the current detection and conditioning circuit, the utility model adopts the high-speed current Hall module CHB-25NP to realize the three-phase current detection , and utilize photoelectric isolating amplifier 26 to isolate, to improve the anti-jamming ability of detection signal, the model of the photoelectric isolating amplifier that this embodiment adopts is HCPL7840, and the model of operational amplifier 27 is LF358, and operational amplifier 27 gives the output voltage of optocoupler HCPL7840 Add a bias of 2V.

所述驱动电路包括电平转换电路和驱动芯片，电平转换电路的输入端为驱动电路的输入端，电平转换电路的输出端连接驱动芯片，驱动芯片的输出端为驱动电路的输出端,本实施方式的无功补偿装置的隔离驱动电路如图17所示，将FPGA发出的PWM信号经过电平转换后，送给驱动芯片2SD315A，该芯片具有4000VAC的隔离电压，设有短路和过流保护功能，将2SD315A的工作模式设置为直接工作模式，这样可在INA和INB两个引脚中输入两个控制信号，并能同时驱动两个功率管。 The driving circuit includes a level shifting circuit and a driving chip, the input end of the level shifting circuit is the input end of the driving circuit, the output end of the level shifting circuit is connected to the driving chip, and the output end of the driving chip is the output end of the driving circuit, The isolated drive circuit of the reactive power compensation device in this embodiment is shown in Figure 17. After the PWM signal sent by the FPGA is level-converted, it is sent to the drive chip 2SD315A. This chip has an isolation voltage of 4000VAC and is equipped with short circuit and overcurrent. Protection function, set the working mode of 2SD315A to direct working mode, so that two control signals can be input into two pins of INA and INB, and two power transistors can be driven at the same time.

基于新型模块化多电平拓扑结构的无功补偿装置的控制方法，具体包括：采用基于瞬时无功功率理论的i_p-i_q电流检测法对电流进行检测；由于无功与谐波电流的检测值为电流补偿控制的给定值，所以无功与谐波电流检测的性能直接影响着STATCOM补偿的效果，采用基于瞬时无功功率理论的i_p-i_q电流检测法，具有较好的实时性，能准确的检测出电网中的有功电流和无功电流的大小，在检测基波无功电流时基本上是无延迟的。 The control method of the reactive power compensation device based on the new modular multi-level topology structure specifically includes: using the i _p -i _q current detection method based on the instantaneous reactive power theory to detect the current; due to the relationship between reactive power and harmonic current The detection value is the given value of current compensation control, so the performance of reactive power and harmonic current detection directly affects the effect of STATCOM compensation. Using the i _p -i _q current detection method based on the theory of instantaneous reactive power has a better Real-time, can accurately detect the size of the active current and reactive current in the grid, basically no delay when detecting the fundamental reactive current.

由于i_p-i_q谐波电流检测法只需三相瞬时电流和a相电网电压的角度信息，因而外部信号检测电路简单。另外，i_p-i_q检测法采用内部的参考正弦信号，没有直接使用系统电压信息参与运算，不受电网电压畸变或不对称的影响。 Since the i _p -i _q harmonic current detection method only needs angle information of three-phase instantaneous current and a-phase grid voltage, the external signal detection circuit is simple. In addition, the i _p -i _q detection method uses the internal reference sinusoidal signal, does not directly use the system voltage information to participate in the calculation, and is not affected by the grid voltage distortion or asymmetry.

首先，将三相负载电流i_a、i_b、i_c变换到dq坐标系下，即 First, transform the three-phase load currents _ia , _ib , and ic into the _dq coordinate system, namely

$[\begin{matrix} {i i}_{d d} \\ {i i}_{q q} \\ {i i}_{00} \end{matrix}] = = {C C}_{22 s the s / / 22 r r} {C C}_{33 s the s / / 22 s the s} [\begin{matrix} {i i}_{a a} \\ {i i}_{b b} \\ {i i}_{c c} \end{matrix}] = = {T T}_{a a b b c c - - d d q q} [\begin{matrix} {i i}_{a a} \\ {i i}_{b b} \\ {i i}_{c c} \end{matrix}] - - - - - - ((1818))$

其中，T_abc-dq为坐标变换矩阵为 Among them, T _abc-dq is the coordinate transformation matrix as

式中，ωt是与电网同步的旋转角度，由锁相环(PLL)来实现。 In the formula, ωt is the rotation angle synchronous with the power grid, which is realized by a phase-locked loop (PLL).

图9为系统无功和谐波电流检测原理图，图中的负载电流i_a、i_b、i_c经式(18)变换得到dq坐标下分量i_d、i_q。由瞬时无功功率理论可知，经坐标变换后得到负载中有功、无功电流分量i_d、i_q，经低通滤波处理后，得到反映负载电流基波分量的分别将i_d、i_q与作差得到反映谐波的有功、无功分量i_d-hrm、i_q-hrm。将以上的i_d-hrm、i_q-hrm和i_q进行组合，即可形成具有实际意义的三种电流，分别是：(1)i_d-hrm+i_q-hrm，为负载中谐波；(2)i_q，为负载中无功电流分量；(3)i_q+i_d-hrm，为负载中的无功和谐波。 Fig. 9 is a schematic diagram of reactive power and harmonic current detection of the system. The load currents _ia , _ib , and _ic in the figure are transformed by _formula (18) to obtain the components id and _iq in the dq coordinates. It can be known from the theory of instantaneous reactive power that the active and reactive current components i _d and i _q in the load are obtained after coordinate transformation, and after low-pass filtering, the fundamental component of the load current is obtained Respectively i _d , i _q and Make a difference to get the active and reactive components i _d-hrm and i _q-hrm reflecting the harmonics. Combining the above i _d-hrm , i _q-hrm and i _q , three currents with practical significance can be formed, namely: (1) i _d-hrm +i _q-hrm , which is the harmonic in the load ; (2) i _q , is the reactive current component in the load; (3) i _q +i _d-hrm , is the reactive power and harmonics in the load.

所述基于新型模块化多电平拓扑结构的无功补偿装置的控制方法包括对功率模块和全桥模块分别进行控制，功率模块的控制包括对功率模块的电容均压控制和基于前馈解耦的电压电流双闭环控制，全桥模块的控制包括对全桥模块电容电压均衡和整个无功补偿装置的环流的抑制。 The control method of the reactive power compensation device based on the new modular multi-level topology includes controlling the power module and the full bridge module respectively, and the control of the power module includes capacitor voltage equalization control of the power module and feed-forward decoupling based on The voltage and current double closed-loop control, the control of the full bridge module includes the balance of the capacitor voltage of the full bridge module and the suppression of the circulating current of the entire reactive power compensation device.

所述功率模块的电容均压控制包括相间电压平衡控制和独立电压平衡控制，所述相间电压平衡控制为：电压外环是使每相2N个功率模块电容电压的平均值与功率模块电容电压的给定值U_ref作差，经PI调节后与该相的环流作差，经电流环比例调节后作为平均电压控制的指令信号，定义为u_Aj，所述与环流i_cir的表达式分别为： The capacitor voltage equalization control of the power module includes phase-to-phase voltage balance control and independent voltage balance control, and the phase-to-phase voltage balance control is: the voltage outer loop is the average value of the capacitor voltages of 2N power modules in each phase It is different from the given value U _ref of the capacitor voltage of the power module, and after PI adjustment, it is different from the circulating current of this phase. After the proportional adjustment of the current loop, it is used as the command signal of the average voltage control, which is defined as u _Aj . and the expressions of the circulation i _cir are:

${i i}_{c c i i r r} = = \frac{11}{22} (({i i}_{P P} + + {i i}_{N N}))$

独立电容电压平衡控制如图5所示，检测每相上、下桥臂的模块电容电压U_cji，将其与模块电容电压的给定值U_ref相比较，经比例调节，再与相应的桥臂电流相乘，由于桥臂电流对于模块电容的充放电是有影响的，需增加符号函数加以修正：当该相上桥臂电流i_p>0时，符号函数为正，取1；当i_p<0时，符号函数为负，取-1。下桥臂与上桥臂类似，最终得到每相上、下桥臂独立电压平衡控制的指令u_BPj和u_BNj。 The independent capacitor voltage balance control is shown in Figure 5. It detects the module capacitor voltage U _cji of the upper and lower bridge arms of each phase, compares it with the given value U _ref of the module capacitor voltage, adjusts the ratio, and then compares it with the corresponding bridge arm Since the bridge arm current has an influence on the charge and discharge of the module capacitor, it is necessary to add a sign function to correct it: when the bridge arm current i _p >0 in this phase, the sign function is positive and takes 1; when i When _p < 0, the sign function is negative and takes -1. The lower bridge arm is similar to the upper bridge arm, and finally obtains instructions u _BPj and u _BNj for independent voltage balance control of the upper and lower bridge arms of each phase.

STATCOM系统的电流控制方法可分直接和间接控制两类。直接控制是直接对换流器的输出电流进行控制，主要有滞环控制和三角波比较控制；间接控制是将STATCOM等效为一个可控的交流电压源，通过调节换流器输出电压的幅值和相位来间接控制输出电流。经比较发现，直接电流控制要想达到理想的控制效果，就需很高的开关频率，造成开关损耗增大，因此不适于在大功率变流系统中应用，因此本实用新型采用基于前馈解耦的间接电流控制方法。 The current control methods of STATCOM system can be divided into direct and indirect control. Direct control is to directly control the output current of the converter, mainly including hysteresis control and triangular wave comparison control; indirect control is to equate the STATCOM to a controllable AC voltage source, and adjust the amplitude of the converter output voltage and phase to indirectly control the output current. After comparison, it is found that if the direct current control wants to achieve the ideal control effect, it needs a very high switching frequency, resulting in increased switching loss, so it is not suitable for application in high-power converter systems. Therefore, the utility model adopts a feed-forward solution based on Coupled indirect current control method.

通过对系统的等效电路分析，将整个变流器的损耗等效为固定电阻R，连接电抗器及线路电感等效为电感L，换流器输出电压为多电平阶梯波，谐波含量小，故可忽略谐波而只考虑其基波分量；认为系统三相对称，交流输出电压与电容电压成线性关系。对于星形接法，在abc坐标系下，由基尔霍夫电压电流定律可得： Through the equivalent circuit analysis of the system, the loss of the entire converter is equivalent to a fixed resistance R, the connecting reactor and line inductance are equivalent to an inductance L, the output voltage of the converter is a multi-level ladder wave, and the harmonic content Small, so the harmonics can be ignored and only the fundamental wave component is considered; the three-phase system is considered to be symmetrical, and the AC output voltage has a linear relationship with the capacitor voltage. For the star connection, in the abc coordinate system, it can be obtained from Kirchhoff's voltage and current law:

$L L \frac{d d}{d d t t} [\begin{matrix} {i i}_{a a} \\ {i i}_{b b} \\ {i i}_{c c} \end{matrix}] = = [\begin{matrix} {v v}_{s the s a a} \\ {v v}_{s the s b b} \\ {v v}_{s the s c c} \end{matrix}] - - [\begin{matrix} {v v}_{c c a a} \\ {v v}_{c c b b} \\ {v v}_{c c c c} \end{matrix}] - - R R [\begin{matrix} {i i}_{a a} \\ {i i}_{b b} \\ {i i}_{c c} \end{matrix}] - - - - - - ((2020))$

直流侧电容瞬时功率为STATCOM直流与交流两侧能量守恒关系，可得到： The instantaneous power of the DC side capacitor is The energy conservation relationship on both sides of STATCOM DC and AC can be obtained:

${NCu NCu}_{d d c c} \frac{{du du}_{d d c c}}{d d t t} = = {v v}_{c c a a} {i i}_{a a} + + {v v}_{c c b b} {i i}_{b b} + + {v v}_{c c c c} {i i}_{c c} - - - - - - ((21 twenty one))$

式中N为直流侧总的电容数目，又由于i_a、i_b、i_c是STATCOM某时刻发出的电流，v_sa,v_sb,v_sc为某时刻电网电压值，v_ca，v_cb，v_cc为STATCOM某时刻发出电压值，C为电容值，u_dc为电容上的电压，且： In the formula, N is the total number of capacitors on the DC side, and since _{ia, i b} _, and _ic are the currents sent by STATCOM at a certain moment, _{vsa, vsb, and vsc} _are _the grid voltage values at a certain moment, _vca , _vcb , v _cc is the voltage value sent by STATCOM at a certain moment, C is the capacitance value, u _dc is the voltage on the capacitance, and:

$\{\begin{matrix} {v v}_{s the s a a} = = \sqrt{22} {U u}_{s the s} s the s i i n no ((ω ω t t)) \\ {v v}_{s the s b b} = = \sqrt{22} {U u}_{s the s} s the s i i n no ((ω ω t t - - \frac{22}{33} π π)) \\ {v v}_{s the s c c} = = \sqrt{22} {U u}_{s the s} s the s i i n no ((ω ω t t + + \frac{22}{33} π π)) \end{matrix} - - - - - - ((22 twenty two))$

$\{\begin{matrix} {v v}_{c c a a} = = {Mu Mu}_{d d c c} s the s i i n no ((ω ω t t - - δ δ)) \\ {v v}_{c c b b} = = {Mu Mu}_{d d c c} sin sin ((ω ω t t - - \frac{22}{33} π π - - δ δ)) \\ {v v}_{c c c c} = = {Mu Mu}_{d d c c} s the s i i n no ((ω ω t t + + \frac{22}{33} π π - - δ δ)) \end{matrix} - - - - - - ((23 twenty three))$

由(20)至(23)得在abc坐标下，计算出STATCOM的数学模型为： From (20) to (23), under the abc coordinates, the mathematical model of STATCOM is calculated as:

$\{\begin{matrix} \frac{{di di}_{a a}}{d d t t} = = \sqrt{22} {U u}_{S S} s the s i i n no ((ω ω t t)) - - {Mu Mu}_{d d c c} s the s i i n no ((ω ω t t - - δ δ)) - - {Ri Ri}_{a a} \\ \frac{{di di}_{b b}}{d d t t} = = \sqrt{22} {U u}_{S S} s the s i i n no ((ω ω t t - - \frac{22}{33} π π)) - - {Mu Mu}_{d d c c} s the s i i n no ((ω ω t t - - \frac{22}{33} π π - - δ δ)) - - {Ri Ri}_{b b} \\ \frac{{di di}_{c c}}{d d t t} = = \sqrt{22} {U u}_{S S} s the s i i n no ((ω ω t t + + \frac{22}{33} π π)) - - {Mu Mu}_{d d c c} s the s i i n no ((ω ω t t + + \frac{22}{33} π π - - δ δ)) - - {Ri Ri}_{c c} \\ \frac{{du du}_{d d c c}}{d d t t} = = \frac{M m}{N N C C} [[{i i}_{a a} s the s i i n no ((ω ω t t - - δ δ)) + + {i i}_{b b} s the s i i n no ((ω ω t t - - \frac{22}{33} π π - - δ δ)) + + {i i}_{c c} s the s i i n no ((ω ω t t + + \frac{22}{33} π π - - δ δ))]] \end{matrix} - - - - - - ((24 twenty four))$

引入dq变换，式(24)在旋转坐标系下变为： Introducing the dq transformation, formula (24) becomes:

$\begin{matrix} \frac{d d}{d d t t} [\begin{matrix} {i i}_{d d} \\ {i i}_{q q} \end{matrix}] = = \frac{d d}{d d t t} (({T T}_{a a b b c c - - d d q q} [\begin{matrix} {i i}_{a a} \\ {i i}_{b b} \\ {i i}_{c c} \end{matrix}])) = = {T T}_{a a b b c c - - d d q q} \frac{d d}{d d t t} [\begin{matrix} {i i}_{a a} \\ {i i}_{b b} \\ {i i}_{c c} \end{matrix}] + + \frac{{dT dT}_{a a b b c c - - d d q q}}{d d t t} [\begin{matrix} {i i}_{a a} \\ {i i}_{b b} \\ {i i}_{c c} \end{matrix}] \\ = = {T T}_{a a b b c c - - d d q q} \frac{d d}{d d t t} [\begin{matrix} {i i}_{a a} \\ {i i}_{b b} \\ {i i}_{c c} \end{matrix}] + + ω ω [\begin{matrix} {i i}_{d d} \\ {i i}_{q q} \end{matrix}] = = [\begin{matrix} - - \frac{R R}{L L} & ω ω \\ - - ω ω & - - \frac{R R}{L L} \end{matrix}] [\begin{matrix} {i i}_{d d} \\ {i i}_{q q} \end{matrix}] + + \frac{11}{L L} [\begin{matrix} {v v}_{s the s d d} - - {v v}_{c c q q} \\ {v v}_{s the s q q} - - {v v}_{c c q q} \end{matrix}] \end{matrix} - - - - - - ((2525))$

式中，v_sd，v_sq，v_cd，v_cq分别为电网电压和STATCOM输出电压的dq分量，i_d，i_q为STATCOM输出电流的dq分量。由于U_S与d轴重合，因此有下式成立： In the formula, v _sd , v _sq , v _cd , v _cq are the dq components of grid voltage and STATCOM output voltage, respectively, and i _d , i _q are the dq components of STATCOM output current. Since U _S coincides with the d axis, the following formula holds:

$[\begin{matrix} {v v}_{s the s d d} \\ {v v}_{s the s q q} \end{matrix}] = = \sqrt{33} {U u}_{s the s} [\begin{matrix} 11 \\ 00 \end{matrix}],, [\begin{matrix} {v v}_{c c d d} \\ {v v}_{c c q q} \end{matrix}] = = \frac{\sqrt{33}}{22} {Mu Mu}_{d d c c} [\begin{matrix} 11 \\ 00 \end{matrix}] [\begin{matrix} c c o o s the s δ δ \\ s the s i i n no δ δ \end{matrix}] - - - - - - ((2626))$

其中：δ为STATCOM输出电压与电网电压的相位差，M为调制比，U_S为电网电压。选择控制量为STATCOM的输出电压的dq轴分量v_cd和v_cq，由于δ＝tg^-1(v_cd/v_cq)，通过对M和δ的控制就会改变STATCOM与电网的功率交换，从而补偿系统无功。根据式(21)可以得到STATCOM输出电压v_cd，v_cq的表达式： Among them: δ is the phase difference between the STATCOM output voltage and the grid voltage, M is the modulation ratio, and U _S is the grid voltage. Select the control quantity as the dq axis components v _cd and v _cq of the output voltage of STATCOM, because δ=tg ^-1 (v _cd /v _cq ), through the control of M and δ, the power exchange between STATCOM and the grid will be changed, thereby compensating the reactive power of the system. According to formula (21), the expressions of STATCOM output voltage v _cd and v _cq can be obtained:

$[\begin{matrix} {v v}_{c c d d} \\ {v v}_{c c q q} \end{matrix}] = = [\begin{matrix} {v v}_{s the s d d} \\ {v v}_{s the s q q} \end{matrix}] + + [\begin{matrix} - - R R & ω ω L L \\ - - ω ω L L & - - R R \end{matrix}] [\begin{matrix} {i i}_{d d} \\ {i i}_{q q} \end{matrix}] - - L L \frac{d d}{d d t t} [\begin{matrix} {i i}_{d d} \\ {i i}_{q q} \end{matrix}] - - - - - - ((2727))$

根据式(27)得到有功、无功电流控制框图如图10所示。可以明显看出，STATCOM系统是一个典型的耦合系统，i_d，i_q通过电抗器耦合，STATCOM输出电压的变化会影响到输出电流的变化，并且dq轴相互影响，不利于控制。通过采取一定措施对dq轴解耦，可以使得控制更为简单，电流变换到dq轴后成为直流量，通过传统线性PI调节即可实现无静差调节。 According to formula (27), the active and reactive current control block diagram is shown in Figure 10. It can be clearly seen that the STATCOM system is a typical coupling system, i _d and i _q are coupled through a reactor, the change of the STATCOM output voltage will affect the change of the output current, and the dq axes affect each other, which is not conducive to control. By taking certain measures to decouple the dq axis, the control can be made simpler. After the current is converted to the dq axis, it becomes a DC flow, and no static difference adjustment can be realized through the traditional linear PI adjustment.

前馈解耦控制策略如下，引入中间变量x₁，x₂： The feed-forward decoupling control strategy is as follows, introducing intermediate variables x ₁ and x ₂ :

$\{\begin{matrix} {x x}_{11} = = {v v}_{s the s d d} - - {v v}_{c c d d} + + {ωLi ω Li}_{q q} \\ {x x}_{22} = = {v v}_{s the s q q} - - {v v}_{c c q q} + + {ωLi ω Li}_{d d} \end{matrix} - - - - - - ((2828))$

由(27)和(28)可得： From (27) and (28), we can get:

$\frac{d d}{d d t t} [\begin{matrix} {i i}_{d d} \\ {i i}_{q q} \end{matrix}] = = [\begin{matrix} - - R R / / L L & 00 \\ 00 & - - R R / / L L \end{matrix}] - - - - - - ((2929))$

$[\begin{matrix} {x x}_{11} \\ {x x}_{22} \end{matrix}] = = [\begin{matrix} {K K}_{11} (({i i}_{d d}^{* *} - - {i i}_{d d})) + + \frac{{K K}_{11}}{{T T}_{11}} &Integral; &Integral; (({i i}_{d d}^{* *} - - {i i}_{d d})) d d t t \\ {K K}_{22} (({i i}_{q q}^{* *} - - {i i}_{q q})) + + \frac{{K K}_{22}}{{T T}_{22}} &Integral; &Integral; (({i i}_{q q}^{* *} - - {i i}_{q q})) d d t t \end{matrix}] - - - - - - ((3030))$

其中，K₁、K₂为比例系数，T₁，T₂为积分系数，i_d和i_q分别为MMC换流器输出的dq坐标轴的电流分量，i_d ^*和i_q ^*分别为负载中的有功电流分量和无功电流分量的给定值； Among them, K ₁ and K ₂ are proportional coefficients, T ₁ and T ₂ are integral coefficients, i _d and i _q are the current components of the dq coordinate axes output by the MMC converter, and i _d ^* and i _q ^* are the load The given value of active current component and reactive current component in

如图11所示，通过这种变换将dq轴的电流设计成两个PI控制器，其输出就是中间变量x₁，x₂，这样就可实现dq轴电流的解耦控制。 As shown in Figure 11, through this transformation, the dq-axis current is designed as two PI controllers, whose outputs are intermediate variables x ₁ and x ₂ , so that the decoupling control of the dq-axis current can be realized.

所述电压电流双闭环控制如图12所示，STATCOM系统由电压外环和电流内环构成，其中，电压外环是控制直流侧的电容电压，使其保持恒值，给定直流电压U_ref与换流器直流侧电容电压进行比较，经电压PI调节，其输出作为有功电流给定值i_d ^*,负载电流经坐标变换、取反处理得到无功电流给定值i_q ^*；将变换器输出的三相电流i_abc经坐标变换得到i_d和i_q，与i_d ^*、i_q ^*进行比较，经电流环PI调节，得到期望的输出电压V_cd、V_cq；对V_cd、V_cq进行坐标逆变换得到静止坐标系下的三相调制波，与载波移相控制器产生的三角载波比较后得到PWM控制信号，经功放驱动MMC模块内相应的功率开关管，从而控制STATCOM输出电压的幅值和相位，达到补偿无功的目的。 The voltage and current double closed-loop control is shown in Figure 12. The STATCOM system is composed of a voltage outer loop and a current inner loop, wherein the voltage outer loop controls the capacitor voltage on the DC side to maintain a constant value, and the given DC voltage U _ref Compared with the capacitor voltage on the DC side of the converter, adjusted by the voltage PI, its output is taken as the given value of active current i _d ^* , and the given value of reactive current i _q ^* is obtained by transforming the load current through coordinate transformation and negation; The three-phase current i _abc output by the device is transformed into i d and i q to obtain i _d and i _q , compared with i _d ^* and i _q ^* , and adjusted by the current loop PI to obtain the expected output voltage V _cd and V _cq ; for V _cd , V _cq performs coordinate inverse transformation to obtain the three-phase modulation wave in the static coordinate system, and compares it with the triangular carrier wave generated by the carrier phase shift controller to obtain the PWM control signal, which drives the corresponding power switch tube in the MMC module through the power amplifier to control the STATCOM output The amplitude and phase of the voltage can achieve the purpose of reactive power compensation.

所述全桥模块电容电压均衡过程包括：每一相上、下桥臂全桥模块电容电压的给定值V_H,ref与全桥模块的实际电容电压进行比较，经PI控制器，其输出乘以该桥臂电流的符号函数后，将生成的全桥模块电压给定值V_r,ref与三角载波比较后得到PWM波，其中，r＝P,N，驱动全桥模块中相应的功率开关管，对全桥模块的电容进行充放电控制，实现全桥模块电容电压的均衡； The full-bridge module capacitor voltage equalization process includes: the given value V _H,ref of the full-bridge module capacitor voltage of each phase upper and lower bridge arms is compared with the actual capacitor voltage of the full-bridge module, and through the PI controller, its output After multiplying by the sign function of the bridge arm current, compare the generated full-bridge module voltage given value V _r,ref with the triangular carrier wave to obtain a PWM wave, where r=P,N, drive the corresponding power in the full-bridge module The switch tube controls the charging and discharging of the capacitors of the full-bridge module to realize the balance of the capacitor voltage of the full-bridge module;

所述环流抑制过程包括：将每一相环流分别与环流的参考值进行比较，此时环流的参考值i_cir,ref＝i_dc/3，得到的结果通过一个比例控制器形成一个全桥模块的电压调整信号，将这个电压调整信号平均分成2份，分别加在该相上、下桥臂全桥模块的电压信号上。 The circulation suppression process includes: comparing the circulation current of each phase with the reference value of the circulation current, at this time, the reference value of the circulation current i _cir,ref = i _dc /3, and the obtained result forms a full-bridge module through a proportional controller Divide the voltage adjustment signal into two parts on average, and add them to the voltage signals of the upper and lower bridge arm full-bridge modules of the phase respectively.

本实用新型通过主程序、捕获中断子程序以及T1周期中断子程序的设计来实现主控制器DSP的程序设计。 The utility model realizes the program design of the main controller DSP through the design of the main program, the capture interrupt subroutine and the T1 period interrupt subroutine.

STATCOM系统软件的整体规划是通过主程序设计来完成的，其主要对DSP系统的工作环境进行配置、系统中相关变量的初始化、各中断的初始化、判断是否开启中断子程序等，接着进入接收和发送数据的循环中，同时等待中断事件的发生。当中断被开启，暂时停止主循环，进入到相应的中断服务子程序中进行各种运算和配置PWM控制信号。当中断完成后，返回主循环，继续等待下一次中断的发生。主程序流程图如图18所示。 The overall planning of the STATCOM system software is completed through the design of the main program, which mainly configures the working environment of the DSP system, initializes related variables in the system, initializes each interrupt, judges whether to open the interrupt subroutine, etc., and then enters the receiving and In the cycle of sending data, wait for the occurrence of interrupt event at the same time. When the interrupt is enabled, temporarily stop the main loop, enter the corresponding interrupt service subroutine to perform various calculations and configure PWM control signals. When the interrupt is completed, return to the main loop and continue to wait for the next interrupt to occur. The main program flow chart is shown in Figure 18.

捕获中断子程序的设计是为了实现数字锁相环，以检测电网的频率。捕获中断子程序的开启则是通过a相电压信号的过零点产生的上升沿进行触发。值得注意的是电网的频率并不是一成不变的50Hz，而是在一个小范围内波动，因此需要进行一个限定判断，具体的实现方法如图19所示。 The capture interrupt subroutine is designed to implement a digital phase-locked loop to detect the frequency of the power grid. The opening of the capture interrupt subroutine is triggered by the rising edge generated by the zero-crossing point of the a-phase voltage signal. It is worth noting that the frequency of the power grid is not a constant 50Hz, but fluctuates within a small range, so a limited judgment is required. The specific implementation method is shown in Figure 19.

T1中断子程序的流程图如图20所示，在该子程序里要完成电压电流的采样、桥臂电流极性的判断、与FPGA的通讯、子模块电压保护、有功无功的计算以及三相调制波的计算等，DSP模块的主要算法都在该子程序中完成。 The flowchart of the T1 interrupt subroutine is shown in Figure 20. In this subroutine, the sampling of voltage and current, the judgment of the polarity of the bridge arm current, the communication with FPGA, the voltage protection of sub-modules, the calculation of active and reactive power, and the three The calculation of the phase modulation wave, etc., and the main algorithms of the DSP module are all completed in this subroutine.

验证过程： Verification process:

为验证系统无功补偿效果，图21是电网补偿之前A相电压、电流波形，可以看出，补偿前电流明显滞后于电压；图22是补偿之后A相电压和电流波形，相电压、电流相位一致。可见，本实用新型对无功有很好的补偿效果。 In order to verify the reactive power compensation effect of the system, Figure 21 is the voltage and current waveforms of phase A before grid compensation. It can be seen that the current before compensation lags behind the voltage obviously; Figure 22 is the voltage and current waveforms of phase A after compensation, and the phase voltage and current phase unanimous. It can be seen that the utility model has a good compensation effect on reactive power.

为验证子模块电容电压均衡效果，a相上、下桥臂中半桥子模块的电容电压如图25所示，可以看到，上、下桥臂子模块电容电压基本稳定在1000V，波动在10V左右。可见，本实用新型具有较好的子模块均压能力。 In order to verify the equalization effect of the capacitor voltage of the sub-modules, the capacitor voltages of the half-bridge sub-modules in the upper and lower bridge arms of phase a are shown in Figure 25. It can be seen that the capacitor voltages of the upper and lower bridge arm sub-modules are basically stable at 1000V, and fluctuate between Around 10V. It can be seen that the utility model has better sub-module pressure equalization capability.

为验证系统对环流的抑制能力，图26给出了采用环流抑制器前后三相环流波形，图27为采用环流抑制器后的三相环流波形，可以看出三相环流得到了有效抑制。可见，本实用新型具有较好的抑制环流能力。 In order to verify the ability of the system to suppress the circulating current, Figure 26 shows the three-phase circulating current waveform before and after using the circulating current suppressor, and Figure 27 shows the three-phase circulating current waveform after using the circulating current suppressor. It can be seen that the three-phase circulating current has been effectively suppressed. It can be seen that the utility model has better ability to suppress circulation.

Claims

1. based on the reactive power compensator of novel modularized many level topological structure, comprise three-phase alternating-current supply, resistance sense load, converter, control circuit, signal deteching circuit and drive circuit, described converter comprises three identical brachium pontis be connected in parallel of structure, each brachium pontis comprises about point symmetry in brachium pontis and the upper brachium pontis be connected in series and lower brachium pontis, it is characterized in that: described upper brachium pontis comprises the inductance of series connection mutually, some half-bridge cells and a H-bridge unit, the inductance of upper brachium pontis and the inductance of lower brachium pontis are connected in series, the mid point of converter three brachium pontis is connected between three-phase alternating-current supply and resistance sense load by conductor in parallel, the input of signal deteching circuit connects the output of three-phase alternating-current supply respectively, the input of resistance sense load, the output of converter, converter three brachium pontis, each half-bridge cells of converter and H-bridge unit, the input of the output connection control circuit of signal deteching circuit, the output of control circuit is connected by drive circuit and converter.

2. according to claim 1 based on the reactive power compensator of novel modularized many level topological structure, it is characterized in that: described control circuit comprises phase-shifting carrier wave controller, first control unit and the second control unit, the output of the first control unit and the output of the second control unit all connect with phase-shifting carrier wave controller, described first control unit comprises the first comparator (1), second comparator (3), 3rd comparator (5), 4th comparator (7), 5th comparator (9), one PI controller (2), 2nd PI controller (4), 3rd PI controller (8), first coordinate converter (6), second coordinate converter (12), first reactor (10) and the second reactor (11), first comparator (1), one PI controller (2), second comparator (3), 2nd PI controller (4) and the 3rd comparator (5) access the first coordinate converter after connecting successively, 4th comparator (7), 3rd PI controller (8) and the 5th comparator (9) access the first coordinate converter (6) after connecting successively, the output of the first coordinate converter (6) connects phase-shifting carrier wave controller, the output of phase-shifting carrier wave controller connects drive circuit, the output of described converter connects the second coordinate converter, first output of the second coordinate converter connects the second comparator (3) and the second reactor (11) respectively, the output of the second reactor (11) connects the 3rd comparator (5), another output of second coordinate converter connects the 4th comparator (7) and the first reactor (10) respectively, the output of the first reactor (10) connects the 5th comparator (9).

3. according to claim 2 based on the reactive power compensator of novel modularized many level topological structure, it is characterized in that: described second control unit comprises the 6th comparator (13), 7th comparator (15), 8th comparator (18), 9th comparator (21), first proportional controller (14), second proportional controller (17), 4th PI controller (16), 5th PI controller (19) sum functions module (20), 6th comparator (13), first proportional controller (14), 7th comparator (15), 4th PI controller (16) and the second proportional controller (17) access the 9th comparator (21) after connecting successively, 8th comparator (18), 5th PI controller (19) sum functions module (20) accesses the 9th comparator (21) after connecting successively, the output of the 9th comparator (21) connects described phase-shifting carrier wave controller.

4. according to claim 2 based on the reactive power compensator of novel modularized many level topological structure, it is characterized in that: described control circuit comprises DSP module and FPGA module, the output of DSP module connects FPGA module, the output of FPGA module connects drive circuit, described first control unit and the second control unit are integrated in DSP module, and described phase-shifting carrier wave controller is integrated in FPGA module.

5. according to claim 1 based on the reactive power compensator of novel modularized many level topological structure, it is characterized in that: the described reactive power compensator based on novel modularized many level topological structure comprises zero cross detection circuit, described zero cross detection circuit comprises voltage sensor (22), comparison circuit (23) and optocoupler (24), the input of voltage sensor (22) connects the output of three-phase alternating-current supply, the output of voltage sensor (22) connects the input of comparison circuit (23), the output of comparison circuit (23) accesses DSP module afterwards by optocoupler (24).

6. according to claim 1 based on the reactive power compensator of novel modularized many level topological structure, it is characterized in that: described drive circuit comprises level shifting circuit and driving chip, the input of level shifting circuit is the input of drive circuit, the output of level shifting circuit connects driving chip, and the output of driving chip is the output of drive circuit.