CN1716758A - Control device and module of permanent magnet synchronous motor - Google Patents

Abstract

The present invention provides a field-weakening vector control device of a permanent magnet synchronous motor, which can realize high-precision and high-response motor torque even in a field-weakening region; in addition, in a system for low-cost current detection, and omits Commonly used in magnetic pole position detector systems. The method for realizing this is to automatically correct the control gain of the calculation unit that generates the d-axis current command by using the frequency command value of the motor. In addition, the d-axis current command generated at no load is obtained by calculation in advance, and is added to the output of the calculation unit that generates the d-axis current command value.

Description

Control device and module of permanent magnet synchronous motor

technical field

The invention relates to a vector control method in a weak magnetic field area of a permanent magnet synchronous motor.

Background technique

As the prior art of the vector control method in the weak magnetic field area, the following two methods are known, that is, as described in Japanese Patent Laid-Open No. 8-182398, the d-axis current command value is listed (table), and the d-axis and q A method in which the current control of the axis is a proportional calculation method; and, as described in JP-A-2002-95300, the terminal voltage of the motor is obtained by the current control section of the d-axis and the q-axis, and the command value of the terminal voltage is obtained by A method of calculating the above-mentioned d-axis current command value by using a proportional-integral operation from the deviation from the above-mentioned terminal voltage.

[Patent Document 1] JP-A-8-182398

[Patent Document 2] JP-A-2002-95300

However, in the method described in Japanese Patent Laid-Open No. 8-182398, the current control is a proportional operation method, so there is no current that complies with the current command value, and the torque accuracy tends to deteriorate; In the above method, since the generation of the d-axis current command is slow, the torque response tends to deteriorate.

An object of the present invention is to provide a "field weakening control device for a permanent magnet synchronous motor" capable of realizing "high precision and high response torque control" even in a field weakening control region.

Contents of the invention

In the present invention, the d-axis current command value can be generated with high response by automatically correcting the integral control gain of the field weakening command calculation unit that generates the d-axis current command value using the frequency command value of the motor.

Furthermore, it is characterized in that a d-axis current command generated at no load is obtained by calculation in advance, and is added to an output of a calculation unit that generates a d-axis current command value.

With the present invention, even in a weak magnetic field region, high-precision, high-response motor torque can be realized.

Description of drawings

FIG. 1 is a block diagram showing a field weakening vector control device for a permanent magnet synchronous motor according to an embodiment of the present invention.

FIG. 2 is an explanatory diagram of a field weakening command calculation unit 8 in the control device of FIG. 1 .

FIG. 3 is an example of a voltage saturation characteristic diagram in the case where there is no field weakening command calculation unit 8 .

FIG. 4 is an example of a voltage saturation characteristic diagram when the field weakening command calculation unit 8 is added.

5 is a configuration diagram showing a field weakening vector control device for a permanent magnet synchronous motor according to another embodiment of the present invention.

FIG. 6 is an explanatory diagram of a field weakening command calculation unit 8 a in the control device of FIG. 5 .

7 is a configuration diagram showing a field weakening vector control device for a permanent magnet synchronous motor according to another embodiment of the present invention.

FIG. 8 is an explanatory diagram of a field weakening command calculation unit 8 b in the control device of FIG. 7 .

9 is a configuration diagram showing a field weakening vector control device for a permanent magnet synchronous motor according to another embodiment of the present invention.

FIG. 10 is an explanatory diagram of a field weakening command calculation unit 8 c in the control device of FIG. 9 .

Fig. 11 is a configuration diagram showing a field weakening vector control device for a permanent magnet synchronous motor according to another embodiment of the present invention.

Fig. 12 is a configuration diagram showing a field weakening vector control device for a permanent magnet synchronous motor according to another embodiment of the present invention.

Fig. 13 is a configuration diagram in the case of applying an embodiment of the present invention to a module.

In the picture:

1-Permanent magnet synchronous motor, 2-Power converter, 3-Current detector, 4-Magnetic pole position detector, 5-Frequency calculation unit, 6-Phase calculation unit, 7, 13-Coordinate transformation unit, 8, 8a, 8b, 8c-weakening magnetic field command calculation unit, 9-d-axis current command calculation unit, 10-q-axis current command calculation unit, 11-voltage vector calculation unit, 12-output voltage calculation unit, 14-current estimation unit, 15- Phase error calculation unit, 21-DC power supply, IDC-input DC bus current detection value, Id ^* -first d-axis current command value, Id ^** -second d-axis current command value, Iq ^* -first q-axis current command value, Iq ^** - second q-axis current command value, V ₁ ^* _ref - output voltage command value in the field weakening region, V ₁ ^* - output voltage value, θc ^* - rotation phase command, ω ₁ ^* - frequency instruction.

Detailed ways

Hereinafter, embodiments of the present invention will be described in detail using the drawings.

FIG. 1 shows a configuration example of a field weakening vector control device for a permanent magnet synchronous motor as an embodiment of the present invention. 1 is a permanent magnet synchronous motor; 2 is a power converter that outputs a voltage proportional to the three-phase AC voltage command value Vu ^* , Vv ^* , Vw ^* ; 21 is a DC power supply; 3 is a three-phase AC current Iu that can be detected , Iv, and Iw current detectors; 4 is a magnetic pole position detector capable of detecting the position detection value θi of every 60° electric angle of the motor; 5 is a frequency calculation unit that calculates the frequency command value ω ₁ ^* according to the position detection value θi ; 6 is a phase calculation unit for calculating the rotation phase command θc ^* of the motor based on the position detection value θi and the frequency command value ω ₁ ^* ; 7 is the detection value Iuc, Ivc, Iwc based on the above-mentioned three-phase alternating current Iu, Iv, Iw , and the rotation phase command θc ^* , the coordinate conversion unit that outputs the current detection values Idc and Iqc of the d-axis and q-axis; 8 is based on the output voltage command value V ₁ ^* _ref and the output voltage value V ₁ ^* in the weak magnetic field Deviation, calculating the first d-axis current command value Id ^* of the field weakening command calculation unit; 9 is based on the first d-axis current command value Id ^* as the output of the weakening field command calculation unit, and the deviation of the d-axis current detection value Idc , the d-axis current command calculation unit that outputs the second d-axis current command value Id ^** ; 10 is to output the second q-axis current command according to the deviation between the first q-axis current command value Iq ^* and the q-axis current detection value Iqc The q- ^axis current command computing unit with the value Iq ^** ; 11 is to calculate the voltage command values Vd* and Vq ^* based on the electrical constant of the motor 1, the second current command values Id ^** , Iq ^** and the frequency command value ω ₁ ^* 12 is the output voltage calculation unit for calculating the output voltage value V ₁ ^* of the power converter according to the voltage command values Vd ^* , Vq ^* ; 13 is the output voltage calculation unit based on the voltage command values Vd ^* , Vq ^* and the rotation phase command θc ^* is a coordinate conversion unit that outputs three-phase AC voltage command values Vu ^* , Vv ^* , and Vw ^* .

First, the basic operation of voltage control and phase control in the vector control system when using the field weakening command calculation which is the characteristic of the present invention will be described.

For the voltage control, the output voltage calculation unit 12 in FIG. 1 calculates the output voltage value V 1 * using the ^d -axis and q-axis voltage command values Vd ^* , _Vq ^* as shown in Equation 1.

[number 1]

${{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="V"\; filenames="A20051008223800151.png,A20051008223800162.png"\; state="\{\{state\}\}">V</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\sqrt{{\mathrm{<span\; class="notranslate">\; Vd</span>}}^{{<span\; class="notranslate">\; *</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; wxya</span>}}^{{<span\; class="notranslate">\; *</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}}<span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>$

The field weakening command calculation unit 8 calculates the first d-axis current command value Id ^* so that the above-mentioned output voltage value V ₁ ^* coincides with the output voltage command value V ₁ ^* _ref in the field weakening region.

In addition, the voltage vector calculation unit 11 calculates the d-axis and q-axis voltage command values Vd ^* and Vq ^* using the second d-axis and q-axis current command values and motor constants shown in the number 2 in advance, and controls the converter The output voltage.

[number 2]

Vd ^* ＝ _R1 ^* Id ^** - _ω1 ^* ·Lq ^* ·Iq ^**

Vq ^* ＝R ₁ ^* ·Iq ^** +ω ₁ ^* ·Ld ^* ·Id ^** +ω ₁ ^* ·Ke ^* ...(2)

Here, R1 ^* is the setting value of the resistance, Ld ^* is the setting value of the d-axis inductance, Lq ^* is the setting value of the q-axis inductance, and Ke ^* is the setting value of the induced voltage constant.

On the other hand, for the phase control, the magnetic pole position detector 4 can grasp the magnetic pole position every 60 degrees electrical angle. In this embodiment, the position detection value θi at this time is

[number 3]

θi=60i+30...(3)

Wherein, i=0, 1, 2, 3, 4, 5. In the frequency calculation unit 5, an average rotation frequency ω ₁ ^* (hereinafter referred to as a frequency command value) in the shortest 60-degree interval is calculated based on the detected position value θi.

[number 4]

ω ₁ ^* ＝Δθ/Δt …(4)

Here, Δθ=θi-θ(i-1), and Δt is the time until the position detection signal in the 60-degree section is detected.

In addition, the phase calculation unit 6 calculates the rotational phase command θc ^* as shown in Equation 5 using the detected position value θi and the frequency command ω ₁ ^* to control the reference phase of the motor 1 .

[number 5]

θ ^* ＝θi+ω ₁ ^* Δt …(5)

The above is the basic operation of voltage control and phase control.

Next, the field weakening command calculation unit 8 in the feedback control system which is the characteristic of the present invention will be described with reference to FIG. 2 .

In the field weakening command calculation unit 8, the deviation between the output voltage command value V ₁ ^* _ref and the output voltage value V ₁ ^* in the field weakening region is input to the integral calculation unit 81 whose integral gain is constant K, and integral calculation is performed. This calculated value is input to the limiter calculation unit 82 whose positive side is limited to "zero", and its output value is the first d-axis current command Id ^* .

Next, through this embodiment, the effect brought by the present invention will be described.

The control device of FIG. 1 considers the case where the first d-axis current command value Id ^* is controlled to be "zero" (the case where the field weakening command calculation is not performed).

Substituting the number 2 into the number 1 from the V ₁ ^* output by the voltage vector calculation unit 11:

[number 6]

${{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="V"\; filenames="A20051008223800151.png,A20051008223800162.png"\; state="\{\{state\}\}">V</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\sqrt{{<span\; class="notranslate">\; (</span>{{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="R"\; filenames="A20051008223800151.png,A20051008223800162.png"\; state="\{\{state\}\}">R</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}{\mathrm{<span\; class="notranslate">\; ID</span>}}^{<span\; class="notranslate">\; *</span><span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; -</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; Q</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>{\mathrm{<span\; class="notranslate">\; Iq</span>}}^{<span\; class="notranslate">\; *</span><span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; +</span>{<span\; class="notranslate">\; (</span>{{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="R"\; filenames="A20051008223800151.png,A20051008223800162.png"\; state="\{\{state\}\}">R</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; Iq</span>}}^{<span\; class="notranslate">\; *</span><span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; +</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; Ld</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; ID</span>}}^{<span\; class="notranslate">\; *</span><span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; +</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>{\mathrm{<span\; class="notranslate">\; Ke</span>}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; )</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 6</span>}<span\; class="notranslate">\; )</span>$

In addition, if the saturation value of V ₁ ^* is assumed to be V ₁ ^* _max , the voltage saturation region forms a relationship of number 7.

[number 7]

V ₁ ^* _max ² ＝(R1 ^* Id ^** -ω ₁ ^* Lq ^* Iq ^** ) ² +(R1 ^* Iq ^** +ω ₁ ^* Ld ^* Id ^** +ω ₁ ^* Ke ^* ) ² …(7)

Here, by rearranging the number 7, the quadratic equation about the frequency command ω ₁ ^* can be obtained,

[number 8]

$\mathrm{<span\; class="notranslate">\; A</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{{<span\; class="notranslate">\; *</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; B</span>}<span\; class="notranslate">\; \&CenterDot;</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; +</span>\mathrm{<span\; class="notranslate">\; C</span>}<span\; class="notranslate">\; =</span>\mathrm{<span\; class="notranslate">\; 0</span>}<span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 8</span>}<span\; class="notranslate">\; )</span>$

in,

A＝(Ld ^* Id ^** ) ² +(Lq ^* Iq ^** ) ² +(Ke ^* +2 Ld ^* Id ^** )

B＝2·R1 ^* ·Iq ^** ·(Ke ^* +(Ld ^* -Iq ^** )·Id ^** )

$\mathrm{<span\; class="notranslate">\; C</span>}<span\; class="notranslate">\; =</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="R"\; filenames="A20051008223800151.png,A20051008223800162.png"\; state="\{\{state\}\}">R</figure-callout></span>}\mathrm{<span\; class="notranslate">\; 1</span>}}^{{<span\; class="notranslate">\; *</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>{\mathrm{<span\; class="notranslate">\; ID</span>}}^{{<span\; class="notranslate">\; *</span><span\; class="notranslate">\; *</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; +</span>{\mathrm{<span\; class="notranslate">\; Iq</span>}}^{{<span\; class="notranslate">\; *</span><span\; class="notranslate">\; *</span>}^{\mathrm{<span\; class="notranslate">\; 2</span>}}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; -</span>{{{{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="V"\; filenames="A20051008223800151.png,A20051008223800162.png"\; state="\{\{state\}\}">V</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}}_{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="max"\; filenames="A20051008223800151.png,A20051008223800181.png"\; state="\{\{state\}\}">max</figure-callout></span>}}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}$

According to number 8, ω ₁ ^* when V ₁ ^* is saturated can be obtained.

[Number 9]

${{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; =</span>\frac{<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; B</span>}<span\; class="notranslate">\; \&PlusMinus;</span>\sqrt{{\mathrm{<span\; class="notranslate">\; B</span>}}^{\mathrm{<span\; class="notranslate">\; 2</span>}}<span\; class="notranslate">\; -</span>\mathrm{<span\; class="notranslate">\; 4</span>}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; A</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="C"\; filenames="A20051008223800151.png,A20051008223800181.png"\; state="\{\{state\}\}">C</figure-callout></span>}}}{\mathrm{<span\; class="notranslate">\; 2</span>}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; A</span>}}<span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 9</span>}<span\; class="notranslate">\; )</span>$

Here, when Id ^** =Id ^* =0 and Iq ^** =τ/KT, the relationship between the motor torque τ and the frequency command ω ₁ ^* is shown in FIG. 3 .

Here, τ is the motor torque, and KT is the torque coefficient.

The solid line shown in FIG. 3 is the boundary line of V ₁ ^* saturation, the upper side of the boundary line is the saturated area, and the lower side is the unsaturated area, which is the practically operable range.

Therefore, there is a problem in the vector control in which the d-axis current command value Id ^* is set to "zero" that the operating range in the high-speed region is limited to be low.

Therefore, in this embodiment, the first d-axis current command value Id* is calculated in such a manner that the output voltage value V ₁ ^* coincides with the output voltage command value V ₁ ^* _ref in the field weakening region, and the ^{Id *} A second d-axis current command value Id ^** is generated, and a voltage vector calculation is performed.

Here, the output voltage command value V ₁ ^* _ref in the field-weakening region is set as the expression 10.

[number 10]

V ₁ ^* _ref < V ₁ ^* _max ...(10)

As a result, since the voltage command values Vd ^* and Vq* are calculated in such a manner that the output voltage value _V1 ^* does not saturate (be a value smaller than _V1 ^* _max ⁾ by the voltage vector calculation unit 11, the voltage in the high-speed region can be enlarged. operating range.

Since the present invention can generate current according to the current command value, high-precision torque control can be realized, and in addition, the operating range can be expanded as shown in FIG. 4 .

In addition, if high torque is required during torque control operation, it is necessary to flow a large current corresponding to the torque. If high torque is required for a continuous time, it will generate heat due to the motor current, and the winding resistance R inside the motor will increase with time. In this way, since the resistance set value calculated by the voltage vector calculation unit does not match the actual resistance value, the necessary voltage cannot be supplied to the motor.

Therefore, by providing a current command calculation unit upstream of the vector calculation unit as shown in FIG. 1 of this embodiment, and controlling the output voltage so that the motor current coincides with the current command value, it is possible to avoid changes in the motor constant. , Hall elements, etc., to provide a control device for an AC motor that does not suffer from insufficient torque from a low speed range.

[Example 2]

Fig. 5 shows another embodiment of the present invention.

This embodiment is a control device for a permanent magnet synchronous motor in which the integral gain of the field weakening command calculation unit in the feedback control mode is changed by using the frequency command ω ₁ ^* .

In Fig. 5, 1-7, 9-13, 21 are the same as those in Fig. 1 . 8a is a field weakening command calculation unit, which automatically corrects the integral gain when integrating the deviation between V ₁ ^* _ref and V ₁ ^* according to the frequency command ω ₁ ^* .

Next, using FIG. 6 , the field weakening command computing unit 8 a that is a feature of the present invention will be described.

In the field weakening command calculation unit 8a, the deviation between the output voltage command value V ₁ ^* _ref and the output voltage value V ₁ ^* in the field weakening region is input to the integral calculation unit 8a1 whose integral gain is constant K, and integral calculation is performed. At this time, the integral gain K is automatically corrected by the frequency _ω1 . The output value of the integral calculation unit 8a1 is input to the limiter calculation unit 82a that limits the positive side to "zero", and the output value thereof is the first d-axis current command Id ^* .

Using this current command value Id ^* , a second current command value Id ^** is generated, voltage command values Vd ^* , Vq ^* are calculated, and the converter output voltage is controlled.

Here, the effects brought about by the present invention will be described through this embodiment.

When the integral gain K used in the operation of the field weakening command is constant, the closed-loop transfer function G _Ф (s) from V ₁ ^* _ref at no load (Iq ^* = 0) to Id ^* is

[number 11]

${\mathrm{<span\; class="notranslate">\; G</span>}}_{\mathrm{<span\; class="notranslate">\; \&Phi;</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; Ld</span>}}^{<span\; class="notranslate">\; *</span>}}}{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span><span\; class="notranslate">\; (</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; K</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&Center\; Dot;</span>{\mathrm{<span\; class="notranslate">\; Ld</span>}}^{<span\; class="notranslate">\; *</span>}}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; the\; s</span>}}<span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 11</span>}<span\; class="notranslate">\; )</span>$

Here, s is the Laplacian operator. According to number 11, Id ^* is generated by a delay, and its response time constant T _Ф is number 12. It can be seen that T _Ф changes according to the frequency command ω ₁ ^* .

[number 12]

${\mathrm{<span\; class="notranslate">\; T</span>}}_{\mathrm{<span\; class="notranslate">\; \&Phi;</span>}}<span\; class="notranslate">\; =</span>\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{\mathrm{<span\; class="notranslate">\; K</span>}<span\; class="notranslate">\; \&CenterDot;</span>{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; Ld</span>}}^{<span\; class="notranslate">\; *</span>}}<span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; \&Center\; Dot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 12</span>}<span\; class="notranslate">\; )</span>$

Therefore, the integral gain K of 8a1 is calculated as shown in Equation 13.

[number 13]

K=1/ω ₁ ^* ω _c /Ld ^* ... (13)

Among them, _ωc is the control response angular frequency (rad/s) of the weak field command operation. Then, the new transfer function G _Ф ′(s) becomes

[number 14]

${{\mathrm{<span\; class="notranslate">\; G</span>}}^{<span\; class="notranslate">\; \text{'}</span>}}_{\mathrm{<span\; class="notranslate">\; \&Phi;</span>}}<span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; S</span>}<span\; class="notranslate">\; )</span><span\; class="notranslate">\; =</span>\frac{\frac{\mathrm{<span\; class="notranslate">\; 1</span>}}{{{\mathrm{<span\; class="notranslate">\; \&omega;</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}}^{<span\; class="notranslate">\; *</span>}<span\; class="notranslate">\; \&CenterDot;</span>{\mathrm{<span\; class="notranslate">\; Ld</span>}}^{<span\; class="notranslate">\; *</span>}}}{\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; +</span>{{\mathrm{<span\; class="notranslate">\; T</span>}}^{<span\; class="notranslate">\; \text{'}</span>}}_{\mathrm{<span\; class="notranslate">\; \&Phi;</span>}}<span\; class="notranslate">\; \&CenterDot;</span>\mathrm{<span\; class="notranslate">\; the\; s</span>}}<span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; \&CenterDot;</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 14</span>}<span\; class="notranslate">\; )</span>$

where the new response time constant T _Ф ′ is

[number 15]

T _Ф ′＝1/ω _c …(15)

In this way, T _Ф ′ can be set independently of the frequency command ω ₁ ^* , and an effect of higher response can be obtained.

In addition, the permanent magnet synchronous motor control device of the mode of changing the integral gain of the field weakening command calculation part under the feedback control mode of this embodiment by using the frequency command ω ₁ ^* can also be applied to the voltage vector shown in FIG. In a control system other than a control system with a current command calculation unit in the upstream of the calculation unit.

[Example 3]

Fig. 7 shows another embodiment of the present invention. The present embodiment is a field weakening vector control device for a permanent magnet synchronous motor when a feedforward method is used in the field weakening command calculation unit.

In FIG. 7 , 1 to 7, 9 to 13, and 21 of the constituent elements are the same as those in FIG. 1 .

Using FIG. 8 , the field weakening command computing unit 8 b in the feedforward control system, which is the characteristic of the present invention, will be described.

In addition to the present embodiment, the d-axis current command generated at no load is obtained by performing calculations in advance.

With respect to the field weakening command calculation section 8b, in the calculation section 8b1, the induced voltage command value (=ω ₁ ^* Ke ^* ) is subtracted from the output voltage command value V ₁ ^* _ref in the field weakening region, and this is subtracted to obtain The value of is divided by the product of ω ₁ ^* and Ld ^* . The output value of the computing unit 8b1 is input to the primary delay filter 8b2. And, the output value of 8b2 is input to the limiter operation part 8b3 which limits the positive side to "zero", and the output value is the 1st d-axis current command Id ^* .

Using this current command value Id ^* , a second current command value Id ^** is generated, and voltage command values Vd ^* , Vq ^* are calculated to control the converter output voltage.

In the high- ^speed region, even if the torque is "zero", V ₁ ^{* is saturated only by the induced voltage command value of Vq *} (=ω ₁ ^* Ke ^* ).

If the d-axis current command value necessary to leave the voltage saturation region is set as Id ^* _ff0 , then

[number 16]

Id ^* _ff0 = (V ₁ ^* max - ω ₁ ^* Ke ^* )/(ω ₁ ^* Ld ^* ) ... (16)

In this way, by setting the first-order delay filter time constant T of 8b2 as in (17), even in the feedforward control mode, the same effect as that of the second embodiment can be obtained.

[number 17]

T=1/ω _c ... (17)

In addition, the field-weakening torque control device for a permanent magnet synchronous motor in the case where the feedforward method is used in the field-weakening command computing part of this embodiment can also be applied to the voltage vector computing part shown in FIG. In a control system other than a control system with a current command calculation unit in the upstream part.

[Example 4]

Fig. 9 shows another embodiment of the present invention. This embodiment is a control device for a permanent magnet synchronous motor when a feedforward control method and a feedback control method are used in the field weakening command calculation unit.

In FIG. 9 , 1 to 7, 9 to 13, and 21 of the constituent elements are the same as those in FIG. 1 . Using FIG. 10 , the field weakening command computing unit 8c in the feedforward control method and the feedback control method, which are the characteristics of the present invention, will be described.

In the field weakening command calculation unit 8c, the calculation unit 8c1 subtracts the induced voltage command value (=ω ₁ ^* Ke ^* ) from the output voltage command value V ₁ ^* _ref in the field weakening region, and subtracts this resultant The value is divided by the product of ω ₁ ^* and Ld ^* .

The output value of the calculation part 8c1 is input to the primary delay filter 8c2. The output value of 8c2 is input to the limiter calculation part 8c3 which restricts the positive side to "zero", and the output value is Id ^* _ff .

In addition, the output voltage command value V ₁ ^* _ref and the output voltage value V ₁ are simultaneously input to the integral calculation unit 8c4 whose integral gain is constant K, and integral calculation is performed. At this time, the integral gain K is automatically corrected by the frequency _ω1 .

The output value of the integral operation part 8c4 is input to the limiter operation part 8c5 which limits the positive side to "zero", and the output value is Id ^* _fb .

Therefore, the first d-axis current command Id ^* is calculated by adding the output value Id ^* _ff of the feedforward control and the output value Id ^* _fb of the feedback control as shown in (18).

[number 18]

Id ^* = Id ^* _ff + Id ^* _fb ... (18)

In this way, the same operation as the above-mentioned embodiment can be obtained, and the effect of higher response can be obtained.

In addition, the control device of the permanent magnet synchronous motor in the case of using the feedforward control method and the feedback control method on the field weakening command calculation part of the present embodiment can also be applied to the voltage vector as shown in FIG. In a control system other than a control system with a current command calculation unit in the upstream of the calculation unit.

[Example 5]

In Embodiments 1 to 4, the detected three-phase AC currents Iu to Iw are detected by the expensive current detector 3 , but they can also be applied to a control device that detects inexpensive currents.

Figure 11 shows this embodiment. In FIG. 11, 1, 2, 4 to 7, 8a, 9 to 13, and 21 of the constituent elements are the same as those shown in FIG. 5 .

14 is a current estimating unit, which estimates the three-phase AC currents Iu, Iv, and Iw in the motor 1 based on the DC current IDC on the input bus of the power converter.

Using these estimated current values Iu^, Iv^, Iw^, in the coordinate conversion unit 7, the d-axis and q-axis current detection values Idc, Iqc are calculated.

Since Id ^* and Idc, and Iq ^* and Iqc are also made to coincide with each other in this current sensorless control method, it can be seen that the operation is the same as that of the above-mentioned embodiment, and the same effect is obtained.

In addition, although in the present embodiment, the form shown in FIG. 6 is used in the field weakening command calculation unit, the same effects can be obtained by using the forms shown in FIGS. 2 , 8 , and 10 .

[Example 6]

Fig. 12 shows another embodiment of the present invention.

This embodiment is suitable for a control device that performs low-cost current detection and omits a magnetic pole position detector.

In FIG. 12, 1, 2, 7, 8a, 9 to 13, and 21 of the constituent elements are the same as those shown in FIG.

6' is a phase calculation unit which integrates the frequency command ω ₁ ^* to calculate the rotation phase command θc ^* .

14 is a current estimating unit, which estimates the three-phase AC currents Iu, Iv, and Iw in the synchronous motor based on the DC current IDC on the input bus of the power converter.

Using the estimated current values Iu^, Iv^, Iw^, the coordinate conversion unit 7 calculates d-axis and q-axis current detection values Idc, Iqc.

In addition, 15 is a phase error computing ^unit , which ^estimates a ^phase error Δθc (= θc ^* -θ).

16 is a frequency estimation part which calculates _ω1 ^** so that a phase error Δθc may become "zero". Also in this position and current sensorless control method, it operates in the same manner as the above-mentioned embodiment, and it can be seen that the same effect can be obtained.

In addition, in this embodiment, although the form of FIG. 6 is used in the field weakening command calculation part, the same effect can be obtained also using the form of FIG. 2, FIG. 8, and FIG. 10.

[Example 7]

An example in which the present invention is applied to a module will be described using FIG. 13 . This example shows the implementation of Example 1. Here, the frequency calculation unit 5, the phase calculation unit 6, the coordinate conversion unit 7, the field weakening command calculation unit 8, the d-axis current command calculation unit 9, the q-axis current command calculation unit 10, the voltage vector calculation unit 11, and the output voltage calculation unit 12. The coordinate conversion unit 13 is composed of a single-chip microcomputer. Furthermore, the above-mentioned single-chip microcomputer and the power converter are housed in one module formed on the same substrate. The term "module" here means a "standardized constituent unit" and is composed of separable hardware/software components. In addition, although it is preferable to configure on the same substrate in terms of production, it is not limited to the same substrate. Therefore, it can also be configured on a plurality of circuit boards built in the same housing. In other embodiments, the same configuration can also be adopted.

As above, through the present invention, it is possible to provide a field-weakening vector control device for a permanent magnet synchronous motor, which can realize high-precision, high-response motor torque even in a field-weakening region; Commonly used in a system that detects a magnetic pole position detector and a system that omits a magnetic pole position detector.

Claims (10)

1. A control device for a permanent magnet synchronous motor, according to the current command value of the second d axis and the q axis calculated by the current command value and the current detection value of the first d axis and the q axis, and the frequency command value, to The output voltage value of the power converter driving the permanent magnet synchronous motor is controlled, and it is characterized in that: A field weakening command calculation unit is provided, and an integral calculation value of a deviation between an output voltage command value and the output voltage value is used as the first d-axis current command value. 2. The control device of the permanent magnet synchronous motor according to claim 1, characterized in that, In the integral operation of the deviation between the output voltage command value and the output voltage value, the integral gain is corrected according to the frequency command value. 3. A control device for a permanent magnet synchronous motor, which controls the output voltage value of the power converter that drives the permanent magnet synchronous motor according to the current command value and the frequency command value of the d axis and the q axis, characterized in that: having a field-weakening command calculation unit for taking an integral calculation value of an output voltage command value and a deviation from the output voltage value as a d-axis current command value, In the integral operation of the deviation between the output voltage command value and the output voltage value, the integral gain is corrected according to the frequency command value. 4. A control device for a permanent magnet synchronous motor, which controls the output voltage value of the power converter that drives the permanent magnet synchronous motor according to the current command value and the frequency command value of the d axis and the q axis, it is characterized in that, The d-axis current command value is calculated using the output voltage command value, frequency command value, and motor constant in the field weakening region. 5. The control device of the permanent magnet synchronous motor according to claim 4, characterized in that, The sum of the output voltage command value and the integral calculation value of the deviation of the output voltage value, and the value calculated using the output voltage command value, the frequency command value, and the motor constant is taken as the d-axis current command value. 6. The control device of the permanent magnet synchronous motor according to claim 5, characterized in that, In the integral operation of the deviation between the output voltage command value and the output voltage value, the integral gain is corrected according to the frequency command value. 7. The control device of the permanent magnet synchronous motor according to claim 4, characterized in that, calculation using the output voltage command value and frequency command value and the motor constant, the induced voltage command value of the motor is subtracted from the output voltage command value, and the subtracted value is divided by the frequency command value and the d-axis product of inductance. 8. The control device of the permanent magnet synchronous motor according to claim 1, characterized in that, The current detection value is a current that reproduces the motor current based on the input DC bus current detection value of the power converter. 9. The control device of the permanent magnet synchronous motor according to any one of claims 1 and 8, characterized in that, The deviation between the rotational phase command and the rotational phase of the motor is calculated using the d-axis and q-axis voltage command values and the detected motor current or the reproduced current, and the deviation is zeroed to calculate the Frequency command value. 10. A module, characterized in that, comprising: A control device as claimed in any one of claims 1 to 9; and, A power converter that converts DC to AC.

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