CN1486008A - Method and apparatus for automatic frequency correction in CDMA system - Google Patents
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Abstract
The invention provides a CDMA system automatic frequency correcting device, including channel estimation unit, frequency error estimation unit, loop filter unit, D/A converter and voltage-controlled oscillator. The channel unit includes delay unit, multiplier, superposition unit and pilot filter. The frequency error estimation unit includes delay memory unit, multiplier, inverse tangent unit and frequency-deviation multipath union unit, and estimates the frequency error to control output frequency of the voltage-controlled oscillator. It adopts pilot signal despreaded by PN juncition and integrated as channel estimated value, and after smoothly processing and simply calculating, it obtains each path of frequency error estimated value.
Description
Technical Field
The present invention relates to the field of wireless communication, and in particular, to an automatic frequency correction method and apparatus in a code division multiple access system.
Background
The mobile communication system of today is basically a cellular mobile communication system. Cellular mobile communication systems have undergone several generations of development, the first generation being analog cellular mobile communication systems employing Frequency Division Multiple Access (FDMA), such as AMPS in the united states, TACS system in the united kingdom, etc.; the second generation is basically a digital cellular mobile communication system employing Time Division Multiple Access (TDMA), such as the european GSM system; the Code Division Multiple Access (CDMA) cellular mobile communication system has the characteristics of simple frequency planning, large system capacity, strong multipath interference resistance and the like, so that the CDMA cellular mobile communication system becomes the main technology of a third-generation cellular mobile communication system.
In the ground, the main characteristic of the communication system channel is multipath propagation. The propagation environment of mobile communication is very complicated by the fact that signals encounter many buildings, trees and undulating terrain during propagation, which causes absorption and penetration of energy and reflection, scattering and diffraction of electric waves. In a mobile communication environment, a signal arriving at an antenna of a mobile station is not from a single path but is a result of the synthesis of a plurality of reflected waves of multiple paths. The CDMA system employs RAKE reception techniques to overcome the effects of mobile channel multipath fading on the signal and employs a pilot channel to estimate multipath channel parameters for maximum ratio combining. Due to the relative instability of the transceiver clock frequency and the doppler effect caused by the relative motion of the transceiver, the receiver local carrier and the received signal carrier frequency have frequency deviation, and even if the parameter obtained by channel estimation is used for maximum ratio combination, the performance of the receiver is reduced along with the increase of the frequency difference. It is therefore necessary to compensate this unknown frequency difference using automatic frequency control techniques to ensure proper operation of the receiver. Considering that the fixed frequency offsets of the receiving and transmitting ends are superposed in the phase information of the multipath fading signal, the multipath channel estimation value is obtained by adopting the pilot frequency, the multipath channel estimation value is simply processed, the frequency deviation of the receiving and transmitting ends can be extracted to obtain the estimation value of the frequency deviation, and then the estimation value is used for adjusting the frequency of the local oscillator, so that the function of automatic frequency correction can be realized. Thus, the closer the frequency offset estimate is to the unknown frequency difference, the more accurate the frequency correction will be.
In the schematic diagram of the receiver with automatic frequency correction shown in fig. 1, the receiver includes mixers 100A, 100B, matched filters 110A, 110B, down- sampling modules 120A, 120B, a RAKE receiver 130, a channel estimation and frequency deviation estimation unit 140, a loop filtering unit 160, a D/a converter 150, a voltage controlled oscillator 170, and a phase shifting unit 190. The received signals y (t) enter the mixers 100A and 100B, respectively, are mixed in the mixer 100A with the local oscillation signal generated by the vco 170 and passing through the phase shift unit 190, and are mixed in the mixer 100B with the local oscillation signal generated by the vco 170. The outputs of mixers 100A, 100B pass through matched filters 110A, 110B, respectively, and are down-sampled by down- sampling modules 120A, 120B to the chip rate, and the down-sampled signals are processed in RAKE receiver 130, such as PN despreading, integration, and weight combining. Channel estimation and frequency deviation estimation unit 140 performs channel estimation and calculates frequency deviation estimation value using effective path data output from RAKE receiver 130 Estimating the frequency deviation
After multiplying by a fixed coefficient K, the signal is filtered by the loop filter unit 160 to obtain Δ w, so that noise interference and other frequency components are eliminated. The voltage signal output from the loop filter unit 160 controls the output frequency of the voltage controlled oscillator 170, thereby completing the automatic frequency correction process. The automatic frequency correction apparatus generally includes a channel estimation and frequency offset estimation unit 140, a loop filter unit 160, a D/a conversion 150, and a voltage controlled oscillator 170. The channel estimation and frequency deviation estimation unit 140 mainly performs channel parameter estimation and instantaneous frequency deviation estimation, and the obtained frequency deviation estimation value is sent to the loop filtering unit 160, and the loop filtering unit 160 filters other combined frequency components and other interference components in the frequency deviation estimation, so as to ensure the performance required by the loop. After the output of the loop filter unit 160 is D/a converted 150, the local reference frequency source vco 170 is adjusted to gradually approach the actual carrier frequency.
U.S. Pat. No. 5, 5764687A, Mobile Demodulator Architecture for a spread spectrum Multiple Access Communication System, proposes an automatic frequency correction method, which uses a cross product method to calculate the frequency deviation for each path, and then adds them to obtain the total frequency difference.
In the method, because the transmitted known information pilot carries the information of the channel, the channel estimation value is represented by the pilot which is used in the RAKE receiver and is subjected to multipath weighting combination, the pilot is a signal subjected to PN despreading and integration, the interference of other channels is removed, and the inter-symbol smoothing is performed through a pilot filter, so that the fluctuation among pilot symbols is small.
Let p (n) be the pilot received at a certain time, and p (n) be regarded as a complex number composed of I, Q two parts, i.e. p (n) ═ pI(n)+pQ(n), then its cross product with the pilot at the previous time instant is:
p(n)×p(n-1)＝pI(n)pQ(n-1)-pI(n-1)pQ(n) (1)
the cross product p (n) x p (n-1) is the imaginary part of the product of p (n-1) multiplied by the imaginary part of p (n). The cross product yields the phase change value.
Is provided with <math> <mrow> <mi>p</mi> <mrow> <mo>(</mo> <mi>n</mi> <mo>-</mo> <mn>1</mn> <mo>)</mo> </mrow> <mo>=</mo> <msup> <mrow> <mi>A</mi> <mn>1</mn> </mrow> <mo>*</mo> </msup> <msup> <mi>e</mi> <mrow> <mi>jΔw</mi> <msub> <mi>T</mi> <mn>1</mn> </msub> </mrow> </msup> <mo>,</mo> </mrow> </math> <math> <mrow> <mi>p</mi> <mrow> <mo>(</mo> <mi>n</mi> <mo>)</mo> </mrow> <mo>=</mo> <msup> <mrow> <mi>A</mi> <mn>2</mn> </mrow> <mo>*</mo> </msup> <msup> <mi>e</mi> <mrow> <mi>jΔw</mi> <msub> <mi>T</mi> <mn>2</mn> </msub> </mrow> </msup> <mo>,</mo> </mrow> </math>
Then <math> <mrow> <mi>p</mi> <mrow> <mo>(</mo> <mi>n</mi> <mo>-</mo> <mn>1</mn> <mo>)</mo> </mrow> <mo>·</mo> <mover> <mi>p</mi> <mo>-</mo> </mover> <mover> <mrow> <mo>(</mo> <mi>n</mi> <mo>)</mo> </mrow> <mo>-</mo> </mover> <mo>=</mo> <mi>A</mi> <mn>1</mn> <mi>A</mi> <mn>2</mn> <msup> <mi>e</mi> <mrow> <mi>jΔw</mi> <mrow> <mo>(</mo> <msub> <mi>T</mi> <mn>1</mn> </msub> <mo>-</mo> <msub> <mi>T</mi> <mn>2</mn> </msub> <mo>)</mo> </mrow> </mrow> </msup> <mo>=</mo> <msup> <mi>Ae</mi> <mi>jΔwT</mi> </msup> <mo>-</mo> <mo>-</mo> <mo>-</mo> <mo>-</mo> <mrow> <mo>(</mo> <mn>2</mn> <mo>)</mo> </mrow> </mrow> </math>
I.e., Im (p (n)) p (n-1)) ═ Asin (Δ wT) (3)
When Δ wT < 1, (3) can be expressed as:
as can be seen from equation (4), the cross product results and the frequency deviation estimate
Proportional relationship, so that the frequency deviation of this path can be obtained. But whenWhen it becomes larger, the linear region is exceeded, and the above relationship is no longer established. The structure of the frequency deviation estimation unit obtained by the method is shown in fig. 2, and comprises 2 delay units, 2 multipliers and 1 adder, wherein I, Q two paths of signals are respectively delayed by the delay units and then multiplied by Q, I signals, and the obtained product is subtracted to obtain the frequency deviation estimation unit
The value of (c). However, the patent does not consider the effect of the final frequency difference on the interference caused by each path.
In the Journal of south University, 2002 2, an article "A Novel AFC in the 3rd Generation Mobile Communication System" describes a method for calculating an estimated value of frequency deviation.
Since the pilot channel signal in a CDMA system is a signal that is always transmitted and known, assuming that the input signal containing the pilot signal is s (t), the impulse response of the multipath channel is:
then, the baseband signal received by the receiving end is:
then
*l＝θl+ Δ wt where | τl-τk|＞1/B，l≠k，l.k＝1，2，......L，αlIs the received amplitude of the path signal, θlIs the received phase, τ, of the path signallIs the reception delay of the signal of the path L, L is the number of multipaths, Δ w is the deviation of the transmission frequency from the reception frequency, nl(t) is complex Gaussian white noise with a mean of 0 and a variance of N0. Let alphal，φlIs an estimate of the amplitude and phase of the l-path signal, then alphal，φlThe maximum likelihood function of (a) is expressed as:
derivation of equation (8) yields:
then the channel estimation value at the time of the ith path n can be obtained:
assuming that the channel remains unchanged for T time, there are
This gives an estimate of the frequency deviation for this path:
in some systems, the frequency offset estimation is performed by using this method, and the schematic structure of the channel estimation unit and the frequency offset estimation unit is shown in fig. 3. The channel estimation unit includes a group of delay units 341, a group of multipliers 342, a group of low pass filters 343 and a comparison selector 344, the number of the delay units 341, the multipliers 342 and the low pass filters 343 is the same and are connected in sequence, the received baseband signal r (k) is subjected to PN despreading after passing through the delay unit 341, the despread signal is output to the comparison selector 344 after passing through the low pass filter 343, the channel estimation value of the strongest path in the multipath is selected, and then the channel estimation value enters the frequency deviation estimation unit, and the frequency deviation estimation unit includes a delay unit 346, a multiplier and an arc tangent unit 345. In the frequency offset estimation unit, the channel estimation value of the strongest path is multiplied by the reciprocal of the channel estimation value of the strongest path through the delay unit 346, and the product is processed by the arc tangent unit 345 to obtain the frequency offset estimation value, and the frequency offset estimation value obtained by the strongest path is used as the frequency offset estimation value of all paths. But doing so can introduce large deviations to the other paths and thus introduce even larger errors into the system. Since the channel estimation signal is obtained by the low-pass filter 343 after the baseband signal is subjected to PN despreading, if the bandwidth of the selected low-pass filter 343 is not appropriate, the obtained signal chips have large fluctuation and interference of other channels remains, and thus the obtained channel estimation value itself has large error, which causes the overall performance of the loop to be degraded and reduces the quality of the communication system.
In the conventional automatic frequency correction apparatus, the loop filtering unit 160 generally employs a fixed step length K, which greatly reduces the convergence rate of the loop adjustment or increases the jitter error of the frequency offset after convergence, because a smaller step length is employed to result in a longer convergence time under the condition of a large frequency offset; and when the output of the voltage-controlled oscillator is close to the carrier frequency, the frequency offset jitter error is determined by the step length, and the smaller the step length is, the smaller the jitter error is.
Disclosure of Invention
The invention aims to solve the technical problem of providing an automatic frequency correction method and device of a code division multiple access system, and solves the problems of large calculation error of a frequency deviation estimated value and contradiction between loop convergence speed and frequency deviation jitter error in the prior art.
The automatic frequency correction method of the code division multiple access system comprises the following steps:
firstly, separating the received signals output by the matched filter according to the effective path, and carrying out pseudo-random code de-spreading on each path of signals;
integrating the despread signals, extracting pilot signals and obtaining channel estimation values;
thirdly, smoothing the channel estimation value;
fourthly, calculating the frequency deviation estimation value of each path according to the smoothed channel estimation value;
fifthly, calculating frequency deviation estimated values of all effective paths according to the weighting of the effective paths;
sixthly, generating a loop filter coefficient by using the frequency deviation value estimated value obtained in the step five;
and seventhly, performing loop filtering and controlling the output frequency of the voltage-controlled oscillator.
The automatic frequency correction device of the code division multiple access system comprises a channel estimation unit, a frequency deviation estimation unit, a loop filter unit, a D/A converter and a voltage-controlled oscillator which are connected in sequence, wherein the channel estimation unit obtains a channel estimation value after despreading and integrating a received signal by a pseudo-random code, outputs the channel estimation value to the frequency deviation estimation unit for frequency deviation estimation, outputs the obtained frequency deviation estimation value to the loop filter unit, and outputs a signal to control the output frequency of the voltage-controlled oscillator by the D/A converter,
the channel estimation unit comprises M delay units, M multipliers, M superposition units and M pilot frequency filters, wherein the signals pass through the M delay units to obtain aligned M-path signals, the aligned M-path signals are multiplied by complex conjugate signals of pseudo-random codes, the complex conjugate signals are superposed in the M superposition units to obtain pilot frequency signals, and finally, the pilot frequency signals are smoothed by the conducting filters to obtain channel estimation values of the M paths;
the frequency deviation estimation unit comprises M delay storage units, M multipliers, M arc tangent units and 1 frequency deviation multi-path merging unit, wherein the channel estimation values of M paths enter the M delay storage units for storage, are calculated in the multipliers and the arc tangent units together with the current channel estimation value to obtain the frequency deviation of the M paths, and are weighted and merged in the frequency deviation multi-path merging unit to finally obtain the frequency deviation estimation values of all paths;
the loop filtering unit comprises a threshold comparison unit, a multiplier and a filtering unit; the output of the frequency deviation estimation unit enters the threshold comparison unit and is used for calculating the loop filter coefficient required currently, multiplying the obtained loop filter coefficient and the frequency deviation estimation value in the multiplier, filtering through the filter unit and outputting to the D/A converter.
Aiming at the condition of multipath signals in a mobile communication system, the invention adopts pilot signals subjected to PN de-spread and integration as channel estimation values, obtains the frequency deviation estimation value of each path meeting the maximum likelihood criterion through smoothing processing and simple calculation, and then adopts the criterion meeting the minimum mean square error to carry out multipath frequency error combination, so that the error of each path is minimized. Meanwhile, the step length of the loop filter is changed according to different conditions of the environment, so that different requirements are met, and automatic frequency correction can be quickly and effectively realized. The method of the invention has simple calculation and high correction speed, and well meets the requirement of the CDMA terminal on the automatic frequency correction performance. In addition, the frequency shift caused by Doppler can be tracked and corrected, and the method can be used together with a RAKE receiver to directly utilize the channel estimation value of the RAKE receiver, thereby reducing the requirement of hardware.
Drawings
Fig. 1 is a schematic diagram of a general receiver with automatic frequency correction.
Fig. 2 is a schematic diagram of a conventional frequency offset estimation unit.
Fig. 3 is a schematic diagram of a conventional channel estimation and frequency offset estimation unit.
Fig. 4 is a schematic diagram of a channel estimation unit 410 and a frequency deviation estimation unit 420 in the automatic frequency correction apparatus of the present invention.
Fig. 5 is a schematic diagram of the multipath combining unit 423 in the frequency offset estimation unit 420 of fig. 4.
Fig. 6 is a schematic diagram of a loop filter unit 600 in the afc apparatus of the present invention.
Detailed Description
The present invention will be described in further detail with reference to the accompanying drawings and examples.
Fig. 1 to 3 illustrate the situation of the conventional automatic frequency correction system and frequency offset estimation unit, which have been described in detail in the background art and are not described herein again.
The invention integrates the advantages of the prior art, adopts the pilot frequency for multipath weighting combination in the RAKE receiver to carry out channel estimation, namely, the received baseband signal is integrated after PN de-spreading, the interference of other channels is removed, the influence of other channels on the channel estimation is eliminated, then the inter-symbol smoothing is carried out, so that the fluctuation between pilot frequency symbols is small, then the channel estimation is carried out on each path by adopting an anti-tangential method, and the multipath is combined by considering the influence of the multipath. The multipath combination adopts the minimum mean square error criterion, so that the error brought to each path by the finally obtained total frequency deviation estimated value is minimum, the error is reduced to the maximum extent, the step length of the loop filter is changed according to different conditions according to the obtained frequency difference, different requirements are met, and the excellent performance of the loop is ensured.
The basic principle of the present invention will be described first.
After the channel estimation value of each path is obtained, the frequency deviation estimation value of a single path can be obtained according to the formula (12):
in the formula TsIs the symbol period of the pilot.
If the influence of all effective multipaths is considered, the multipaths need to be combined, and the combining method meets the minimum mean square error criterion.
Suppose that the frequency difference obtained at a certain time isThe 1 st path obtains a frequency difference of
The amplitude of the pilot is betalTo make
At a minimum, derivative thereof
Obtaining:
wherein beta isl＝|cl(n)| (16)
In the loop filter unit, letThe step size of the loop filter used for the frequency offset estimation at the nth time is deltanThen, the step size at the nth output is determined by the following equation:
when the absolute value of the frequency deviation estimated value is larger than the set threshold, a larger step length is adopted to shorten the tracking time (frequency adjustment stage); when the absolute value of the estimated value of the frequency deviation is smaller than the set threshold, a smaller step length k is adopted2To improve the accuracy of the automatic frequency correction (frequency locking phase). In addition, a plurality of thresholds can be set for more precise control. The threshold can be selected according to different channel conditions, if the channel condition is poor, a slightly larger threshold can be selected to prevent the step length of the loop filter from changing all the time, and if the channel condition is good, a smaller threshold can be selected to achieve stability more quickly.
The output of the loop filter unit is used for controlling the output frequency of the voltage-controlled oscillator after D/A conversion, so that the output frequency gradually approaches to the carrier frequency value of the received signal.
Fig. 4 is a schematic diagram of a channel estimation unit 410 and a frequency deviation estimation unit 420 in the automatic frequency correction apparatus of the present invention. The channel estimation unit 410 includes a delay unit 411, a multiplication unit 412, an overlap unit 413, and a pilot filter 414; the frequency offset estimation unit 420 includes a delay storage unit 421, a multiplication unit, an arctangent unit 422, and a frequency offset multipath combining unit 423. The received signal gets multiple fingers (M) of the RAKE receiver through different delay units 411. The number of fingers M of the RAKE receiver is not necessarily the same as the actual number of paths L, and if L < M, M-L fingers are turned off; if L > M, the signals of the first M strongest paths of all L paths are demodulated, ignoring the remaining L-M paths.
In one embodiment of the present invention, assuming that M is 3, the aligned three-path signal is obtained by passing through the matched filter and the down-sampled signal r (k) and through the different delay units 411A, 411B, 411C. Each path signal is multiplied by the complex conjugate of the PN code in multipliers 412A, 412B, and 412C, and then the despread signals are added within a certain length N to obtain a pilot signal, which aims to eliminate interference of other channels to the pilot channel, the added length N should be as long as possible to ensure accuracy of pilot estimation, and the added length N should ensure that parameters of the channel do not change within the length interval, where N is 64 in the embodiment. In order to smooth the symbols of the pilot and the output between the symbols, the pilot filter 414A, 414B, 414C is used for smoothing, the pilot filter is a low pass filter, which may be an Infinite Impulse Response (IIR) filter or a Finite Impulse Response (FIR) filter, and the function of the low pass filter is to reduce the variation between the pilot symbols and the symbols and smooth the pilot, and in this embodiment, a first order IIR filter is selected. The output of the pilot filters 414A, 414B, 414C is the final channel estimate. These channel estimation values are stored in delay storage units 421A, 421B, 421C, and calculated with the current channel estimation value in multipliers and arc tangent units 422A, 422B, 422C, respectively, according to formula (13) to obtain an instantaneous frequency difference estimation value of each path, and then the frequency differences of each path are weighted and combined in a frequency offset multi-path combining unit 423, so that a final output frequency difference estimation value can be obtained
The schematic diagram of the frequency offset multipath combining unit 423 is shown in fig. 5, and includes M weighting units 4231 and an adder 4232. In the embodiment of the present invention, M is 3, and the frequency differences of the three paths are added in weighting units 4231A, 4231, and 4231C, respectivelyIn which beta isl1-1 … L is determined according to equation (16), and <math> <mrow> <mi>β</mi> <mo>=</mo> <munderover> <mi>Σ</mi> <mrow> <mi>l</mi> <mo>=</mo> <mn>1</mn> </mrow> <mi>L</mi> </munderover> <msub> <mi>β</mi> <mi>l</mi> </msub> <mo>.</mo> </mrow> </math> the selection of the weight value meets the minimum mean square error criterion, so that the finally obtained frequency difference and the error of each path deviation are minimized, thereby reducing the error brought to the system and improving the overall performance of the system. The weighted results are added in the adder 4232 to obtain the frequency offset estimation value
The variable step size loop filter unit 600 of the present invention is shown in fig. 6 and includes a threshold comparison unit 610, a multiplier 620 and a filter unit 630. In an embodiment of the present invention, the frequency offset estimation value obtained as described above
Entering a threshold comparison unit 610, performing threshold control according to a formula (17), determining different step lengths according to the threshold size, and obtaining a step length and a frequency offset estimation valueThe multiplication is performed in the multiplier 620, so as to accelerate the convergence rate of the filtering loop and ensure that the converged frequency offset jitter error is small, and then the filtering is performed through the filtering unit 630 to obtain Δ w, which is output to the D/a converter to control the output frequency of the voltage controlled oscillator.
The invention has been realized in the forward link simulation of cdma-20001 x, and the simulation proves that the invention can effectively correct the frequency difference between the transmission carrier and the receiving carrier, can achieve the stability within 0.1 second, and can track the Doppler frequency shift to a certain extent, thereby reducing the influence of the Doppler frequency shift on the system and improving the overall performance of the system. It is apparent that the present invention can also be used in several rate sets of cdma 20001x reverse links using pilots. In addition, the present invention can also be used in a WCDMA system.
Claims (6)
1. An automatic frequency correction method for a code division multiple access system, comprising the steps of:
firstly, separating the received signals output by the matched filter according to the effective path, and carrying out pseudo-random code de-spreading on each path of signals;
integrating the despread signals, extracting pilot signals and obtaining channel estimation values;
thirdly, smoothing the channel estimation value;
fourthly, calculating the frequency deviation estimation value of each path according to the smoothed channel estimation value;
fifthly, calculating frequency deviation estimated values of all effective paths according to the weighting of the effective paths;
sixthly, generating a loop filter coefficient by using the frequency deviation value estimated value obtained in the step five;
and seventhly, performing loop filtering and controlling the output frequency of the voltage-controlled oscillator.
2. The automatic frequency correction method of claim 1, wherein the formula for calculating the frequency deviation estimate of each path in step four is
Wherein,is an estimate of the frequency deviation of the path, cl(n) is the channel estimate at time n of the first path, cl(n-1) is a channel estimation value at the time of the l-th path (n-1), TsIs the symbol period of the pilot.
3. The automatic frequency correction method according to claim 1 or 2, wherein the calculation formula of all the effective path frequency deviation estimation values in the fifth step is
βl＝|cl(n)|，
Wherein,
is an estimate of the frequency deviation, beta, of all the effective paths at time nlIs the pilot amplitude of the l-th path,
is the estimated frequency deviation value of the path I obtained by the calculation in the step four, cl(n) is a channel estimation value at the time of the nth path, and L is the number of multipaths.
4. The automatic frequency correction method according to claim 3, wherein the sixth step further comprises determining whether the absolute value of the estimated frequency deviation value at the current time is greater than a predetermined threshold, and if so, applying a larger loop filter coefficient k1(ii) a If the value is less than the threshold value, a smaller loop filter coefficient k is adopted2。
5. An automatic frequency correction device of a code division multiple access system comprises a channel estimation unit (410), a frequency deviation estimation unit (420), a loop filter unit (600), a D/A converter and a voltage-controlled oscillator which are connected in sequence, wherein the channel estimation unit (410) despreads and integrates a received signal by a pseudo-random code to obtain a channel estimation value, outputs the channel estimation value to the frequency deviation estimation unit (420) for frequency deviation estimation, outputs the obtained frequency deviation estimation value to the loop filter unit (600), and outputs a signal to control the output frequency of the voltage-controlled oscillator by the D/A converter,
the channel estimation unit (410) comprises M delay units (411), M multipliers (412), M superposition units (413) and M pilot filters (414), wherein the signals pass through the M delay units (411) to obtain aligned M-path signals, the aligned M-path signals are multiplied by complex conjugate signals of pseudo-random codes, the aligned M-path signals are superposed in the M superposition units (413) to obtain pilot signals, and finally, the pilot signals are smoothed by the pass filter (414) to obtain channel estimation values of the M paths;
the frequency deviation estimation unit (420) comprises M delay storage units (421), M multipliers, M arc tangent units (422) and 1 frequency offset multi-path merging unit (423), wherein channel estimation values of M paths enter the M delay storage units (421) for storage, are calculated in the multipliers and the arc tangent units (422) together with the current channel estimation values to obtain frequency deviations of the M paths, and are weighted and merged in the frequency offset multi-path merging unit (423) to finally obtain frequency deviation estimation values of all paths.
6. The automatic frequency correction device according to claim 5, characterized in that the loop filter unit (600) comprises a threshold comparison unit (610), a multiplier (620) and a filter unit (630); the output of the frequency deviation estimation unit (420) enters the threshold comparison unit (610) for calculating the currently required loop filter coefficient, and the obtained loop filter coefficient and the frequency deviation estimation value are multiplied in the multiplier (620), and then filtered by the filtering unit (630) and output to the D/A converter.
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Cited By (8)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN100367690C (en) * | 2004-06-04 | 2008-02-06 | 中兴通讯股份有限公司 | Frequency deviation estimation and correction method in TD-SCDMA system |
| CN1777160B (en) * | 2004-11-15 | 2010-09-29 | 上海宣普实业有限公司 | Coarse frequency correcting method |
| CN1780271B (en) * | 2004-11-19 | 2011-05-18 | 上海宣普实业有限公司 | Path merge and related method |
| CN101312350B (en) * | 2007-05-23 | 2012-01-25 | 联发科技股份有限公司 | Communication system and demodulation method for detecting burst noise and reducing its influence |
| CN101557243B (en) * | 2008-04-11 | 2013-09-11 | 中兴通讯股份有限公司 | Device and method for carrying out self-adaptive filtering to pilot channel |
| CN106534033A (en) * | 2016-12-06 | 2017-03-22 | 西安电子科技大学 | Combined OFDM/OQAM time frequency synchronization method under multipath channel |
| CN108600132A (en) * | 2018-04-08 | 2018-09-28 | 深圳市盛路物联通讯技术有限公司 | A kind of transmitting frequency calibration method, system, equipment and computer readable storage medium |
| CN110392008A (en) * | 2018-04-23 | 2019-10-29 | 北京展讯高科通信技术有限公司 | Baseband carrier frequency tracking, device and terminal |
-
2002
- 2002-09-24 CN CNA02137239XA patent/CN1486008A/en active Pending
Cited By (10)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN100367690C (en) * | 2004-06-04 | 2008-02-06 | 中兴通讯股份有限公司 | Frequency deviation estimation and correction method in TD-SCDMA system |
| CN1777160B (en) * | 2004-11-15 | 2010-09-29 | 上海宣普实业有限公司 | Coarse frequency correcting method |
| CN1780271B (en) * | 2004-11-19 | 2011-05-18 | 上海宣普实业有限公司 | Path merge and related method |
| CN101312350B (en) * | 2007-05-23 | 2012-01-25 | 联发科技股份有限公司 | Communication system and demodulation method for detecting burst noise and reducing its influence |
| CN101557243B (en) * | 2008-04-11 | 2013-09-11 | 中兴通讯股份有限公司 | Device and method for carrying out self-adaptive filtering to pilot channel |
| CN106534033A (en) * | 2016-12-06 | 2017-03-22 | 西安电子科技大学 | Combined OFDM/OQAM time frequency synchronization method under multipath channel |
| CN106534033B (en) * | 2016-12-06 | 2019-11-05 | 西安电子科技大学 | OFDM/OQAM time frequency combined synchronizing method under a kind of multipath channel |
| CN108600132A (en) * | 2018-04-08 | 2018-09-28 | 深圳市盛路物联通讯技术有限公司 | A kind of transmitting frequency calibration method, system, equipment and computer readable storage medium |
| CN110392008A (en) * | 2018-04-23 | 2019-10-29 | 北京展讯高科通信技术有限公司 | Baseband carrier frequency tracking, device and terminal |
| CN110392008B (en) * | 2018-04-23 | 2022-05-31 | 北京紫光展锐通信技术有限公司 | Baseband carrier frequency tracking method and device and terminal |
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