CN1230990C - Searching for signals in a communications system - Google Patents

Abstract

A method and system simplify matched filters for CDMA systems using a two-stage search. A coarse stage together with a fine stage produces the positions (groups) of received signal path rays. In the first stage, an oversampled digital signal (240) is decimated, and the decimated signal (415) is applied to a matched filter (420) to finally produce an approximate position (460). In the second stage, the oversampled signal (240) is shifted based on the determined approximate position (460) and then correlated with the generated code (440), and a more precise position is then selected from the output of the correlation (450) . Alternatively, a shifted version of the generated code (440) is correlated with the oversampled signal (240), and more precise locations are selected from the output of those correlations (450).

Description

Search for signals in communication systems

technical field

The present invention relates generally to the field of communications and, by way of non-limiting example, in particular to tuning signal path rays in a wireless communications system, such as a Code Division Multiple Access (CDMA) system.

Background technique

Mobile wireless communications are becoming increasingly important to provide security, convenience, improved productivity, and simple conversational desires to wireless communication service users. An excellent mobile wireless communication option is cellular communication. Cellular phones can be found, for example, in vehicles, briefcases, purses, and even pockets. As cellular telephone users and the types of services provided proliferate, new wireless system standards are being developed to meet these demands.

For example, CDMA, Wideband-CDMA (W-CDMA), etc. are being implemented to improve spectral efficiency and introduce new features. In CDMA or W-CDMA (hereinafter collectively referred to as "CDMA"), signal fading is combated by combining multiple received rays of different signal paths in a rake receiver. The positions (in time) of the signal path rays are first found by using a searcher. Subsequently, the path rays are combined by using a maximum ratio combiner (MRC). Searchers are conventionally implemented as one or more matched filters and a peak detector. The signal path rays are matched to certain pilot sequences, which results in some peaks representing the positions of the individual path rays. A peak detector detects these resulting peaks.

Implementing a searcher is a computationally complex endeavor; therefore, it is ideal to detect path rays only once. After detection, the path ray is thus traced as long as possible by using a path ray tracer. Tracking continues until the received signal quality reaches (eg drops) a predetermined threshold. Thereafter, tracking is stopped and a new search is started. The computational complexity of a searcher arises, at least in part, from the number of delayed candidates that the searcher must consider to locate a path ray. The greater the number of delay candidates, the greater the cost in terms of hardware, processing time, power consumption, silicon real estate, and so on. Therefore, there is a need for a means to reduce the total number of delay candidates that a searcher must consider when locating different signal path rays.

Contents of the invention

The needs of the prior art are met by the method and system of the present invention. For example, as heretofore unrealized, it would be beneficial to reduce the total number of delay candidates that a receiver searcher must consider when locating different signal path rays. In fact, it would be beneficial if a searcher split the matching process into coarse signal matching and fine signal matching in order to reduce the number of delay elements involved in computing signal path ray positions.

In one embodiment, the present invention involves searching for signal path rays in a CDMA system. The present invention is directed to performing a primary coarse search of the signal path rays to determine their general location and then performing a secondary fine search to determine their precise location.

The methods and systems of the present invention are generally directed to simplifying matched filters in CDMA receivers. The matched filter is simplified by reducing the number of delay candidates that must be processed when searching for the location of a signal path ray to be received. The simplification of the matched filter is accomplished by implementing a two-stage signal path ray position searcher. The first coarse stage locates the approximate location of a signal path ray. The second refinement stage locates the signal path rays more correctly. The more precise position can then be forwarded to a set of rake fingers in a spread spectrum receiver.

In one embodiment, an analog received signal is oversampled in an analog-to-digital conversion. In other words, the analog signal is sampled more than once per chip. This oversampled signal is then decimated to reduce the number of entries in the digital signal. The decimated signal is applied to a matched filter, which may consist of at least one finite impulse response (FIR) filter. A peak detector detects an approximate position in the output from the FIR filter.

In response to the determined approximate position or positions, the oversampled signal is shifted. A code generator generates a code corresponding to expected data to be received. The shifted oversampled signal is correlated with the generated code, and a comparator selects the more precise position from the result of the correlated set. In another embodiment, the generated code is shifted and then correlated with the oversampled signal. In addition, a comparator selects a more precise position from the results of the related group.

The technical advantages of the present invention include but are not limited to the following. It should be understood that a particular embodiment may not include any, let alone all, of the following exemplary technical advantages.

An important technical advantage of the present invention is that it reduces the complexity of the searcher in a CDMA receiver by reducing the number of delay elements that the searcher must use. This therefore reduces power consumption and reduces the amount of silicon space occupied by the searcher.

Another important technical advantage of the present invention is that it enables the search to be done using a two-stage scheme, thereby simplifying the computational complexity associated with the first stage.

Yet another important technical advantage of the present invention is the ability to first detect path rays with a coarse temporal resolution and then determine the position of path rays by adjusting them with a finer resolution.

The above and other features of the present invention are explained in detail hereinafter with reference to illustrative examples shown in the accompanying drawings. It should be understood by those skilled in the art that the described embodiments are provided for purposes of illustration and understanding and that numerous equivalent embodiments are contemplated.

Description of drawings

A more complete understanding of the method and system of the present invention can be had by referring to the following detailed description when taken in conjunction with the accompanying drawings, in which:

Figure 1 illustrates an exemplary portion of a wireless communication system according to the present invention;

Figure 2 illustrates exemplary transmit/receive equipment of the wireless communication system of Figure 1 in accordance with the present invention;

Figure 3 illustrates an exemplary air interface format according to an embodiment of the present invention;

Figure 4A illustrates signal path ray detection according to an exemplary embodiment of the present invention;

FIG. 4B illustrates signal path ray detection according to another exemplary embodiment of the present invention;

Figure 5A illustrates an exemplary higher level diagram of signal path ray detection according to the exemplary embodiment of Figures 4A and 4B of the present invention;

Figure 5B illustrates another exemplary higher level diagram of signal path ray detection in accordance with the exemplary embodiment of Figures 4A and 4B of the present invention; and

FIG. 6 illustrates an exemplary method in flow chart form of detecting signal path rays in two stages according to the invention.

Detailed ways

In the following description, for purposes of explanation and not limitation, specific details are set forth, such as particular circuits, logical modules (e.g., implemented in software, hardware, firmware, etc.), techniques, etc., in order to provide a comprehensive overview of the present invention. comprehensive understanding. It will be apparent, however, to one skilled in the art that the present invention may be practiced in other embodiments that depart from these specific details. In other instances, detailed descriptions of well-known methods, devices, logic code (hardware, software, firmware, etc.), etc. are omitted so as not to obscure the invention with unnecessary detail.

The preferred embodiment of the invention and its advantages are best understood by referring to Figures 1-6 of the drawings, like numerals being used for like and corresponding parts of the various drawings. It should be understood that the figures reflect the real (I) and complex (Q) parts of the entire signal value (set) (I+jQ).

The air interface aspects of International Mobile Telecommunications 2000 (IMT-2000), a so-called third generation standard, are used to describe an embodiment of the invention. However, it should be understood that the principles of the present invention are applicable to other wireless (or wired) communication standards (or systems), especially those using spread spectrum techniques, such as those based on some types of Code Division Multiple Access (CDMA) schemes, such as direct Sequential (DS) CDMA (eg, W-CDMA, IS-95-A, etc.), Frequency Hopping (FH) CDMA, Time-dodging CDMA, Frequency-Time Dodging CDMA, etc., all These are generally referred to herein as CDMA.

Referring now to FIG. 1, an exemplary portion of a wireless communication system in accordance with the present invention is illustrated generally at 100. Referring now to FIG. Wireless communication system 100 includes (portion of) a base transmit/receive antenna 105 , a base transmitter/receiver (ie, a transceiver (TRX) ) section 110 , and mobile stations 115 and 125 . Although only two mobile stations 115 and 125 are shown in FIG. 1, it should be understood that the wireless communication system 100 may include more than two mobile stations. Also illustrated are transmission 120 (from mobile station 115) and transmission 130 (from mobile station 125). As is known in the art, reflections, delays, etc. cause multiple signals (e.g., transmit signals 130A, 130B) of the transmission (e.g., transmission 130) to be received by the base station transmit/receive antenna 105 and subsequently processed by the base station TRX section 110. and 130C).

Referring now to FIG. 2, exemplary transmit/receive equipment of a wireless communication system of FIG. 1 is illustrated generally at 200 in accordance with the present invention. An information-carrying signal 205 is input to a spreader 210, which spreads the signal over a wide frequency range. The spread signal is modulated at a modulator 215 and then transmitted from an antenna 220 . Antenna 220 may be, for example, an antenna of one of mobile stations 115 and 125 . Transmission 225 is received at antenna 230 (among several different signals (eg, signal path rays) arriving at different times). The antenna 230 may be, for example, the base station transmit/receive antenna 105 . It should be noted, however, that the (receiving) antenna 230 may correspond to a mobile station and the (transmitting) antenna 220 may correspond to a base station in accordance with the principles of the present invention. Thus, for example, the two-stage search principle of the invention can also be implemented in cooperation with the receiver of the mobile station.

Continuing now with reference to FIG. 2 , antenna 230 receives transmission 225 , which may include multiple signals. Transmission 225 is processed by radio frequency (RF) section 235 , which forwards a signal 240 to rake receiver 245 . The rake receiver 245 combines the multiple signals to obtain an improved signal 280 ; the improved signal 280 is then forwarded to post-processing 290 . The rake receiver 245 includes a rake branch 255 and a combiner 275 . As part of, or perhaps only associated with, the rake receiver 245 are: a searcher 250 and a path tracker 260 . Searcher 250 , rake branch 255 , and path tracer 260 receive as input signal (group) 240 , which includes multiple signals of transmission 225 .

In accordance with the present invention, searcher 250 implements a two-stage search scheme to locate one or more of the signals of signal(s) 240, as described in more detail below with reference to FIGS. 4-6. The searcher 250 informs the rake branch 255 along line 265 of the determined location (group). Once the searcher 250 has located the signal path rays, the path tracer 260 attempts to track them whenever possible. Individual adjustments to continue tracing the signal path rays are transmitted from path tracer 260 to rake branch 255 along line 270 . Using position and tracking information from lines 265 and 270, respectively, rake branch 255 extracts signal path rays by despreading the received information in a manner known in the art. The extracted signal path rays are output to a combiner 275 which may utilize maximum ratio combining (MRC) to produce a modified signal 280 .

Referring now to FIG. 3, an exemplary air interface format is illustrated generally at 300 in accordance with an embodiment of the present invention. In a CDMA system, data is segmented into portions of predetermined duration specified by a given CDMA standard. These parts are in turn segmented into smaller and smaller parts until the smallest part, the chip, is reached. More specifically, information segmentation in a broadcast channel (BCCH) according to the IMT-2000 standard (which is a W-CDMA standard) is illustrated at 300 . A superframe 305 has a duration of 720 ms and is divided into 72 frames 310 . Each frame 310 is segmented into fifteen slots 315 and each slot 315 is also segmented into ten (10) symbols 320 . Finally, each symbol 320 consists of two hundred fifty-six (256) chips 325 .

Radio waves travel a calculable distance during each chip depending on the duration of the chip. For example, under the W-CDMA IMT2000 standard, radio waves propagate approximately 78.0 m in one duration corresponding to one chip 325 . In the W-CDMA IMT2000 standard, one chip 325 duration is defined to be 0.26 μs long. Therefore, 3*10 ⁸ m/s x 0.26*10 ⁻⁶ S=78.0 m/S, where the quantity 3*10 ⁸ m/s is equal to the speed of radio waves. Within a distance of 78.0m, some path rays can reach a CDMA receiver. Therefore, the received data is usually digitized by oversampling the chips in order to increase the resolution of the path ray arrival time detection. While oversampling enhances the searcher's performance, it also unfortunately increases its complexity, since the matched filter must therefore deal with the greater number of delay elements due to the oversampling. This complexity works against the extent to which it increases hardware requirements and/or processing time.

Referring now to FIG. 4A, signal path ray detection is illustrated generally at 250A in accordance with an exemplary embodiment of the present invention. Searcher 250A detects signal path rays when the undecimated received data is shifted and correlated with the generated code. Referring now to FIG. 4B, signal path ray detection is illustrated generally at 250B in accordance with another exemplary embodiment of the present invention. Searcher 250B detects signal path rays when the generated codes are shifted and correlated with undecimated received data.

As noted above, to increase the resolution of path ray arrival time detection in a CDMA system, the received data is preferably oversampled several times (eg, at least more than once) per chip. The oversampling rate can be defined as the number of times the received signal is sampled per chip. This oversampling creates a need for an increased complexity of the matched filter(s), since more delay elements are required for the implementation. However, in accordance with the principles of the present invention, this added complexity is prevented (e.g., reduced) by dividing the matching process/means into two (2) stages: Coarse Signal Matching (denoted "Stage 1") and Fine Signal Matching (denoted as "Phase 2").

Continuing now with searchers 250A and 250B of FIGS. 4A and 4B , respectively, coarse signal matching ("Stage 1") is described. One purpose of coarse signal matching is to roughly locate signal path rays. First, however, the incoming signal transmission (group) 225 (of FIG. 2) is converted from analog to digital using an A/D converter 405 by oversampling a few times per chip. This A/D converter 405 may, for example, be part of the RF section 235, rake receiver 245 or other components (not shown). (Thus, signal 240 may always be digital (e.g., if A/D converter 405 is part of RF part 235 (of FIG. 2 )) or may be analog at one point and digital at a later point ( For example, if the A/D converter 405 is part of the rake receiver 245).) (now digital) signal 240 is decimated at a decimation section 410 to produce a decimated signal 415 which is equal to the number of components making up the signal 240 has fewer ingredients than The decimated signal 415 is then applied to a matched filter 420 . Matched filter 420 may be matched to the pilot signal of transmit (group) 225 .

Matched filter 420 may use at least one FIR filter 425 to approximately locate signal path rays. Alternatively, it could eg use a set of correlators or the like. The decimated signal 415 (eg, instead of the (digital) signal 240 ) is provided to the FIR filter 425 in order to reduce the total number of delay elements processed by the FIR filter 425 . The approximate position of the signal path rays can be determined by applying a peak detector 427 to the output of the matched filter 420 . Matched filter 420 for coarse signal matching produces a detected approximate location 460 (eg, output from peak detector 427).

The decimation factor of the decimation section 410 is preferably equal to or less than the oversampling rate of the A/D converter 405 . If the decimation factor is less than the oversampling rate, the FIR matched filter can still detect signal path rays at a resolution higher than the chip resolution. On the other hand, if the decimation factor is equal to or greater than the oversampling rate, the filters detect signal path rays with a resolution equal to or less than the chip resolution, respectively.

Continuing now with searchers 250A and 250B of FIGS. 4A and 4B , respectively, fine signal matching ("Stage 2") is described. Fine signal matching is performed using the approximate positions (groups) of signal path rays detected by coarse signal matching. The approximate position (set) is provided as one or more delayed candidates (as indicated by "D") from the coarse signal match. A code generator generates a pattern of expected data at these approximate positions (sets), and this generated pattern is correlated with the undecimated received data with (extra) sample resolution. In an exemplary embodiment (eg, as illustrated in searcher 250A of FIG. 4A ), by shifting the undecimated received data with (additional) sample resolution and then correlating with the generated pattern, the signal The exact position (group) of the path ray. By comparing the resulting correlation values the precise position of the signal path ray can be determined. In another exemplary embodiment (eg, as illustrated in searcher 250B of FIG. 4B ), the pattern generated by shifting is then correlated with the undecimated received data with (additional) sample resolution, and the signal can be detected The exact position (group) of the path ray. By comparing the resulting correlation values the exact position of the signal path ray can be determined, a selected one (group) of which can be forwarded as output (group).

Continuing now with FIGS. 4A and 4B , a fine signal is performed using the detected approximate positions (sets) 460 (e.g., delay candidates (sets) "D") of signal path rays received from the coarse signal match ("Stage 1"). Signal matching ("Stage 2"). A code generator 435 generates a pattern of expected data as generated code data 440 . Referring now only to FIG. 4A, the detected approximate position (group) 460 ("D") and (additional) sampled signal 240 are applied to shifters 430(D-M/C)...430(D)...430 (D+M/C), which delays the (additional) sampled signal 240 from "-M/C" (eg, by shifting) to "+M/C" units. Element "C", as explained further below with reference to Tables 1-3, relates to (sub)chip resolution. More specifically, in some embodiments "C" is proportional to the inverse of the (sub)chip resolution. For example, if a particular embodiment operates at quarter-chip resolution, then "C" equals four (4) in that particular embodiment. Shifters 430(D)-M/C)...430(D)...430(D+M/C) produce as output shifted (additional) sampled signals 400(D-M/C).. .400(D)...400(D+M/C). The shifted (extra) sampled signals 400(D-M/C)...400(D)...400(D+M/C) and the generated code data 440 are correlated in a correlation element 445. Referring now only to FIG. 4B, the detected approximate position (set) 460 ("D") and generated code data 440 are applied to shifters 430(D-M/C)...430(D)...430( D+M/C), which delays the generated code data 440 from "-M/C" (eg, by shifting) to "+M/C" units. Shifters 430(D-M/C)...430(D)...430(D+M/C) produce as output shifted generated code data 460(D-M/C)...460(D) ...460(D+M/C). The shifted generated code data 460(D-M/C)...460(D)...460(D+M/C) and the (additional) sampled signal 240 are correlated in a correlation element 445.

Continuing now collectively with searchers 250A and 250B of FIGS. 4A and 4B respectively, the correlated values (e.g., shifted (extra) sampled signals 400(D-M/C)...400(D).. .400(D+M/C) and generated code data 440, and shifted generated code data 460(D-M/C)...460(D)...460(D+M/C ) and the (additional) sampled signal 240 are applied to a correlation element 445. More specifically, each of the shifters 430(D-M/C)...430(D)...430(D+M/C) is correlated is a mixer detector 445(D-M/C)'...445(D)'...445(D+M/C)' that receives the correlated values (e.g., it may be a multiplying mixer etc.). Correlation is accomplished by applying the output (groups) of mixer detectors 445(D-M/C)'...445(D)'...445(D+M/C) to A corresponding set of: (i) coherent integrators 445(D-M/C)"...445(D)"...445(D+M/C)" (e.g., each of which may be a low-pass or band-pass suppressable narrow-band filter, etc.); (ii) amplitude generating part 445(D-M/C)...445(D)...445(D+M/C); and (iii) non- Coherent integrator 445(D-M/C)""...445(D)""...445(D+M/C)"".

If n=1, the amplitude generating part 445(D-M/C)...445(D)...445(D+M/C)generates the signal amplitude; if n=2, then generates the amplitude Value squared. The amplitude generation section is used to achieve non-coherent integration by taking away the signal phase. Thus, robust integration can be produced because phase variations in the channel do not affect the result. Such protection from phase changes can be achieved eg by squaring the signal (when n=2) or by generating only the magnitude (when n=1). The latter is advantageously cheaper to implement in terms of silicon area and power consumption (ie, magnitude generation), while the former (ie, squared) advantageously provides slightly better performance. Correlation values 450(D-M/C)...450(D)...450(D+M/C) from non-coherent integrator 445(D-M/C)""...445(D)"".. .445(D+M/C)"" to output. A comparison section 455 selects the highest correlation value from the correlation values 450(D-M/C)...450(D)...450(D+M/C) and uses it as a more precise fine position in line 265 on output. The comparing section 455 may select a correlation value from the correlation values 450(D-M/C)...450(D)...450(D+M/C) having the maximum value. Alternatively, a more complex scheme can be used, for example, to select the best candidate.

Referring now to FIG. 5A , an exemplary higher level diagram of signal path ray detection according to the exemplary embodiment of FIGS. 4A and 4B of the present invention is illustrated generally at 500 . The searchers 500 operate in parallel. Referring now to FIG. 5B , another exemplary higher level diagram of signal path ray detection according to the exemplary embodiment of FIGS. 4A and 4B of the present invention is illustrated generally at 550 . The searchers 550 operate in series. Each searcher 500 and 550 begins with "Stage 1" (as identified above with reference to Figures 4A and 4B) blocks 505 and 555, respectively. Each searcher 500 and 550 includes one or more "Stage 2". It should be noted that "Stage 2" of searchers 500 and 550 need not include comparison portion 455 of searchers 250A and 250B (of FIGS. Sections 515 and 570 are completed.

"Phase 1" blocks 505 and 555 generate a plurality of delay candidates D ₁ ...D _k . The value of "k" can be, for example, five (5) or six (6). In the searcher 500, delay candidates _D1 ... _Dk are generated approximately simultaneously by the "Stage 1" block 505 and sent as a vector to the "Stage 2" (as identified above with reference to Figures 4A and 4B) block 510. "Phase 2" blocks 510(1)...510(k) each produce one output out of a total of "k" outputs which are then compared in comparison section 515 which simultaneously receives delayed candidates D1...Dk as input. In searcher 550, delay candidates _D1 ... _Dk are generated substantially simultaneously by "Stage 1" block 555 and sent as a vector to "Stage 2" block 560. The "Phase 2" block 560 is operated "k" times iteratively (eg, in series). The successively generated "k" outputs of the "Phase 2" block 560 are in memory 565 in locations 1...k, respectively. Since each of these "k" outputs actually includes "2M+1" (sub-)outputs, each storage location 1...k of memory 565 may contain "2M+1" memory slots. The "k" outputs (or, more correctly, the "k*(2M+1)" outputs) are then passed in parallel to the comparison section 570, which simultaneously receives delay candidates D ₁ ... D _k as input.

With respect to both searchers 500 and 550, the "k" from multiple "Stage 2" blocks 510(1)...510(k) or from a single "Stage 2" block 560 (e.g. via memory 565) (or "k*(2M+1)") outputs are compared in comparing sections 515 and 570, respectively. The comparison sections 515 and 570 may select the "k" (or "k*(2M+1 )″) “L” maxima of the outputs, as explained in more detail with reference to Table 3 below. The selected "L" outputs may be used in a rake receiver (eg, rake receiver 245 of FIG. 2) to combine corresponding signal path rays, eg, using MRC.

One exemplary comparison of comparing sections 515 and 570 is now described with reference to Tables 1-3 for purposes of illustration and not limitation. Assume that the intent is to use two (2) "Stage 2" blocks (e.g., two "Stage 2" blocks 510(1) and 510(2) or a single "Stage 2" block 560 that operates twice) to locate two Peak (eg, "L=2"), each "Level 2" block operates at quarter-chip resolution (eg, "C=4"). Considering the case when "M=2" (and thus each "Stage 2" has "2M+1" outputs), the number of correlators and thus outputs per stage is equal to five (5). In Table 1 below, the output of the previous "Stage 1" block 505 or 555 is given as [1, 2]. The resulting output of the two "phase 2" blocks is thus: Phase 2: 1 Phase 2: 2 Correlator 1 D1-2/C=0.5 D2-2/C=1.5 Correlator 2 D1-1/C=0.75 D2-1/C＝1.75 Correlator 3 D1=1.0 D2=2.0 Correlator 4 D1+1/C=1.25 D2+1/C=2.25 Correlator 5 D1+2/C=1.5 D2+2/C=2.5

Table 1 (L=2; M=2; C=4; first stage output [1, 2]).

In another example, consider that the output of the "j"th correlator of the "i"th "Phase 2" block is denoted as OUT(i,j), as in Table 2 below: OUT(1,j) OUT(2,j) Correlator 1 140 120 Correlator 2 121 80 Correlator 3 70 30 Correlator 4 60 20 Correlator 5 120 10

Table 2(OUT("i"th stage 2 block, "j"th correlator)).

The final delay value utilized by the rake receiver can be selected by observing and analyzing these exemplary values. In this example, assume that the goal is to select two (2) (eg, L=2) best delay candidates. There are many possible ways to choose these two (2) best delay candidates. A straightforward approach is to first determine the delay value with the largest output, which is the OUT(1,1) delay candidate with 0.50 chip delay. Thereafter, all outputs closer to half a chip are set equal to zero. Table 3 below reflects this setting to zero: OUT(1,1)=140 OUT(2,1)=120 OUT(1,2)=0 OUT(2, 2) = 80 OUT(1,3)=0 OUT(2,3)=30 OUT(1,4)=60 OUT(2,4)=20 OUT (1, 5) = 120 OUT(2,5)=10

Table 3(OUT("i"th stage 2 block, "j"th correlator)).

From the values in Table 3, the next largest output value is selected, which is the delay candidate for OUT(1,5) and OUT(2,1), where the delay is equal to 1.5 chips. This process can be repeated if more delay candidates are to be determined. In this example, the two "Phase 2" stages overlap at a delay of 1.5 chips. It should be noted that it is possible to avoid this overlap by carefully adjusting the delays when they are provided to the "Stage 2" stage.

It should be understood that the elements of Figures 2 and 4-5 need not be separate physical devices. For example, they may alternatively be logical modules in which respective functions are performed by separate entities, overlapping entities, some combination thereof, and so on. Furthermore, they may also consist of one or more software programs or subroutines running on a general-purpose microprocessor, such as a digital signal processor (DSP), or a specialized processing unit. Other possibilities for implementing the principles of the invention will become apparent to those of ordinary skill in the art upon reading and understanding this disclosure, particularly with reference to Figures 2 and 4-6 and the texts associated therewith.

Referring now to FIG. 6 , an exemplary method in flow chart form of detecting signal path rays in two stages in accordance with the present invention is illustrated generally at 600 . Flowchart 600 begins with the receipt of a signal (block 605). Preferably, the signal can be converted from analog to digital by oversampling multiple times per chip. A coarse signal matching stage follows (block 610). As part of the coarse signal matching stage (block 610), the (additionally) sampled signal is decimated (block 615). The decimated signal may then be applied to a filter (block 620). The filter may for example be a FIR filter which is part of a set of matched filters. The filtered signal may then be applied to a detector to determine the approximate position (set) of the signal path ray (decision 625). The detector can be, for example, a peak detector. It should be understood that while the invention is directed to dividing the search process/means into two phases, at least part of the coarse and fine signal matching phases may occur in parallel.

The fine signal matching stage (block 630) may use the approximate position (group) as a guideline for shifting the at least one value being correlated. In one exemplary embodiment, the undecimated (over) sampled signal may be shifted (block 635), and the shifted undecimated (over) sampled signal may be correlated with the generated code (block 645). The correlation results can then be compared, and the highest correlation value can be selected in order to determine the fine location (group) of signal path rays (block 655). In another exemplary embodiment, the generated code may be shifted (block 640), and the shifted generated code may be correlated to the undecimated (over) sampled signal (block 650). The correlation results can then be compared, and the highest correlation value can be selected in order to determine the fine location (group) of signal path rays (block 660). After determining the fine position (set) of signal path rays as part of the fine signal matching stage (block 630), the fine position (set) of signal path rays may be provided to the rake branch (block 665) for further processing of the received signal.

While preferred embodiments of the method and system of the present invention have been illustrated in the drawings and described in the preceding detailed description, it is to be understood that the invention is not limited to the disclosed embodiments, but is capable of numerous reconfigurations , modifications and substitutions without departing from the spirit and scope of the invention as set forth and defined by the appended claims.

Claims (25)

1. A method for locating a signal path ray in a communication system, comprising the steps of: receive a signal; decimating the signal to produce a decimated signal; processing the decimated signal to generate at least one first position with a first precision; and Processing said received signal and a generated code using said at least one first position with a first precision to produce at least one second position with a second precision, said first precision being less precise than said second precision. 2. The method according to claim 1, further comprising the steps of: sampling said signal in an analog-to-digital conversion multiple times per chip prior to said decimating step; and Wherein, said signal in said extracting step comprises a sampled signal. 3. The method of claim 1, wherein said communication system comprises a wireless Code Division Multiple Access (CDMA) communication system. 4. The method according to claim 1, wherein said step of processing said decimated signal to generate at least one first position comprises: applying said decimated signal to at least one filter to generate said at least one first position location steps. 5. The method according to claim 4, wherein said step of applying said decimated signal to at least one filter to generate said at least one first position comprises: applying said decimated signal to at least one matched filter at least one finite impulse response (FIR) filter of the filter. 6. The method according to claim 4, wherein said step of processing said decimated signal to generate at least one first position further comprises: applying an output of said at least one filter to a peak detector in order to determine said The step of at least one first position. 7. The method of claim 1 , wherein said step of processing said signal and a generated code using said at least one first location to generate at least one second location comprises: responding to said at least one first location The step of shifting one of said signal and said generated code to create a shifted variable and a non-shifted variable. 8. The method according to claim 7, wherein said step of processing said signal and a generated code using said at least one first position to generate at least one second position further comprises: combining said shifted variable with The step of correlating said non-shifted variables to produce a plurality of correlated values. 9. The method according to claim 8, wherein said step of processing said signal and a generated code using said at least one first location to generate at least one second location further comprises: comparing said plurality of correlation values The step of comparing to select said at least one second location. 10. The method of claim 8, wherein said shifted variable comprises said received signal and said non-shifted variable comprises said generated code. 11. The method of claim 8, wherein said shifted variable comprises said generated code and said non-shifted variable comprises said received signal. 12. The method according to claim 1, further comprising the step of forwarding said at least one second position to a rake branch in order to initiate a subsequent maximum ratio combining (MRC) of said received signals. 13. A receiver system for locating a signal path ray in a communication system, comprising: a decimation section that decimates a signal according to the decimation factor; at least one filter connected to said decimation section, said at least one filter being associated with determining a first position of said signal with a first precision; a code generator section adapted to generate at least one pattern; at least one shifter connected to said at least one filter to receive said first position; at least one correlator that correlates a version of the signal with a version of the at least one code pattern; at least one comparison section that selects a highest value from the output associated values, and the second position output from said comparison section consists of said highest value or a related value; and wherein the first accuracy of the first location is less precise than a second accuracy of the second location. 14. The receiver system according to claim 13, wherein said shifter shifts said signal, said version of said signal is a shifted version of said signal, and said at least one pattern of The version is a non-shifted version of the at least one pattern. 15. The receiver system according to claim 13, wherein said shifter shifts said at least one pattern, said version of said signal is a non-shifted version of said signal, and said at least The version of a pattern is a shifted version of the at least one pattern. 16. The receiver system according to claim 13, further comprising: an analog-to-digital converter, said analog-to-digital converter converting said signal to a digitally sampled signal before said decimating section decimates said signal. 17. The receiver system of claim 16, wherein the sampling rate of said analog-to-digital converter is such that an analog version of said signal is sampled multiple times per chip. 18. The receiver system of claim 17, wherein said sampling rate and said decimation factor are at least in part determinants of the accuracy of said first location. 19. The receiver system according to claim 13, further comprising a peak detector; and Wherein said at least one filter comprises a plurality of matched filters, said plurality of matched filters comprises at least one finite impulse response (FIR) filter, one input of said peak detector is controlled by said at least one FIR filter consists of an output of the peak detector, and the first position consists of an output of the peak detector. 20. The receiver system of claim 13, wherein said at least one correlator comprises a plurality of correlators, each of said plurality of correlators comprising a multiplicative mixer and an integrator. 21. The receiver system of claim 13, wherein said communication system comprises a wireless Code Division Multiple Access (CDMA) communication system. 22. The receiver system according to claim 13 , further comprising: a comparison section and a plurality of rake branches, said comparison section receiving at least one output from said at least one correlator and sending to said plurality of rake branches At least one of the provides a second location. 23. A method for searching a signal path ray in a Code Division Multiple Access (CDMA) communication system, comprising the steps of: receive a signal; determining a rough location of said signal with a first precision; determining a fine location of the signal based at least in part on the coarse location, wherein the fine location has a second precision, the first precision being less precise than the second precision; and The fine location is provided to the rake branch. 24. A method according to claim 23, wherein said step of determining said coarse position of said signal comprises the step of decimating said signal, said signal having been oversampled. 25. The method of claim 23, wherein said step of determining said fine location of said signal based at least in part on said coarse location comprises the step of: generate a pattern; shifting in response to the coarse position; correlating said pattern with said signal, at least one of said pattern and said signal having been shifted in said shifting step; and The fine location is selected in response to the correlating step.