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CN117977922A - Driving control method of high power factor high frequency quasi-resonant three-phase AC/DC converter - Google Patents

Driving control method of high power factor high frequency quasi-resonant three-phase AC/DC converter Download PDF

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CN117977922A
CN117977922A CN202410141734.5A CN202410141734A CN117977922A CN 117977922 A CN117977922 A CN 117977922A CN 202410141734 A CN202410141734 A CN 202410141734A CN 117977922 A CN117977922 A CN 117977922A
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frequency
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output
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phase
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初中原
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JIANGSU EASTONE TECHNOLOGY CO LTD
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JIANGSU EASTONE TECHNOLOGY CO LTD
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0012—Control circuits using digital or numerical techniques
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0067—Converter structures employing plural converter units, other than for parallel operation of the units on a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/084—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters using a control circuit common to several phases of a multi-phase system
    • H02M1/0845—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters using a control circuit common to several phases of a multi-phase system digitally controlled (or with digital control)
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/08—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters
    • H02M1/088—Circuits specially adapted for the generation of control voltages for semiconductor devices incorporated in static converters for the simultaneous control of series or parallel connected semiconductor devices
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4225—Arrangements for improving power factor of AC input using a non-isolated boost converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4241—Arrangements for improving power factor of AC input using a resonant converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/02—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC
    • H02M5/04—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters
    • H02M5/22—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M5/225—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode comprising two stages of AC-AC conversion, e.g. having a high frequency intermediate link
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/02—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC
    • H02M5/04—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters
    • H02M5/22—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M5/275—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M5/293—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M5/2932—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage, current or power
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/02—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC
    • H02M5/04—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters
    • H02M5/22—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M5/275—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M5/297—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal for conversion of frequency
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/02—Conversion of AC power input into DC power output without possibility of reversal
    • H02M7/04—Conversion of AC power input into DC power output without possibility of reversal by static converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/02—Conversion of AC power input into DC power output without possibility of reversal
    • H02M7/04—Conversion of AC power input into DC power output without possibility of reversal by static converters
    • H02M7/12—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/02—Conversion of AC power input into DC power output without possibility of reversal
    • H02M7/04—Conversion of AC power input into DC power output without possibility of reversal by static converters
    • H02M7/12—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/2173—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a biphase or polyphase circuit arrangement
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/02—Conversion of AC power input into DC power output without possibility of reversal
    • H02M7/04—Conversion of AC power input into DC power output without possibility of reversal by static converters
    • H02M7/12—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Rectifiers (AREA)

Abstract

The invention discloses a driving control method of a high-power factor high-frequency quasi-resonance three-phase alternating-current-direct-current converter, which comprises the following steps: acquiring a line voltage sampling value between a first phase voltage input end and a second phase voltage input end; acquiring a first sampling current of a power frequency AC-high frequency AC circulating inverter input by a first phase voltage; acquiring a synchronous pulse according to a second sampling current in the high-frequency cyclic power factor corrector; four high-frequency switching tubes in the power frequency AC-high-frequency AC circulating inverter are controlled by four AND gates; obtaining the output of the PFC PWM pulse width modulator according to the output of the boost inductor current regulator, the synchronous pulse and the output of the pulse oscillator; and controlling the switching states of two high-frequency switching tubes in the high-frequency circulating power factor corrector according to the output of the PFC PWM pulse width modulator. The invention provides a stable switching tube driving method for the high-power factor high-frequency quasi-resonance three-phase alternating-current-direct-current converter.

Description

高功率因数高频准谐振三相交直流变换器的驱动控制方法Driving control method of high power factor high frequency quasi-resonant three-phase AC/DC converter

技术领域Technical Field

本发明涉及变换器驱动控制技术领域，具体涉及一种用于高功率因数高频准谐振三相交直流变换器的驱动控制方法。The invention relates to the technical field of converter drive control, and in particular to a drive control method for a high power factor, high frequency, quasi-resonant three-phase AC/DC converter.

背景技术Background technique

三相AC/DC变换器拓扑结构的所有功率变换器及其衍生产品，包括开关电源，应用领域非常广泛，例如单晶炉电源，电解电源，充电电源等。All power converters with three-phase AC/DC converter topology and their derivatives, including switching power supplies, have a wide range of applications, such as single crystal furnace power supplies, electrolytic power supplies, charging power supplies, etc.

三相电压型PWM高频整流电路由六个功率开关管组成，也可看成是多个升压(Boost)电路的组合，因此是升压型整流电路，其输出直流电压从交流电源电压的峰值附近向高调节，如果向低调节会使输入电流波型失真，功率因数降低，甚至不能工作，因此要求开关管的耐压要高于三相线电压峰值的3～4倍。The three-phase voltage-type PWM high-frequency rectifier circuit is composed of six power switch tubes, which can also be regarded as a combination of multiple boost circuits. Therefore, it is a boost rectifier circuit. Its output DC voltage is adjusted upward from the peak value of the AC power supply voltage. If it is adjusted downward, the input current waveform will be distorted, the power factor will be reduced, and it may even fail to work. Therefore, the withstand voltage of the switch tube is required to be 3 to 4 times higher than the peak value of the three-phase line voltage.

因此，如何提供一种三相交直流变换器的驱动控制方法，以降低现有变换器驱动方法对开关管的耐压值要求，从而提高开关管使用寿命，成为目前亟待解决的问题。Therefore, how to provide a driving control method for a three-phase AC/DC converter to reduce the withstand voltage requirements of the existing converter driving method on the switch tube, thereby increasing the service life of the switch tube, has become a problem that needs to be solved urgently.

发明内容Summary of the invention

有鉴于此，本发明实施例提供了一种用于高功率因数高频准谐振三相交直流变换器的驱动控制方法，以解决现有技术中变换器驱动方法对开关管的耐压值要求高，导致开关管使用寿命不够长，需要经常维护更换的问题。In view of this, an embodiment of the present invention provides a driving control method for a high power factor, high frequency, quasi-resonant three-phase AC/DC converter to solve the problem that the converter driving method in the prior art has high requirements on the withstand voltage of the switching tube, resulting in the switching tube having a short service life and requiring frequent maintenance and replacement.

本发明实施例提供了一种高功率因数高频准谐振三相交直流变换器的驱动控制方法，包括：The embodiment of the present invention provides a driving control method for a high power factor high frequency quasi-resonant three-phase AC/DC converter, comprising:

获取第一相电压输入端和第二相电压输入端之间的线电压采样值；Acquire a line voltage sampling value between the first phase voltage input terminal and the second phase voltage input terminal;

获取第一相电压输入的工频AC-高频AC循环逆变器的第一采样电流；Acquire a first sampled current of an industrial frequency AC-high frequency AC cycle inverter inputted with a first phase voltage;

根据高频循环功率因数校正器中经过升压电感的第二采样电流获取同步脉冲；Acquiring a synchronization pulse according to a second sampled current passing through a boost inductor in a high frequency cycle power factor corrector;

将线电压采样值、第一采样电流、同步脉冲作为四个与门的间接控制输入，通过四个与门对工频AC-高频AC循环逆变器中的四个高频开关管进行控制；The line voltage sampling value, the first sampling current, and the synchronization pulse are used as indirect control inputs of four AND gates, and four high-frequency switching tubes in the industrial frequency AC-high frequency AC cycle inverter are controlled through the four AND gates;

将直流输出电压与直流输出给定值之间的差值、线电压采样值和第一采样电流这三个值进行相乘后，再与第二采样电流作差，作为升压电感电流调节器的输入；The difference between the DC output voltage and the DC output given value, the line voltage sampling value and the first sampling current are multiplied together, and then the difference is made with the second sampling current as the input of the boost inductor current regulator;

根据升压电感电流调节器的输出、同步脉冲和脉冲振荡器的输出，得到PFC PWM脉宽调制器的输出；The output of the PFC PWM pulse width modulator is obtained according to the output of the boost inductor current regulator, the synchronization pulse and the output of the pulse oscillator;

根据PFC PWM脉宽调制器的输出控制高频循环功率因数校正器中的两个高频开关管的开关状态；Controlling the switching states of two high-frequency switching tubes in the high-frequency cycle power factor corrector according to the output of the PFC PWM pulse width modulator;

其中，高功率因数高频准谐振三相交直流变换器包括三个单相线电压输入电路；每一相线电压输入电路包括：通过工频AC-高频AC循环逆变器得到高频交流电压；高频交流电压经过高频变压器升压；再经过高频循环功率因数校正器变换为直流电压。Among them, the high power factor high frequency quasi-resonant three-phase AC/DC converter includes three single-phase line voltage input circuits; each phase line voltage input circuit includes: obtaining a high-frequency AC voltage through an industrial frequency AC-high frequency AC cycle inverter; the high-frequency AC voltage is boosted by a high-frequency transformer; and then converted into a DC voltage by a high-frequency cycle power factor corrector.

可选地，基准电压为接地电压。Optionally, the reference voltage is a ground voltage.

可选地，根据高频循环功率因数校正器中经过升压电感的第二采样电流获取同步脉冲，包括：Optionally, obtaining a synchronization pulse according to a second sampled current passing through a boost inductor in a high frequency cycle power factor corrector comprises:

将第二采样电流与基准电压通过第一比较器进行比较后输出，得到同步脉冲。The second sampling current is compared with the reference voltage through the first comparator and then output to obtain a synchronization pulse.

可选地，将线电压采样值、第一采样电流、同步脉冲作为四个与门的间接控制输入，通过四个与门对工频AC-高频AC循环逆变器中的四个高频开关管进行控制，包括：Optionally, the line voltage sampling value, the first sampling current, and the synchronization pulse are used as indirect control inputs of four AND gates, and four high-frequency switching tubes in the industrial frequency AC-high frequency AC cycle inverter are controlled through the four AND gates, including:

将线电压采样值和基准电压分别作为第二比较器的正端输入和负端输入；将线电压采样值和基准电压分别作为第三比较器的负端输入和正端输入；The line voltage sampling value and the reference voltage are used as the positive terminal input and the negative terminal input of the second comparator respectively; the line voltage sampling value and the reference voltage are used as the negative terminal input and the positive terminal input of the third comparator respectively;

第二比较器的输出分别作为第一与门的第一输入和第二与门的第一输入；The output of the second comparator is used as the first input of the first AND gate and the first input of the second AND gate respectively;

第三比较器的输出分别作为第三与门的第一输入和第四与门的第一输入；The output of the third comparator is respectively used as the first input of the third AND gate and the first input of the fourth AND gate;

将第一采样电流与设定值通过第四比较器进行比较后输出，作为逆变PWM脉宽调制模块的第一输入；脉冲振荡器产生的脉冲作为逆变PWM脉宽调制模块的第二输入；逆变PWM脉宽调制模块的第一输出分别作为第一与门的第二输入和第三与门的第二输入；逆变PWM脉宽调制模块的第二输出分别作为第二与门的第二输入和第三与门的第二输入；The first sampled current is compared with the set value by the fourth comparator and then output as the first input of the inverter PWM pulse width modulation module; the pulse generated by the pulse oscillator is used as the second input of the inverter PWM pulse width modulation module; the first output of the inverter PWM pulse width modulation module is used as the second input of the first AND gate and the second input of the third AND gate respectively; the second output of the inverter PWM pulse width modulation module is used as the second input of the second AND gate and the second input of the third AND gate respectively;

使系统过压过流保护电路在起动状态输出高电平，作为第一与门的第三输入、第二与门的第三输入、第三与门的第三输入和第四与门的第三输入。The system overvoltage and overcurrent protection circuit is made to output a high level in the starting state as the third input of the first AND gate, the third input of the second AND gate, the third input of the third AND gate and the third input of the fourth AND gate.

可选地，还包括：将同步脉冲输入至脉冲振荡器。Optionally, the method further includes: inputting a synchronization pulse to a pulse oscillator.

可选地，在将直流输出电压与直流输出给定值相减后，通过电压调节器进行PI运算，输出到乘法器；乘法器接收线电压采样值和第一采样电流，对三个值进行相乘输出，作为升压电感电流的动态设定值。Optionally, after subtracting the DC output voltage from the DC output given value, a PI operation is performed through the voltage regulator and output to the multiplier; the multiplier receives the line voltage sampling value and the first sampling current, multiplies the three values and outputs them as the dynamic setting value of the boost inductor current.

可选地，将动态设定值与第二采样电流相减后，通过升压电感电流调节器进行PI运算，以控制PFC PWM脉宽调制器的脉冲宽度。Optionally, after subtracting the dynamic setting value from the second sampled current, a PI operation is performed through the boost inductor current regulator to control the pulse width of the PFC PWM pulse width modulator.

可选地，还包括：当乘法器输出的动态设定值到达动态设置值峰值，动态设定值瞬间下降，升压电感电流调节器的输出快速下降，PFC PWM脉宽调制器的输出脉冲快速变为零。Optionally, it also includes: when the dynamic setting value output by the multiplier reaches the dynamic setting value peak value, the dynamic setting value drops instantly, the output of the boost inductor current regulator drops rapidly, and the output pulse of the PFC PWM pulse width modulator quickly becomes zero.

可选地，还包括：由两个运算放大器、六个电阻和两个二极管组成的精密整流电路对第一采样电流或第二采样电流进行信号放大。Optionally, the method further includes: a precision rectifier circuit composed of two operational amplifiers, six resistors and two diodes for amplifying the signal of the first sampling current or the second sampling current.

本发明的有益效果：Beneficial effects of the present invention:

将三相分成三个单相线电压，第一级将工频交流通过循环逆变器直接将工频交流变换成高频交流，中间没有直流环节，第二级通过高频变压器和高频循环功率因数校正器变换成直流电压，同时进行功率因数校正，功率因数校正中间也没有直流环节，三个独立的直流电压并联后经过π型滤波输出。该电路拓扑结构只有两级开关电路，中间去掉两个直流环节，同时采用高频准谐振方式，可有效减小开关损耗。另外功率因数校正环节放在直流输出前，开关管上的最大耐压就是输出电压。本发明实施例为这种变换器提供了驱动方法，保证其稳定运行。The three phases are divided into three single-phase line voltages. The first stage directly converts the industrial frequency AC into high-frequency AC through a cyclic inverter, without a DC link in the middle. The second stage converts it into a DC voltage through a high-frequency transformer and a high-frequency cyclic power factor corrector, and performs power factor correction at the same time. There is no DC link in the power factor correction. The three independent DC voltages are connected in parallel and output through a π-type filter. This circuit topology has only two switching circuits, two DC links are removed in the middle, and a high-frequency quasi-resonance method is used, which can effectively reduce switching losses. In addition, the power factor correction link is placed before the DC output, and the maximum withstand voltage on the switch tube is the output voltage. The embodiment of the present invention provides a driving method for this converter to ensure its stable operation.

附图说明BRIEF DESCRIPTION OF THE DRAWINGS

通过参考附图会更加清楚的理解本发明的特征和优点，附图是示意性的而不应理解为对本发明进行任何限制，在附图中：The features and advantages of the present invention will be more clearly understood by referring to the accompanying drawings, which are schematic and should not be construed as limiting the present invention in any way. In the accompanying drawings:

图1示出了本发明实施例中一种AB相、BC相、CA相逆变桥PWM控制电路图；FIG1 shows a PWM control circuit diagram of an AB phase, BC phase, and CA phase inverter bridge in an embodiment of the present invention;

图2示出了本发明实施例中一种AB相、BC相、CA相PFC PWM控制电路图；FIG2 shows a PFC PWM control circuit diagram of AB phase, BC phase, and CA phase in an embodiment of the present invention;

图3示出了本发明实施例中一种基于高功率因数高频准谐振的三相交直流变换器的结构图；FIG3 shows a structural diagram of a three-phase AC-DC converter based on high power factor and high frequency quasi-resonance in an embodiment of the present invention;

图4示出了本发明实施例中一种基于高功率因数高频准谐振的三相交直流变换器的DSP模块接线图；FIG4 shows a DSP module wiring diagram of a three-phase AC-DC converter based on high power factor and high frequency quasi-resonance in an embodiment of the present invention;

图5示出了本发明实施例中一种三相准谐振电流采样电路图；FIG5 shows a three-phase quasi-resonant current sampling circuit diagram in an embodiment of the present invention;

图6示出了本发明实施例中一种三相升压电感电流采样电路图；FIG6 shows a three-phase boost inductor current sampling circuit diagram in an embodiment of the present invention;

图7示出了本发明实施例中一种AB相工频交流输入电压正半波、准谐振电流正半波、升压电感储能等效电路图；FIG7 shows an equivalent circuit diagram of an AB phase power frequency AC input voltage positive half-wave, a quasi-resonant current positive half-wave, and a boost inductor energy storage in an embodiment of the present invention;

图8示出了本发明实施例中一种AB相工频交流输入电压正半波、准谐振电流正半波、升压电感放能等效电路图；FIG8 shows an equivalent circuit diagram of an AB phase power frequency AC input voltage positive half-wave, a quasi-resonant current positive half-wave, and a boost inductor energy discharge in an embodiment of the present invention;

图9示出了本发明实施例中一种第一高频电容放电等效电路图；FIG9 shows a first high-frequency capacitor discharge equivalent circuit diagram in an embodiment of the present invention;

图10示出了本发明实施例中一种AB相工频交流输入电压正半波、准谐振电流负半波、升压电感储能等效电路图；FIG10 shows an equivalent circuit diagram of an AB phase power frequency AC input voltage positive half-wave, a quasi-resonant current negative half-wave, and a boost inductor energy storage in an embodiment of the present invention;

图11示出了本发明实施例中一种AB相工频交流输入电压正半波、准谐振电流负半波、升压电感放能等效电路图；FIG11 shows an equivalent circuit diagram of an AB phase power frequency AC input voltage positive half-wave, a quasi-resonant current negative half-wave, and a boost inductor energy discharge in an embodiment of the present invention;

图12示出了本发明实施例中一种第二高频电容放电等效电路图；FIG12 shows a second high-frequency capacitor discharge equivalent circuit diagram according to an embodiment of the present invention;

图13示出了本发明实施例中一种AB相工频交流输入电压负半波、准谐振电流正半波、升压电感储能等效电路图；FIG13 shows an equivalent circuit diagram of an AB phase power frequency AC input voltage negative half-wave, a quasi-resonant current positive half-wave, and a boost inductor energy storage in an embodiment of the present invention;

图14示出了本发明实施例中一种AB相工频交流输入电压负半波、准谐振电流正半波、升压电感放能等效电路图；FIG14 shows an equivalent circuit diagram of an AB phase power frequency AC input voltage negative half-wave, a quasi-resonant current positive half-wave, and a boost inductor energy release in an embodiment of the present invention;

图15示出了本发明实施例中一种第二高频电容放电等效电路图；FIG15 shows a second high-frequency capacitor discharge equivalent circuit diagram according to an embodiment of the present invention;

图16示出了本发明实施例中一种AB相工频交流输入电压负半波、准谐振电流负半波、升压电感储能等效电路图；FIG16 shows an equivalent circuit diagram of an AB phase power frequency AC input voltage negative half-wave, a quasi-resonant current negative half-wave, and a boost inductor energy storage in an embodiment of the present invention;

图17示出了本发明实施例中一种AB相工频交流输入电压负半波、准谐振电流负半波、升压电感放能等效电路图；FIG17 shows an equivalent circuit diagram of an AB phase power frequency AC input voltage negative half-wave, a quasi-resonant current negative half-wave, and boost inductor energy release in an embodiment of the present invention;

图18示出了本发明实施例中一种第一高频电容放电等效电路图；FIG18 shows a first high-frequency capacitor discharge equivalent circuit diagram in an embodiment of the present invention;

图19示出了本发明实施例中一种基于高功率因数高频准谐振的三相交直流变换器的AB相主电路的主要工作波形图；FIG19 shows a main operating waveform diagram of the AB phase main circuit of a three-phase AC/DC converter based on high power factor and high frequency quasi-resonance in an embodiment of the present invention;

附图标记：Reference numerals:

1110、2110、3110：输入滤波电路；1111、2111、3111：滤波电容；1112、2112、3112：共模电感；1113、2113、3113：滤波电容；1114、2114、3114：LV电压传感器；1120、2120、3120：工频AC-高频AC循环逆变器；1121、1122、1123、1124、2121、2122、2123、2124、3121、3122、3123、3124：带有寄生二极管的高频开关管；1125、1126、2125、2126、3125、3126：高频电容；1130、2130、3130：高频变压模块；1131、2131、3131：高频变压器；1132、2132、3132：准谐振电容；1133、2133、3133：高频电流互感器；1140、2140、3140：高频循环功率因数校正器；1141、1142、2141、2142、3141、3142：带有寄生二极管的高频开关管；1143、1144、2143、2144、3143、3144：高频二极管；1145、2145、3145：升压电感；150：π型滤波电路；151、153：滤波电容；152：滤波电感；154：LV电压传感器；155：LA电流传感器；410、420、430：交流输入电压采样模块；440、450、460：高频逆变电流采样模块；47-、480、490：升压电感电流采样模块；500：输出直流电压采样模块；510：输出直流电流采样模块；520：ADC接口；1530、2530、3530：单相线电压逆变桥PWM控制模块；1540、2540、3540：变压器次级循环PFC控制模块；550：逆变桥控制PWM输出接口；560：循环PFC控制PWM输出接口；570：逆变桥驱动电路；580：PFC驱动电路；1531、2531、3531：脉冲振荡器；1532、1534、1535、2532、2534、2535、3532、3534、3535：比较器；1533、2533、3533：逆变PWM脉宽调制模块；1536、2536、3536：系统过压过流保护；1537、2537、3537：与门逻辑电路；441、442、451、452、461、462、471、472、481、482、491、492：运算放大器；R441、R442、R443、R444、R445、Ria1、R451、R452、R453、R454、R455、Rib1、R461、R462、R463、R464、R465、Ric1、R471、R472、R473、R474、R475、Ria2、R481、R482、R483、R484、R485、Rib2、R491、R492、R493、R494、R495、Ric2：电阻；D441、D442、D451、D452、D461、D462、D471、D472、D481、D482、D491、D492：二极管；1541、2541、3541：输出电压调节器；1542、2542、3542：乘法器；1543、2543、3543：升压电感电流调节器；1544、1545、2544、2545、3544、3545：比较器；1546、2546、3546：PFC PWM脉宽调制器。1110, 2110, 3110: input filter circuit; 1111, 2111, 3111: filter capacitor; 1112, 2112, 3112: common mode inductor; 1113, 2113, 3113: filter capacitor; 1114, 2114, 3114: LV voltage sensor; 1120, 2120, 3120: industrial frequency AC-high frequency AC cycle inverter; 1121, 1122, 1123, 1124, 2121, 212 2. 2123, 2124, 3121, 3122, 3123, 3124: high-frequency switch tube with parasitic diode; 1125, 1126, 2125, 2126, 3125, 3126: high-frequency capacitor; 1130, 2130, 3130: high-frequency transformer module; 1131, 2131, 3131: high-frequency transformer; 1132, 2132, 3132: quasi-resonant capacitor; 1133, 2133, 3133: high-frequency current Transformer; 1140, 2140, 3140: high-frequency cycle power factor corrector; 1141, 1142, 2141, 2142, 3141, 3142: high-frequency switch tube with parasitic diode; 1143, 1144, 2143, 2144, 3143, 3144: high-frequency diode; 1145, 2145, 3145: boost inductor; 150: π-type filter circuit; 151, 153: filter capacitor; 152: filter inductor ; 154: LV voltage sensor; 155: LA current sensor; 410, 420, 430: AC input voltage sampling module; 440, 450, 460: high-frequency inverter current sampling module; 47-, 480, 490: boost inductor current sampling module; 500: output DC voltage sampling module; 510: output DC current sampling module; 520: ADC interface; 1530, 2530, 3530: single-phase line voltage inverter bridge PWM control module ; 1540, 2540, 3540: transformer secondary cycle PFC control module; 550: inverter bridge control PWM output interface; 560: cycle PFC control PWM output interface; 570: inverter bridge drive circuit; 580: PFC drive circuit; 1531, 2531, 3531: pulse oscillator; 1532, 1534, 1535, 2532, 2534, 2535, 3532, 3534, 3535: comparator; 15 33, 2533, 3533: inverter PWM pulse width modulation module; 1536, 2536, 3536: system overvoltage and overcurrent protection; 1537, 2537, 3537: AND gate logic circuit; 441, 442, 451, 452, 461, 462, 471, 472, 481, 482, 491, 492: operational amplifier; R441, R442, R443, R444, R445, Ria1, R451, R452 , R453, R454, R455, Rib1, R461, R462, R463, R464, R465, Ric1, R471, R472, R473, R474, R475, Ria2, R481, R482, R483, R484, R485, Rib2, R491, R492, R493, R494, R495, Ric2: resistor; D441, D442, D451, D452 , D461, D462, D471, D472, D481, D482, D491, D492: diodes; 1541, 2541, 3541: output voltage regulator; 1542, 2542, 3542: multiplier; 1543, 2543, 3543: boost inductor current regulator; 1544, 1545, 2544, 2545, 3544, 3545: comparator; 1546, 2546, 3546: PFC PWM pulse width modulator.

具体实施方式Detailed ways

为使本发明实施例的目的、技术方案和优点更加清楚，下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例是本发明一部分实施例，而不是全部的实施例。基于本发明中的实施例，本领域技术人员在没有作出创造性劳动前提下所获得的所有其他实施例，都属于本发明保护的范围。In order to make the purpose, technical solution and advantages of the embodiments of the present invention clearer, the technical solution in the embodiments of the present invention will be clearly and completely described below in conjunction with the drawings in the embodiments of the present invention. Obviously, the described embodiments are part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative work are within the scope of protection of the present invention.

如图1～4所示，本发明实施例提供了一种用于高功率因数高频准谐振三相交直流变换器的驱动控制方法，其中，高功率因数高频准谐振三相交直流变换器的结构如图3所示，包括三个单相线电压输入电路；每一相线电压输入电路包括：通过工频AC-高频AC循环逆变器得到高频交流电压；高频交流电压经过高频变压器升压；再经过高频循环功率因数校正器变换为直流电压。每一相电路中工频AC-高频AC循环逆变器的高频开关管与高频循环功率因数校正器中的高频开关管驱动方法都相同，不同点在于线电压的选取。As shown in Figures 1 to 4, an embodiment of the present invention provides a driving control method for a high power factor high frequency quasi-resonant three-phase AC/DC converter, wherein the structure of the high power factor high frequency quasi-resonant three-phase AC/DC converter is shown in Figure 3, including three single-phase line voltage input circuits; each phase line voltage input circuit includes: obtaining a high frequency AC voltage through an industrial frequency AC-high frequency AC cycle inverter; the high frequency AC voltage is boosted by a high frequency transformer; and then converted into a DC voltage by a high frequency cycle power factor corrector. The high frequency switch tube driving method of the industrial frequency AC-high frequency AC cycle inverter and the high frequency switch tube in the high frequency cycle power factor corrector in each phase circuit are the same, and the difference lies in the selection of line voltage.

以下按ua相输入为例，驱动控制方法包括：Taking u a phase input as an example, the drive control method includes:

S1，获取A相电压输入端和B相电压输入端之间的线电压采样值uabi。S1, obtaining a line voltage sampling value u abi between the A-phase voltage input terminal and the B-phase voltage input terminal.

如图3所示，在ua和ub两路输入之间，设置LV电压传感器，获取线电压采样值uabi。As shown in FIG3 , an LV voltage sensor is set between the two inputs u a and u b to obtain the line voltage sampling value u abi .

S2，获取A相电压输入的工频AC-高频AC循环逆变器的第一采样电流ia1～。S2, obtaining a first sampling current i a1~ of the industrial frequency AC-high frequency AC cycle inverter with A phase voltage input.

如图5和图6所示，由两个运算放大器、六个电阻和两个二极管组成的精密整流电路，对采样电流进行放大。As shown in FIG5 and FIG6 , a precision rectifier circuit composed of two operational amplifiers, six resistors and two diodes amplifies the sampled current.

S3，根据高频循环功率因数校正器中经过升压电感的第二采样电流ia2～获取同步脉冲。S3, obtaining a synchronization pulse according to a second sampling current ia2~ passing through the boost inductor in the high frequency cycle power factor corrector.

如图2所示，将第二采样电流与基准电压通过第一比较器1544进行比较后输出，得到同步脉冲Psy。As shown in FIG. 2 , the second sampling current is compared with the reference voltage by the first comparator 1544 and then output to obtain a synchronization pulse P sy .

在具体实施例中，通过图4所示的DSP处理器及周围电路，将模拟信号转换为数字信号，例如ia1～对应Ia1-，ia2～对应Ia2-。In a specific embodiment, the analog signal is converted into a digital signal through the DSP processor and surrounding circuits shown in FIG. 4 , for example, i a1~ corresponds to I a1 −, and i a2~ corresponds to I a2− .

S4，将线电压采样值uabi、第一采样电流Ia1-、同步脉冲Psy作为四个与门的间接控制输入，通过四个与门对工频AC-高频AC循环逆变器中的四个高频开关管进行控制。S4, taking the line voltage sampling value u abi , the first sampling current I a1 -, and the synchronization pulse P sy as indirect control inputs of four AND gates, and controlling four high-frequency switching tubes in the industrial frequency AC-high frequency AC cycle inverter through the four AND gates.

如图1所示，将线电压采样值uabi和基准电压GND分别作为第二比较器1534的正端输入和负端输入；将线电压采样值uabi和基准电压GND分别作为第三比较器1535的负端输入和正端输入；As shown in FIG1 , the line voltage sampling value u abi and the reference voltage GND are used as the positive input and negative input of the second comparator 1534 respectively; the line voltage sampling value u abi and the reference voltage GND are used as the negative input and positive input of the third comparator 1535 respectively;

第二比较器1534的输出分别作为第一与门1的第一输入和第二与门2的第一输入；The output of the second comparator 1534 is used as the first input of the first AND gate 1 and the first input of the second AND gate 2 respectively;

第三比较器1535的输出分别作为第三与门3的第一输入和第四与门4的第一输入；The output of the third comparator 1535 is used as the first input of the third AND gate 3 and the first input of the fourth AND gate 4 respectively;

将第一采样电流Ia1-与设定值Ia1s通过第四比较器1532进行比较后输出，作为逆变PWM脉宽调制模块1533的第一输入；脉冲振荡器1531产生的脉冲作为逆变PWM脉宽调制模块1533的第二输入；逆变PWM脉宽调制模块1533的第一输出分别作为第一与门1的第二输入和第三与门3的第二输入；逆变PWM脉宽调制模块1533的第二输出分别作为第二与门2的第二输入和第三与门3的第二输入；The first sampling current I a1 - is compared with the set value I a1s by the fourth comparator 1532 and then output as the first input of the inverter PWM pulse width modulation module 1533; the pulse generated by the pulse oscillator 1531 is used as the second input of the inverter PWM pulse width modulation module 1533; the first output of the inverter PWM pulse width modulation module 1533 is used as the second input of the first AND gate 1 and the second input of the third AND gate 3; the second output of the inverter PWM pulse width modulation module 1533 is used as the second input of the second AND gate 2 and the second input of the third AND gate 3;

使系统过压过流保护电路1536在起动状态输出高电平，作为第一与门1的第三输入、第二与门2的第三输入、第三与门3的第三输入和第四与门4的第三输入。The system overvoltage and overcurrent protection circuit 1536 is made to output a high level in the starting state, which serves as the third input of the first AND gate 1, the third input of the second AND gate 2, the third input of the third AND gate 3 and the third input of the fourth AND gate 4.

S5，将直流输出电压Uoc与直流输出给定值U* oc之间的差值、线电压采样值uabi和第一采样电流Ia1-这三个值进行相乘后，再与第二采样电流Ia2-作差，作为升压电感电流调节器1543的输入。S5, multiply the difference between the DC output voltage U oc and the DC output given value U * oc , the line voltage sampling value u abi and the first sampling current I a1- , and then subtract them from the second sampling current I a2- as the input of the boost inductor current regulator 1543.

如图2所示，在将直流输出电压Uoc与直流输出给定值U* oc相减后，通过输出电压调节器1541进行PI运算，输出到乘法器1542；乘法器1542接收线电压采样值uabi和第一采样电流Ia1-，对三个值进行相乘输出，作为升压电感电流的动态设定值。As shown in FIG2 , after the DC output voltage U oc is subtracted from the DC output given value U * oc , a PI operation is performed through the output voltage regulator 1541 and the result is output to the multiplier 1542; the multiplier 1542 receives the line voltage sampling value u abi and the first sampling current I a1- , multiplies the three values and outputs them as the dynamic setting value of the boost inductor current.

S6，根据升压电感电流调节器1543的输出、同步脉冲Psy和脉冲振荡器的输出Fp和第二采样电流Ia2-同基准电压Ia2-LMT的比较结果，得到PFC PWM脉宽调制器的输出。基准电压Ia2-LMT为第二采样电流Ia2-的限值，在升压电感的储能过程中，第二采样电流Ia2-逐渐增大直至到达基准电压Ia2-LMT，升压电感储能结束，开关管Q1141、Q1142关断，升压电感进入放电阶段。S6, according to the output of the boost inductor current regulator 1543, the synchronous pulse Psy and the output Fp of the pulse oscillator and the comparison result of the second sampling current Ia2- and the reference voltage Ia2-LMT , the output of the PFC PWM pulse width modulator is obtained. The reference voltage Ia2-LMT is the limit value of the second sampling current Ia2- . During the energy storage process of the boost inductor, the second sampling current Ia2- gradually increases until it reaches the reference voltage Ia2-LMT . The energy storage of the boost inductor ends, the switch tubes Q1141 and Q1142 are turned off, and the boost inductor enters the discharge stage.

PFC PWM脉宽调制器正常工作。The PFC PWM pulse width modulator operates normally.

将动态设定值与第二采样电流Ia2-相减后，通过升压电感电流调节器进行PI运算，以控制PFC PWM脉宽调制器的脉冲宽度。After subtracting the dynamic setting value from the second sampling current I a2- , a PI operation is performed through the boost inductor current regulator to control the pulse width of the PFC PWM pulse width modulator.

当乘法器输出的动态设定值到达动态设置值峰值，动态设定值瞬间下降，升压电感电流调节器的输出快速下降，PFC PWM脉宽调制器的输出脉冲快速变为零。When the dynamic setting value output by the multiplier reaches the dynamic setting value peak value, the dynamic setting value drops instantly, the output of the boost inductor current regulator drops rapidly, and the output pulse of the PFC PWM pulse width modulator quickly becomes zero.

S7，根据PFC PWM脉宽调制器的输出控制高频循环功率因数校正器中的两个高频开关管的开关状态。S7, controlling the switching states of two high-frequency switching tubes in the high-frequency cycle power factor corrector according to the output of the PFC PWM pulse width modulator.

通过以下AB相实例进行说明：The following AB phase example is used to illustrate:

如图7所示，第一阶段，AB相线电压为正半波，准谐振电流正半波，升压电感储能：As shown in Figure 7, in the first stage, the AB phase line voltage is a positive half-wave, the quasi-resonant current is a positive half-wave, and the boost inductor stores energy:

AB相逆变桥PWM控制1530的脉冲振荡器1531输出脉冲Fp，逆变PWM控制1533输出PWM脉冲Pf1，AB相采样电压uabi为正，比较器1534输出Up为正，比较器1535输出Un为零，系统工作正常时过压过流保护为开放，起动信号RUN为起动状态，Runb为高电平，Q1121输出高电平，通过逆变桥驱动电路570，驱动开关管1121导通，电容1125充电到uab，电流iab2的回路为：ua→共模电感1112→开关管1121→开关管1124的反并联二极管→变压器1131的初级绕组→电容1126→共模电感1112→ub；同时振荡器1531输出脉冲Fp控制PFC PWM脉宽调制器1546，Q1141、Q1142输出高电平，通过PFC驱动电路580输出驱动脉冲，驱动开关管1141、1142同时导通，电流iab3的回路为：变压器1131的次级绕组电压上正→升压电感1145→开关管1141→开关管1142→谐振电容1132→变压器次级绕组下负；AB相PFC PWM脉宽调制器1540的采样输出电压Uoc和给定值U* oc相减，误差值通过输出电压调节器1541进行PI运算，其输出送到乘法器1542，同时乘法器1542接收AB相线电压采样值uabi和逆变电流采样值Ia1-,三个值相乘输出作为升压电感电流的动态设定值，其值和升压电感采样电流Ia2-相减，误差值通过升压电感电流调节器1543进行PI运算，控制PFC PWM脉宽调制器1546的脉冲宽度。The pulse oscillator 1531 of the AB phase inverter bridge PWM control 1530 outputs a pulse Fp, the inverter PWM control 1533 outputs a PWM pulse Pf1, the AB phase sampling voltage u abi is positive, the comparator 1534 outputs Up is positive, the comparator 1535 outputs Un is zero, the overvoltage and overcurrent protection is open when the system works normally, the start signal RUN is in the start state, Runb is high level, Q1121 outputs a high level, and the inverter bridge drive circuit 570 drives the switch tube 1121 to turn on, the capacitor 1125 is charged to u ab , and the loop of the current i ab2 is: u a → common mode inductor 1112 → switch tube 1121 → anti-parallel diode of switch tube 1124 → primary winding of transformer 1131 → capacitor 1126 → common mode inductor 1112 → u b ; at the same time, the oscillator 1531 outputs a pulse Fp to control PFC PWM pulse width modulator 1546, Q1141, Q1142 output high level, output driving pulse through PFC driving circuit 580, drive switch tube 1141, 1142 to turn on at the same time, the loop of current iab3 is: the voltage of secondary winding of transformer 1131 is positive → boost inductor 1145 → switch tube 1141 → switch tube 1142 → resonant capacitor 1132 → transformer secondary winding is negative; the sampled output voltage Uoc of AB phase PFC PWM pulse width modulator 1540 is subtracted from the given value U * oc , the error value is PI-operated through output voltage regulator 1541, and its output is sent to multiplier 1542, and multiplier 1542 receives AB phase line voltage sampling value u abi and inverter current sampling value I a1 -, and the three values are multiplied and output as the dynamic setting value of boost inductor current, and its value and boost inductor sampling current I a2 -subtracted, the error value is subjected to PI operation by the boost inductor current regulator 1543 to control the pulse width of the PFC PWM pulse width modulator 1546.

如图8所示，第二阶段，AB相线电压为正半波，准谐振电流正半波，升压电感放电：As shown in Figure 8, in the second stage, the AB phase line voltage is a positive half-wave, the quasi-resonant current is a positive half-wave, and the boost inductor is discharged:

当升压电感电流iab3达到乘法器输出升压电感电流的动态设定值峰值，动态设定值瞬间下降，升压电感电流调节器1543输出快速下降，PFC PWM脉宽调制器1546的输出脉冲快速变为零，Q1141、Q1142输出零电平，通过PFC驱动电路580输出低电平，驱动开关管1141、1142同时关断，这时电流iab3的回路为：变压器1131的次级绕组电压上正→升压电感1145→二极管1143→滤波电路150→直流输出负载→开关管1142的反并联二极管→谐振电容1132→变压器次级绕组下负，这个过程升压电感1145和谐振电容1132谐振。When the boost inductor current i ab3 reaches the peak value of the dynamic setting value of the boost inductor current output by the multiplier, the dynamic setting value drops instantly, the output of the boost inductor current regulator 1543 drops rapidly, the output pulse of the PFC PWM pulse width modulator 1546 quickly becomes zero, Q1141 and Q1142 output zero level, and the PFC drive circuit 580 outputs a low level to drive the switch tubes 1141 and 1142 to turn off at the same time. At this time, the loop of the current i ab3 is: the positive voltage of the secondary winding of the transformer 1131 → the boost inductor 1145 → the diode 1143 → the filter circuit 150 → the DC output load → the anti-parallel diode of the switch tube 1142 → the resonant capacitor 1132 → the negative voltage of the secondary winding of the transformer. In this process, the boost inductor 1145 and the resonant capacitor 1132 resonate.

如图9所示，第三阶段，电容1125放电：As shown in FIG9 , in the third stage, the capacitor 1125 is discharged:

当电流iab2下降到最小设定值时，采样电流Ia1-，和设定值Ia1s比较(这个设定值决定当开关管1121关断时电容1125上的电压放电到零)，比较器1532输出低电平，逆变PWMM控制1533输出PWM脉冲Pf1输出低电平，Q1121输出低电平，通过逆变桥驱动电路570，驱动开关管1121关断，这时电容1125放电一直到零。When the current i ab2 drops to the minimum set value, the sampled current I a1 - is compared with the set value I a1s (this set value determines that the voltage on the capacitor 1125 is discharged to zero when the switch tube 1121 is turned off), the comparator 1532 outputs a low level, the inverter PWMM control 1533 outputs a PWM pulse Pf1 outputs a low level, Q1121 outputs a low level, and drives the switch tube 1121 to be turned off through the inverter bridge drive circuit 570. At this time, the capacitor 1125 discharges to zero.

如图10所示，第四阶段，AB相线电压为正半波，准谐振电流负半波，升压电感储能：As shown in Figure 10, in the fourth stage, the AB phase line voltage is a positive half-wave, the quasi-resonant current is a negative half-wave, and the boost inductor stores energy:

当电流iab3下降到零时，采样电流Ia2-为零，比较器1544输出Psy零电平，控制脉冲振荡器1531输出零电平，逆变PWMM控制1533输出PWM脉冲Pf2,Q1122输出高电平，通过逆变桥驱动电路570，驱动开关管1122导通，电容1126充电到uab，电流iab2的回路为：ua→共模电感1112→开关管1122→开关管1123的反并联二极管→变压器1131的初级绕组→电容1125→ub，同时振荡器1531输出脉冲Fp控制PFC PWM脉宽调制器1546，Q1141、Q1142输出高电平，通过PFC驱动电路580输出驱动脉冲1141、11421，开关管1141、1142同时导通，电流iab3的回路为：变压器1131的次级绕组电压下正→谐振电容1132→开关管1142→开关管1141→升压电感1145→变压器次级绕组上负；AB相PFC PWM脉宽调制器1540的采样输出电压Uoc和给定值U* oc相减，误差值通过输出电压调节器1541进行PI运算，其输出送到乘法器1542，同时乘法器1542接收AB相线电压采样值uabi和逆变电流采样值Ia1-，三个值相乘输出作为升压电感电流的动态设定值，其值和升压电感采样电流Ia2-相减，误差值通过升压电感电流调节器1543进行PI运算，控制PFC PWM脉宽调制器1546的脉冲宽度。When the current i ab3 drops to zero, the sampling current I a2 - is zero, the comparator 1544 outputs P sy zero level, the control pulse oscillator 1531 outputs zero level, the inverter PWMM control 1533 outputs PWM pulse Pf2, Q1122 outputs high level, and drives the switch tube 1122 to turn on through the inverter bridge drive circuit 570, and the capacitor 1126 is charged to u ab . The loop of the current i ab2 is: u a → common mode inductor 1112 → switch tube 1122 → anti-parallel diode of switch tube 1123 → primary winding of transformer 1131 → capacitor 1125 → u b . At the same time, the oscillator 1531 outputs pulse Fp to control the PFC PWM pulse width modulator 1546, Q1141 and Q1142 output high level, and outputs drive pulses 1141 and 11421 through the PFC drive circuit 580. The switches 1141 and 1142 are turned on at the same time, and the current i The loop of ab3 is: positive voltage on the secondary winding of transformer 1131 → resonant capacitor 1132 → switch tube 1142 → switch tube 1141 → boost inductor 1145 → negative voltage on the secondary winding of transformer; the sampled output voltage U oc of the AB phase PFC PWM pulse width modulator 1540 is subtracted from the given value U * oc , and the error value is subjected to PI operation through the output voltage regulator 1541, and its output is sent to the multiplier 1542. At the same time, the multiplier 1542 receives the AB phase line voltage sampled value u abi and the inverter current sampled value I a1 -, and the three values are multiplied and output as the dynamic setting value of the boost inductor current, and its value is subtracted from the boost inductor sampled current I a2 -, and the error value is subjected to PI operation through the boost inductor current regulator 1543 to control the pulse width of the PFC PWM pulse width modulator 1546.

如图11所示，第五阶段，AB相线电压为正半波，准谐振电流负半波，升压电感放电：As shown in Figure 11, in the fifth stage, the AB phase line voltage is a positive half-wave, the quasi-resonant current is a negative half-wave, and the boost inductor is discharged:

当升压电感电流iab3达到乘法器输出升压电感电流的动态设定值峰值，动态设定值瞬间下降，升压电感电流调节器1543输出快速下降，PFC PWM脉宽调制器1546的输出脉冲快速变为零，Q1141、Q1142输出零电平，通过PFC驱动电路580输出低电平，驱动开关管1141、1142同时关断，这时电流iab3的回路为：变压器1131的次级绕组电压下正→谐振电容1132→二极管1144→滤波电路150→直流输出负载→开关管1141的反并联二极管→升压电感1145→变压器次级绕组上负，这个过程升压电感1145和谐振电容1132谐振。When the boost inductor current i ab3 reaches the peak value of the dynamic setting value of the boost inductor current output by the multiplier, the dynamic setting value drops instantly, the output of the boost inductor current regulator 1543 drops rapidly, the output pulse of the PFC PWM pulse width modulator 1546 quickly becomes zero, Q1141 and Q1142 output zero level, and the PFC drive circuit 580 outputs a low level to drive the switch tubes 1141 and 1142 to turn off at the same time. At this time, the loop of the current i ab3 is: positive voltage on the secondary winding of the transformer 1131 → resonant capacitor 1132 → diode 1144 → filter circuit 150 → DC output load → anti-parallel diode of the switch tube 1141 → boost inductor 1145 → negative on the secondary winding of the transformer. In this process, the boost inductor 1145 resonates with the resonant capacitor 1132.

如图12所示，第六阶段，电容1126放电：As shown in FIG. 12 , in the sixth stage, the capacitor 1126 is discharged:

当电流iab2下降到最小设定值时，采样电流Ia1-，和设定值Ia1s比较(这个设定值决定当开关管1122关断时电容1126上的电压放电到零)，比较器1532输出低电平，逆变PWM控制1533输出PWM脉冲Pf2输出低电平，Q1122输出低电平，通过逆变桥驱动电路570，驱动开关管1122关断，这时电容1126放电一直到零；当电流iab3下降到零时，采样电流Ia2-为零，比较器1544输出Psy零电平，控制脉冲振荡器1531输出高电平，进入下一个准谐振周期。When the current i ab2 drops to the minimum set value, the sampled current I a1 - is compared with the set value I a1s (this set value determines that the voltage on the capacitor 1126 discharges to zero when the switch tube 1122 is turned off), the comparator 1532 outputs a low level, the inverter PWM control 1533 outputs a PWM pulse Pf2 outputs a low level, Q1122 outputs a low level, and drives the switch tube 1122 to be turned off through the inverter bridge drive circuit 570. At this time, the capacitor 1126 discharges to zero; when the current i ab3 drops to zero, the sampled current I a2 - is zero, the comparator 1544 outputs P sy zero level, and the control pulse oscillator 1531 outputs a high level to enter the next quasi-resonance cycle.

如图13所示，第七阶段，AB相线电压为负半波，准谐振电流正半波，升压电感储能：As shown in Figure 13, in the seventh stage, the AB phase line voltage is a negative half-wave, the quasi-resonant current is a positive half-wave, and the boost inductor stores energy:

当AB相线电压为负半波时，AB相采样电压uabi为负，比较器1534输出Un为正，比较器1535输出Up为零，Q1123输出高电平，通过逆变桥驱动电路570，驱动开关管1123导通，电容1126充电到uba，电流iab2的回路为：ub→共模电感1112→电容1125→变压器1131的初级绕组→开关管1123→开关管1122的反并联二极管→ua,同时振荡器1531输出脉冲Fp控制PFCPWM脉宽调制器1546，Q1141、Q1142输出高电平，通过PFC驱动电路580输出，驱动开关管1141、1142同时导通，电流iab3的回路为：变压器1131的次级绕组电压上正→升压电感1145→开关管1141→开关管1142→谐振电容1132→变压器次级绕组下负；AB相PFC PWM脉宽调制器1540的采样输出电压Uoc和给定值U* oc相减，误差值通过输出电压调节器1541进行PI运算，其输出送到乘法器1542，同时乘法器1542接收AB相线电压采样值uabi和逆变电流采样值Ia1-,三个值相乘输出作为升压电感电流的动态设定值，其值和升压电感采样电流Ia2-相减，误差值通过升压电感电流调节器1543进行PI运算，控制PFC PWM脉宽调制器1546的脉冲宽度。When the AB phase line voltage is a negative half-wave, the AB phase sampling voltage u abi is negative, the comparator 1534 outputs Un as positive, the comparator 1535 outputs Up as zero, Q1123 outputs a high level, and drives the switch tube 1123 to conduct through the inverter bridge drive circuit 570, and the capacitor 1126 is charged to u ba . The loop of the current i ab2 is: u b → common mode inductor 1112 → capacitor 1125 → primary winding of transformer 1131 → switch tube 1123 → anti-parallel diode of switch tube 1122 → u a . At the same time, the oscillator 1531 outputs a pulse Fp to control the PFCPWM pulse width modulator 1546, Q1141 and Q1142 output a high level, and output through the PFC drive circuit 580 to drive the switch tubes 1141 and 1142 to conduct at the same time. The current i The loop of ab3 is: the voltage of the secondary winding of the transformer 1131 is positive on the upper side → the boost inductor 1145 → the switch tube 1141 → the switch tube 1142 → the resonant capacitor 1132 → the negative on the lower side of the secondary winding of the transformer; the sampled output voltage U oc of the AB phase PFC PWM pulse width modulator 1540 is subtracted from the given value U * oc , and the error value is subjected to PI operation through the output voltage regulator 1541, and its output is sent to the multiplier 1542. At the same time, the multiplier 1542 receives the AB phase line voltage sampling value u abi and the inverter current sampling value I a1 -, and the three values are multiplied and output as the dynamic setting value of the boost inductor current, and its value is subtracted from the boost inductor sampling current I a2 -, and the error value is subjected to PI operation through the boost inductor current regulator 1543 to control the pulse width of the PFC PWM pulse width modulator 1546.

如图14所示，第八阶段，AB相线电压为负半波，准谐振电流正半波，升压电感放电：As shown in Figure 14, in the eighth stage, the AB phase line voltage is a negative half-wave, the quasi-resonant current is a positive half-wave, and the boost inductor is discharged:

当升压电感电流iab3达到乘法器输出升压电感电流的动态设定值峰值，动态设定值瞬间下降，升压电感电流调节器1543输出快速下降，PFC PWM脉宽调制器1546的输出脉冲快速变为零，Q1141、Q1142输出零电平，通过PFC驱动电路580输出低电平，驱动开关管1141、1142同时关断，这时电流iab3的回路为：变压器1131的次级绕组电压上正→升压电感1145→二极管1143→滤波电路150→直流输出负载→开关管1142的反并联二极管→谐振电容1132→变压器次级绕组下负，这个过程升压电感1145和谐振电容1132谐振。When the boost inductor current i ab3 reaches the peak value of the dynamic setting value of the boost inductor current output by the multiplier, the dynamic setting value drops instantly, the output of the boost inductor current regulator 1543 drops rapidly, the output pulse of the PFC PWM pulse width modulator 1546 quickly becomes zero, Q1141 and Q1142 output zero level, and the PFC drive circuit 580 outputs a low level to drive the switch tubes 1141 and 1142 to turn off at the same time. At this time, the loop of the current i ab3 is: the positive voltage of the secondary winding of the transformer 1131 → the boost inductor 1145 → the diode 1143 → the filter circuit 150 → the DC output load → the anti-parallel diode of the switch tube 1142 → the resonant capacitor 1132 → the negative voltage of the secondary winding of the transformer. In this process, the boost inductor 1145 and the resonant capacitor 1132 resonate.

如图15所示，第九阶段，电容1126放电：As shown in FIG. 15 , in the ninth stage, the capacitor 1126 is discharged:

当电流iab2下降到最小设定值时，采样电流Ia1-，和设定值Ia1s比较(这个设定值决定当开关管1123关断时电容1126上的电压放电到零)，比较器1532输出低电平，逆变PWMM控制1533输出PWM脉冲Pf1输出低电平，Q1123输出低电平，通过逆变桥驱动电路570，驱动开关管1123关断，这时电容1126放电一直到零。When the current i ab2 drops to the minimum set value, the sampled current I a1 - is compared with the set value I a1s (this set value determines that the voltage on the capacitor 1126 is discharged to zero when the switch tube 1123 is turned off), the comparator 1532 outputs a low level, the inverter PWMM control 1533 outputs a PWM pulse Pf1 outputs a low level, Q1123 outputs a low level, and drives the switch tube 1123 to be turned off through the inverter bridge drive circuit 570. At this time, the capacitor 1126 discharges to zero.

如图16所示，第十阶段，AB相线电压为负半波，准谐振电流负半波，升压电感储能：As shown in Figure 16, in the tenth stage, the AB phase line voltage is a negative half-wave, the quasi-resonant current is a negative half-wave, and the boost inductor stores energy:

当电流iab3下降到零时，采样电流Ia2-为零，比较器1544输出Psy零电平，控制脉冲振荡器1531输出零电平，逆变PWM控制1533输出PWM脉冲Pf2,Q1124输出高电平，通过逆变桥驱动电路570，驱动开关管1124导通，电容1125充电到uba，电流iab2的回路为：ub→共模电感1112→电容1126→变压器1131的初级绕组→开关管1124→开关管1121的反并联二极管→ua,同时振荡器1531输出脉冲Fp控制PFC PWM脉宽调制器1546，Q1141、Q1142输出高电平，通过PFC驱动电路580输出，驱动开关管1141、1142同时导通，电流iab3的回路为：变压器1131的次级绕组电压下正→谐振电容1132→开关管1142→开关管1141→升压电感1145→变压器次级绕组上负；AB相PFC PWM脉宽调制器1540的采样输出电压Uoc和给定值U* oc相减，误差值通过输出电压调节器1541进行PI运算，其输出送到乘法器1542，同时乘法器1542接收AB相线电压采样值uabi和逆变电流采样值Ia1-,三个值相乘输出作为升压电感电流的动态设定值，其值和升压电感采样电流Ia2-相减，误差值通过升压电感电流调节器1543进行PI运算，控制PFC PWM脉宽调制器1546的脉冲宽度。When the current i ab3 drops to zero, the sampling current I a2 - is zero, the comparator 1544 outputs P sy zero level, the control pulse oscillator 1531 outputs zero level, the inverter PWM control 1533 outputs PWM pulse Pf2, Q1124 outputs high level, and drives the switch tube 1124 to turn on through the inverter bridge drive circuit 570, and the capacitor 1125 is charged to u ba . The loop of the current i ab2 is: u b → common mode inductor 1112 → capacitor 1126 → primary winding of transformer 1131 → switch tube 1124 → anti-parallel diode of switch tube 1121 → u a . At the same time, the oscillator 1531 outputs pulse Fp to control the PFC PWM pulse width modulator 1546, Q1141 and Q1142 output high level, and output through the PFC drive circuit 580 to drive the switch tubes 1141 and 1142 to turn on at the same time. The current i The loop of ab3 is: positive voltage on the secondary winding of transformer 1131 → resonant capacitor 1132 → switch tube 1142 → switch tube 1141 → boost inductor 1145 → negative voltage on the secondary winding of transformer; the sampled output voltage U oc of AB phase PFC PWM pulse width modulator 1540 is subtracted from the given value U * oc , and the error value is subjected to PI operation through the output voltage regulator 1541, and its output is sent to the multiplier 1542. At the same time, the multiplier 1542 receives the AB phase line voltage sampled value u abi and the inverter current sampled value I a1 -, and the three values are multiplied and output as the dynamic setting value of the boost inductor current, and its value is subtracted from the boost inductor sampled current I a2 -, and the error value is subjected to PI operation through the boost inductor current regulator 1543 to control the pulse width of the PFC PWM pulse width modulator 1546.

如图17所示，第十一阶段，AB相线电压为负半波，准谐振电流负半波，升压电感放电：As shown in Figure 17, in the eleventh stage, the AB phase line voltage is a negative half-wave, the quasi-resonant current is a negative half-wave, and the boost inductor discharges:

当升压电感电流iab3达到乘法器输出升压电感电流的动态设定值峰值，动态设定值瞬间下降，升压电感电流调节器1543输出快速下降，PFC PWM脉宽调制器1546的输出脉冲快速变为零，Q1141、Q1142输出零电平，通过PFC驱动电路580输出低电平，驱动开关管1141、1142同时关断，这时电流iab3的回路为：变压器1131的次级绕组电压下正→谐振电容1132→二极管1144→滤波电路150→直流输出负载→开关管1141的反并联二极管→升压电感1145→变压器次级绕组上负，这个过程升压电感1145和谐振电容1132谐振。When the boost inductor current i ab3 reaches the peak value of the dynamic setting value of the boost inductor current output by the multiplier, the dynamic setting value drops instantly, the output of the boost inductor current regulator 1543 drops rapidly, the output pulse of the PFC PWM pulse width modulator 1546 quickly becomes zero, Q1141 and Q1142 output zero level, and the PFC drive circuit 580 outputs a low level to drive the switch tubes 1141 and 1142 to be turned off at the same time. At this time, the loop of the current i ab3 is: positive voltage on the secondary winding of the transformer 1131 → resonant capacitor 1132 → diode 1144 → filter circuit 150 → DC output load → anti-parallel diode of the switch tube 1141 → boost inductor 1145 → negative voltage on the secondary winding of the transformer. In this process, the boost inductor 1145 resonates with the resonant capacitor 1132.

如图18所示，第十二阶段，电容1125放电：As shown in FIG. 18 , in the twelfth stage, the capacitor 1125 is discharged:

当电流iab2下降到最小设定值时，采样电流Ia1-，和设定值Ia1s比较(这个设定值决定当开关管1124关断时电容1125上的电压放电到零)，比较器1532输出低电平，逆变PWM控制1533输出PWM脉冲Pf2输出低电平，Q1124输出低电平，通过逆变桥驱动电路570，驱动开关管1124关断，这时电容1125放电一直到零；当电流iab3下降到零时，采样电流Ia2-为零，比较器1544输出Psy零电平，控制脉冲振荡器1531输出高电平，进入下一个准谐振周期。When the current i ab2 drops to the minimum set value, the sampled current I a1 - is compared with the set value I a1s (this set value determines that the voltage on the capacitor 1125 is discharged to zero when the switch tube 1124 is turned off), the comparator 1532 outputs a low level, the inverter PWM control 1533 outputs a PWM pulse Pf2 outputs a low level, Q1124 outputs a low level, and the switch tube 1124 is driven to be turned off through the inverter bridge drive circuit 570. At this time, the capacitor 1125 discharges to zero; when the current i ab3 drops to zero, the sampled current I a2 - is zero, the comparator 1544 outputs P sy zero level, and the control pulse oscillator 1531 outputs a high level to enter the next quasi-resonance cycle.

下面以AB相为例阐述变压器次级循环PFC控制的工作原理：The following takes the AB phase as an example to explain the working principle of the transformer secondary cycle PFC control:

1、AB相升压电感储能：1. AB phase boost inductor energy storage:

AB相逆变桥PWM控制1530的脉冲振荡器1531输出脉冲Fp，控制PFC PWM脉宽调制器1546，Q1141、Q1142输出高电平，通过PFC驱动电路580输出驱动脉冲，驱动开关管1141、1142同时导通，AB相PFC PWM脉宽调制器1540的采样输出电压Uoc和给定值U* oc相减，误差值通过输出电压调节器1541进行PI运算，其输出送到乘法器1542，同时乘法器1542接收AB相线电压采样值uabi和逆变电流采样值Ia1-，三个值相乘输出作为升压电感电流的动态设定值，其值和升压电感采样电流Ia2-相减，误差值通过升压电感电流调节器1543进行PI运算，控制PFC PWM脉宽调制器1546的脉冲宽度。The pulse oscillator 1531 of the AB phase inverter bridge PWM control 1530 outputs a pulse Fp to control the PFC PWM pulse width modulator 1546. Q1141 and Q1142 output a high level and output a drive pulse through the PFC drive circuit 580 to drive the switch tubes 1141 and 1142 to be turned on at the same time. The sampled output voltage U oc of the AB phase PFC PWM pulse width modulator 1540 is subtracted from the given value U * oc . The error value is subjected to PI operation through the output voltage regulator 1541, and its output is sent to the multiplier 1542. At the same time, the multiplier 1542 receives the AB phase line voltage sampling value u abi and the inverter current sampling value I a1- . The three values are multiplied and output as the dynamic setting value of the boost inductor current. The value is subtracted from the boost inductor sampling current I a2- . The error value is subjected to PI operation through the boost inductor current regulator 1543 to control the pulse width of the PFC PWM pulse width modulator 1546.

2、AB相升压电感放电：2. AB phase boost inductor discharge:

当升压电感电流iab3达到乘法器输出升压电感电流的动态设定值峰值，动态设定值瞬间下降，升压电感电流调节器1543输出快速下降，PFC PWM脉宽调制器1546的输出脉冲快速变为零，Q1141、Q1142输出零电平，通过PFC驱动电路580输出低电平，驱动开关管1141、1142同时关断；当电流iab3下降到零时，采样电流Ia2-为零，比较器1544输出Psy零电平，控制脉冲振荡器1531输出零电平。When the boost inductor current i ab3 reaches the peak value of the dynamic setting value of the boost inductor current output by the multiplier, the dynamic setting value drops instantly, the output of the boost inductor current regulator 1543 drops rapidly, the output pulse of the PFC PWM pulse width modulator 1546 quickly becomes zero, Q1141 and Q1142 output zero level, and the PFC drive circuit 580 outputs a low level to drive the switch tubes 1141 and 1142 to turn off at the same time; when the current i ab3 drops to zero, the sampling current I a2- is zero, the comparator 1544 outputs P sy zero level, and the control pulse oscillator 1531 outputs zero level.

如图19所示，图中列出了三相无中间直流环节高功率因数高频准谐振AC/DC变换器AB相的主要工作波形，其中uab为输入交流电压波形，iab1为未经滤波的交流输入具有高频成分的工频电流，iab为经滤波后的交流输入工频电流，G1121、G1122、G1123、G1124是准谐振变换器1120的开关管1121、1122、1123、1124的栅极驱动，ua1b1是准谐振变换器输出高频方波电压，幅值包络线是工频正弦波，iab2是准谐振变换器的谐振电流，幅值包络线是工频正弦波，uc1d1是高频变压器1131的副边电压，G1141、G1142是开关管1141、1142的栅极驱动脉冲，ue1d1是高频循环功率因数校正器1140的输入电压，iab3是升压电感1145的电流，幅值包络线为工频正弦波，Uo是输出直流电压。As shown in FIG19 , the main working waveforms of the AB phase of the three-phase high power factor high frequency quasi-resonant AC/DC converter without an intermediate DC link are listed in the figure, wherein u ab is the input AC voltage waveform, i ab1 is the unfiltered AC input power frequency current with high frequency components, i ab is the filtered AC input power frequency current, G1121, G1122, G1123, G1124 are the gate drives of the switch tubes 1121, 1122, 1123, 1124 of the quasi-resonant converter 1120, u a1b1 is the high frequency square wave voltage output by the quasi-resonant converter, and the amplitude envelope is the power frequency sine wave, i ab2 is the resonant current of the quasi-resonant converter, and the amplitude envelope is the power frequency sine wave, u c1d1 is the secondary voltage of the high frequency transformer 1131, G1141, G1142 are the gate drive pulses of the switch tubes 1141, 1142, u e1d1 is the input voltage of the high frequency cycle power factor corrector 1140, i ab3 is the current of the boost inductor 1145, the amplitude envelope is a power frequency sine wave, and Uo is the output DC voltage.

将三相分成三个单相线电压，第一级将工频交流通过循环逆变器直接将工频交流变换成高频交流，中间没有直流环节，第二级通过高频变压器和高频循环功率因数校正器变换成直流电压，同时进行功率因数校正，功率因数校正中间也没有直流环节，三个独立的直流电压并联后经过π型滤波输出。该电路拓扑结构只有两级开关电路，中间去掉两个直流环节，同时采用高频准谐振方式，可有效减小开关损耗。另外功率因数校正环节放在直流输出前，开关管上的最大耐压就是输出电压。本发明实施例为这种变换器提供了驱动方法，保证其稳定运行。The three phases are divided into three single-phase line voltages. The first stage directly converts the industrial frequency AC into high-frequency AC through a cyclic inverter, without a DC link in the middle. The second stage converts it into a DC voltage through a high-frequency transformer and a high-frequency cyclic power factor corrector, and performs power factor correction at the same time. There is no DC link in the power factor correction. The three independent DC voltages are connected in parallel and output through a π-type filter. This circuit topology has only two switching circuits, two DC links are removed in the middle, and a high-frequency quasi-resonance method is used, which can effectively reduce switching losses. In addition, the power factor correction link is placed before the DC output, and the maximum withstand voltage on the switch tube is the output voltage. The embodiment of the present invention provides a driving method for this converter to ensure its stable operation.

虽然结合附图描述了本发明的实施例，但是本领域技术人员可以在不脱离本发明的精神和范围的情况下作出各种修改和变型，这样的修改和变型均落入由所附权利要求所限定的范围之内。Although the embodiments of the present invention have been described in conjunction with the accompanying drawings, those skilled in the art may make various modifications and variations without departing from the spirit and scope of the present invention, and such modifications and variations are all within the scope defined by the appended claims.

Claims (9)

1. A drive control method of a high-power factor high-frequency quasi-resonance three-phase alternating-current-direct-current converter is characterized by comprising the following steps:
Acquiring a line voltage sampling value between a first phase voltage input end and a second phase voltage input end;
acquiring a first sampling current of a power frequency AC-high frequency AC circulating inverter input by a first phase voltage;
acquiring synchronous pulses according to a second sampling current passing through the boost inductor in the high-frequency circulating power factor corrector;
The line voltage sampling value, the first sampling current and the synchronous pulse are used as indirect control inputs of four AND gates, and four high-frequency switching tubes in the power frequency AC-high-frequency AC circulating inverter are controlled through the four AND gates;
multiplying the difference between the DC output voltage and the DC output given value, the line voltage sampling value and the first sampling current, and then making a difference with the second sampling current to serve as the input of the boost inductor current regulator;
obtaining the output of the PFC PWM pulse width modulator according to the output of the boost inductor current regulator, the output of the synchronous pulse and the output of the pulse oscillator;
Controlling the switching states of two high-frequency switching tubes in the high-frequency circulating power factor corrector according to the output of the PFC PWM pulse width modulator;
The high-power factor high-frequency quasi-resonant three-phase alternating-current/direct-current converter comprises three single-phase line voltage input circuits; each phase line voltage input circuit comprises: obtaining high-frequency alternating current voltage through the power frequency AC-high-frequency AC circulating inverter; the high-frequency alternating voltage is boosted by a high-frequency transformer; and then the high-frequency cyclic power factor corrector is used for converting the high-frequency cyclic power factor corrector into direct current voltage.
2. The driving control method of the high-power factor high-frequency quasi-resonant three-phase ac/dc converter according to claim 1, wherein the reference voltage is a ground voltage.
3. The method for driving and controlling a high-power factor high-frequency quasi-resonant three-phase ac/dc converter according to claim 1, wherein obtaining the synchronization pulse according to the second sampling current through the boost inductor in the high-frequency cyclic power factor corrector comprises:
and comparing the second sampling current with the reference voltage through a first comparator and outputting the comparison result to obtain the synchronous pulse.
4. The driving control method of the high-power factor high-frequency quasi-resonant three-phase AC/dc converter according to claim 1, wherein the line voltage sampling value, the first sampling current, and the synchronization pulse are used as indirect control inputs of four and gates, and four high-frequency switching tubes in the power frequency AC-high-frequency AC cyclic inverter are controlled by the four and gates, comprising:
Taking the line voltage sampling value and the reference voltage as positive end input and negative end input of a second comparator respectively; taking the line voltage sampling value and the reference voltage as negative end input and positive end input of a third comparator respectively;
the output of the second comparator is respectively used as a first input of a first AND gate and a first input of a second AND gate;
the output of the third comparator is respectively used as a first input of a third AND gate and a first input of a fourth AND gate;
Comparing the first sampling current with a set value through a fourth comparator and outputting the comparison result as a first input of an inversion PWM (pulse-Width modulation) module; the pulse generated by the pulse oscillator is used as a second input of the inversion PWM module; the first output of the inversion PWM module is respectively used as the second input of the first AND gate and the second input of the third AND gate; the second output of the inversion PWM module is respectively used as the second input of the second AND gate and the second input of the third AND gate;
and enabling the system overvoltage and overcurrent protection circuit to output a high level in a starting state to serve as a third input of the first AND gate, a third input of the second AND gate, a third input of the third AND gate and a third input of the fourth AND gate.
5. The drive control method of a high-power factor high-frequency quasi-resonant three-phase ac/dc converter of claim 4, further comprising: the synchronization pulse is input to the pulse oscillator.
6. The method for driving and controlling a high-power factor high-frequency quasi-resonant three-phase ac/dc converter according to claim 1, wherein after subtracting said dc output voltage from said dc output set value, PI operation is performed by a voltage regulator, and outputted to a multiplier; and the multiplier receives the line voltage sampling value and the first sampling current, multiplies the three values and outputs the three values as dynamic setting values of the boost inductor current.
7. The method according to claim 6, wherein the PI operation is performed by the boost inductor current regulator after subtracting the second sampling current from the dynamic set value, so as to control the pulse width of the PFC PWM pulse width modulator.
8. The drive control method of a high power factor high frequency quasi-resonant three-phase ac/dc converter of claim 6, further comprising: when the dynamic set value output by the multiplier reaches the peak value of the dynamic set value, the dynamic set value drops instantaneously, the output of the boost inductor current regulator drops rapidly, and the output pulse of the PFC PWM pulse width modulator becomes zero rapidly.
9. The drive control method of a high-power factor high-frequency quasi-resonant three-phase ac/dc converter according to claim 1, further comprising: and the precise rectification circuit consists of two operational amplifiers, six resistors and two diodes and is used for amplifying the signals of the first sampling current or the second sampling current.
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CN113595415A (en) * 2021-06-15 2021-11-02 袁源兰 AC/DC resonant converter
CN116961459A (en) * 2023-08-01 2023-10-27 江苏麦洛电气有限公司 High-frequency resonant circulation conversion inverter
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CN101383555A (en) * 2008-10-07 2009-03-11 江南大学 Single-phase inverter synthesis three-phase high power factor conversion power supply
CN102340249A (en) * 2011-08-03 2012-02-01 江南大学 High power factor soft switching three-phase induction coil synthesis heating power supply
CN209105053U (en) * 2018-11-06 2019-07-12 南京熊猫电子股份有限公司 A kind of isolated form three-phase AC/DC conversion device
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