CN117176258B - Digital modulation method and device - Google Patents

Abstract

The present invention relates to the technical field of optical fiber communication, and in particular, to a digital modulation method and device. The digital modulation method includes: at the transmitting end, before inputting the high-frequency carrier signal to the IQ modulator, using the polarization control algorithm in the microcontroller to roughly adjust the bias voltage of the IQ modulator, and roughly locking the bias point Near the linear point, data is then added for fine adjustment, and the DC component and AC component of the optical power are detected to ensure the stability of the IQ modulator's working state; at the receiving end, a coherent demodulation algorithm is used for adaptive equalization processing, and carrier recovery is used The algorithm performs frequency offset recovery and phase recovery on the output results of the coherent demodulation algorithm. The polarization control algorithm ensures the polarization stability of the optical signal and alleviates the noise growth rate of the optical signal during transmission. The coherent demodulation algorithm and carrier recovery algorithm realize signal demodulation and correct data recovery. Effectively improves the signal transmission distance in light.

Description

A digital modulation method and device

Technical Field

The present invention relates to the field of optical fiber communication technology, and in particular to a digital modulation method and device.

Background Art

Optical communication technology is a communication technology that uses light as a carrier for information transmission to transmit data and information through optical fiber or free space. It has the advantages of high bandwidth, low loss, and anti-interference, and has become one of the most important technologies in the field of modern communications.

In the existing technology, optical communication uses mature erbium-doped fiber amplifiers and wavelength division multiplexing technology to simply and effectively solve the relay transmission and capacity expansion problems of optical communication. However, with the outbreak of mobile Internet, the data traffic of communication networks has grown rapidly, and the pressure faced by backbone networks has increased sharply. However, the potential of erbium-doped fiber amplifiers and wavelength division multiplexing technology has become smaller and smaller, while polarization modulation has strong anti-interference ability for optical signals and can effectively reduce signal distortion and bit error rate.

The polarization state of the optical signal may change during the transmission process, for example, due to factors such as the nonlinear effect and dispersion of the optical fiber, resulting in polarization rotation, cross-coupling and other phenomena. These effects will reduce the polarization retention of the optical signal, thereby limiting the transmission distance. Therefore, in order to ensure the polarization state of the optical signal, it is necessary to improve the accuracy of the bias control in the IQ modulator, thereby achieving a longer transmission distance.

Summary of the invention

Therefore, the technical problem to be solved by the present invention is to overcome the problem that the transmission capacity of optical communication in the prior art needs to be improved urgently.

In order to solve the above technical problems, the present invention provides a digital modulation method, comprising:

At the transmitting end, the digital signals after spread spectrum processing of different users are superimposed, the superimposed digital signals are converted into analog signals, radio frequency modulation is performed, and the analog signals are modulated into high-frequency carrier signals;

Before inputting the high-frequency carrier signal into the IQ modulator, the bias voltage of the IQ modulator is roughly adjusted using the polarization control algorithm in the single-chip microcomputer. The bias voltage value is adjusted within a preset bias voltage range according to the minimum and maximum values of the optical power so as to lock the bias point near the linear point.

After the bias voltage is roughly adjusted, the high-frequency carrier signal is input to the IQ modulator, and the bias voltage is continuously tracked using the polarization control algorithm in the single-chip microcomputer. The bias voltage is first increased and then decreased with a step value to obtain two DC components of optical power of different magnitudes, and two AC components of optical power of different magnitudes. The bias voltages corresponding to the smaller DC component and AC component are respectively taken as the new bias voltages, and the process is continuously iterated until the bias voltage no longer changes.

The high-frequency carrier signal is converted into an optical signal through an IQ modulator, and the power of the optical signal is amplified by a small signal-to-noise amplifier. After completion, the optical signal enters the optical fiber for transmission;

At the receiving end, the optical signal after optical fiber transmission is first amplified by a backward Raman amplifier, and then adaptively equalized using a coherent demodulation algorithm. The output of the coherent demodulation algorithm is then frequency-recovered and phase-recovered using a carrier recovery algorithm to restore it to a high-frequency carrier signal.

The restored high-frequency carrier signal is demodulated into an analog signal, then converted into a digital signal, and despread to separate and process the mathematical signals of each user from the mixed digital signals.

In one embodiment of the present invention, the bias voltage of the IQ modulator is roughly adjusted by using a polarization control algorithm in a single chip microcomputer connected to the IQ modulator, and the bias value is adjusted within a preset bias range according to the minimum and maximum values of the optical power so as to lock the bias point near the linear point. The specific steps include:

The single chip microcomputer quickly scans the bias voltages Vbias_XI and Vbias_XQ of the X polarization arm I and the X polarization arm Q respectively, obtains the bias voltages Vbias_XI and Vbias_XQ of the X polarization arm I and the X polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, and controls the signals of the X polarization arm I and the X polarization arm Q to be near the maximum power point;

After obtaining the bias voltages Vbias_XI and Vbias_XQ of the X polarization arm I and the X polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, the single chip microcomputer quickly scans the bias voltage Vbias_XP of the X polarization component to obtain the bias voltage Vbias_XP of the X polarization component that makes the DC component of the signal light power received by the built-in diode the average value of the maximum value and the minimum value, and controls the X polarization component signal to be located near the 90° phase point. At this time, the bias voltage Vbias_XP of the X polarization component is the initial bias voltage of the X polarization component;

After obtaining the bias voltage Vbias_XP of the X-polarization component that makes the DC component of the signal light power received by the built-in diode the average value of the maximum value and the minimum value, the single-chip microcomputer quickly scans the bias voltages Vbias_XI and Vbias_XQ of the X-polarization arm I and the X-polarization arm Q to obtain the bias voltages Vbias_XI and Vbias_XQ of the X-polarization arm I and the X-polarization arm Q that make the DC component of the signal light power received by the built-in diode minimum, and controls the X-polarization arm I and the X-polarization arm Q signals to be located near the optimal polarization point. At this time, the bias voltages Vbias_XI and Vbias_XQ of the X-polarization arm I and the X-polarization arm Q are the initial X-polarization arm I bias voltage and the initial X-polarization arm Q bias voltage;

The single chip microcomputer quickly scans the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q respectively, obtains the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, and controls the signals of the Y polarization arm I and the Y polarization arm Q to be near the maximum power point;

After obtaining the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, the single-chip microcomputer quickly scans the bias voltage Vbias_YP of the Y polarization component to obtain the bias voltage Vbias_YP of the Y polarization component that makes the DC component of the signal light power received by the built-in diode the average value of the maximum value and the minimum value, and controls the Y polarization component signal to be located near the 90° phase point. At this time, the bias voltage Vbias_YP of the Y polarization component is the initial bias voltage of the Y polarization component;

After obtaining the bias voltage Vbias_YP of the Y polarization component that makes the DC component of the signal light power received by the built-in diode the average of the maximum and minimum values, the single-chip microcomputer quickly scans the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q to obtain the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q that minimize the DC component of the signal light power received by the built-in diode, and controls the Y polarization arm I and the Y polarization arm Q signals to be located near the optimal polarization point. At this time, the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q are the initial Y polarization arm I bias voltage and the initial Y polarization arm Q bias voltage.

In one embodiment of the present invention, the high-frequency carrier signal is input into the IQ modulator, and the polarization control algorithm in the single-chip microcomputer continuously tracks the bias voltage, and obtains two DC components of optical power of different sizes and two AC components of optical power of different sizes by increasing the bias voltage with a step value first and then decreasing, and respectively taking the bias voltage corresponding to the smaller DC component and the smaller AC component as the new bias voltage, and continuously iterating until the bias voltage no longer changes. The specific steps include:

A high-frequency carrier signal is added for modulation to obtain the optical signals output by the X polarization arm and the Y polarization arm;

The single chip microcomputer controls the initial bias voltage of the X polarization component, and makes its step value increase first and then decrease, so as to obtain two AC components of optical power of different magnitudes, and takes the bias voltage corresponding to the smaller power AC component as the bias voltage of the new X polarization component;

After obtaining the bias voltage of the new X polarization component, the single chip microcomputer controls the initial X polarization arm I bias voltage and the initial X polarization arm Q bias voltage, so that the step value thereof is increased first and then decreased, to obtain two optical power DC components of different sizes before and after, and the bias voltage corresponding to the smaller power DC component is taken as the new X polarization arm I bias voltage and the new X polarization arm Q bias voltage;

The single chip microcomputer controls the initial bias voltage of the Y polarization component, and makes its step value increase first and then decrease, so as to obtain two AC components of optical power with different magnitudes, and the bias voltage corresponding to the smaller power AC component is taken as the bias voltage of the new Y polarization component;

After obtaining the bias voltage of the new Y polarization component, the single chip microcomputer controls the initial Y polarization arm I bias voltage and the initial Y polarization arm Q bias voltage, so that the step value thereof is increased first and then decreased, to obtain two DC components of optical power of different sizes before and after, and the bias voltage corresponding to the smaller power DC component is taken as the new Y polarization arm I bias voltage and the new Y polarization arm Q bias voltage;

Repeat the above steps and iterate continuously until the bias voltage of the X polarization arm I, the bias voltage of the X polarization arm Q, the bias voltage of the Y polarization arm I, and the bias voltage of the Y polarization arm Q no longer change, and determine them as the optimal linear point, and lock the bias voltage of the X polarization component of the bias voltage and the bias voltage of the Y polarization component to the 90° phase point to keep the orthogonal phase of the I and Q paths.

In one embodiment of the present invention, the adaptive equalization processing using a coherent demodulation algorithm includes: using an FIR filter to eliminate the signal group delay, and then using a decision maker to make decisions and data decoding on the signal after the group delay is eliminated, and outputting the original transmitted signal without carrier recovery.

In one embodiment of the present invention, the FIR filter is used to eliminate the signal group delay, and the weighted square error cost function of the group delay model is:

,

in, is the passband of the FIR filter, is the stop band of the FIR filter, is the amplitude weight in the cost error function; is the expected amplitude-frequency response of the channel, is the expected group delay frequency response of the channel, is the angular frequency;

is the channel frequency response function:

,

in, is the group delay channel impulse response sequence: , is the channel filter length;

is the group delay frequency response function:

,

in, is the phase-frequency characteristic function of the system, It represents the real part operation;

By discretizing the frequency, the weighted square error cost function becomes:

;

Set the group delay channel impulse response sequence The weighted square error cost function is minimized, and the expression of the group delay channel FIR filter model is obtained as follows: .

In one embodiment of the present invention, the carrier recovery algorithm is used to perform frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm, including: using an Mth power frequency offset estimation algorithm to compensate for the frequency offset of the signal, and using a Viterbi-Viterbi phase estimation algorithm to compensate for the phase of the signal.

In one embodiment of the present invention, the Mth power frequency offset estimation algorithm includes a fourth power frequency offset estimation algorithm, and the specific steps include:

S1. Remove the phase damage introduced by the laser line width:

Set the received signal phase to ,in represents the information phase, represents the phase caused by the laser line width, represents the noise phase;

For the sampled signal The conjugate and initial signal Multiplying them, we get:

,

in It is a slow-changing signal, so the sample values before and after are zero. After this step, the laser line width can be eliminated. The phase damage caused by

S2, remove the residual modulation phase:

Ideally, the residual modulation phase The value of ,but In , can Remove;

Will Divided into real and imaginary parts, the X polarization state is:

,

The real part is:

,

The imaginary part is:

,

in, , They are the initial signals of the I and Q signals of X polarization, , They are sampling signals of I and Q signals in X polarization state respectively;

The X polarization state is divided into two square operations and then the fourth power operation is performed to obtain the X polarization state:

,

;

The calculation process of the Y polarization state is the same as that of the X polarization state;

S3, remove noise phase:

Noise phase In the case of The arithmetic mean of the real and imaginary parts of the X polarization state and the Y polarization state after the fourth power operation is calculated for each signal sample. The phase results obtained are all within the ideal phase value. Then take the average of the phase results to get the ideal phase value ;

S4. Calculate the carrier frequency offset estimate:

The arithmetic mean of the real and imaginary parts of the X polarization state and the Y polarization state obtained in S3 is , , , ;

If the real part of the X polarization state And the imaginary part , then the angle ;

If the real part of the X polarization state And the imaginary part , then the angle ;

If the real part of the X polarization state , then the angle remains unchanged, ;

If the real part of the X polarization state , then when the imaginary part is greater than 0, the angle ; When the imaginary part is less than 0, the angle ;

The calculation of the argument of the Y polarization state is the same as that of the X polarization state;

in , ;

Get frequency deviation value , ;

The frequency deviation values X polarization state and Y polarization state are used to make corrections respectively, and the real and imaginary parts of the corrected X polarization state are obtained as follows:

,

;

The real and imaginary parts of the corrected Y polarization state are:

,

;

After frequency deviation correction, the restored high-frequency carrier signal is output.

In one embodiment of the present invention, a CDMA spreading code is used to perform a spreading process on an initial digital signal to generate a mask code.

In one embodiment of the present invention, at the receiving end, quantization processing and sampling mapping are performed on each channel of the despread digital signal, and the digital signal is restored to the original binary data through multi-level digital modulation decoding.

The present invention also provides a digital modulation device, comprising:

The transmitting terminal comprises:

A spread spectrum module is used to superimpose the digital signals after spread spectrum processing of different users, convert the superimposed digital signals into analog signals, perform radio frequency modulation, and modulate the analog signals into high-frequency carrier signals;

An IQ modulator, whose input end is connected to the output end of the spectrum spreading module, and is used to convert the high-frequency carrier signal into an optical signal;

A single-chip computer connected to the IQ modulator, used to use a polarization control algorithm to roughly adjust the bias voltage of the IQ modulator before the high-frequency carrier signal is input to the IQ modulator, and adjust the bias value within a preset bias range according to the minimum and maximum values of the optical power so as to lock the bias point near the linear point; after the high-frequency carrier signal is input to the IQ modulator, the polarization control algorithm is used to continuously track the bias voltage, and the bias voltage that first increases and then decreases with a step value is used to obtain two DC components of optical power of different sizes, and two AC components of optical power of different sizes, and the bias voltages corresponding to the smaller DC component and the smaller AC component are respectively taken as new bias voltages, and the bias voltage is continuously iterated until the bias voltage no longer changes;

A small signal-to-noise amplifier, whose input end is connected to the IQ modulator and whose output end is connected to the optical fiber link, is used to amplify the power of the optical signal;

The receiving terminal comprises:

A backward Raman amplifier, the input end of which is connected to the optical fiber link, is used to amplify the optical signal after optical fiber transmission;

A carrier recovery module, whose input end is connected to the output end of the backward Raman amplifier, is used to perform adaptive equalization processing using a coherent demodulation algorithm, and use a carrier recovery algorithm to perform frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm to restore it to a high-frequency carrier signal;

The despreading module is used to demodulate the restored high-frequency carrier signal into an analog signal, then convert it into a digital signal, and then despread it.

The above technical solution of the present invention has the following advantages compared with the prior art:

A digital modulation method described in the present invention uses a polarization control algorithm in a single-chip microcomputer to adjust the bias voltage of an IQ modulator at a transmitting end. The polarization control algorithm is divided into two stages: coarse adjustment and fine adjustment. In the coarse adjustment stage, no signal data is added for rapid adjustment, and the bias point is roughly locked near a linear point. Then, data is added for fine adjustment. By detecting the DC component and AC component of the optical power, the stability of the working state of the IQ modulator is ensured, thereby ensuring the polarization stability of the optical signal, and easing the noise growth rate of the optical signal during transmission, thereby enabling ultra-long-distance transmission of the signal in the optical fiber.

The digital modulation method described in the present invention uses a coherent demodulation algorithm at the receiving end to perform adaptive equalization processing on the received signal, effectively reducing the bit error rate in signal transmission, suppressing channel noise and interference by eliminating the group delay distortion generated during the transmission process, and being able to extract information from the original signal to the maximum extent, reducing the error and distortion introduced during the transmission process, thereby improving the reliability of the data. The carrier recovery algorithm is used to perform frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm, and the carrier frequency and phase of the transmitting end can be accurately recovered at the receiving end, achieving frequency and phase synchronization between the receiving end and the transmitting end, thereby achieving signal demodulation and correct data recovery, and improving the transmission distance of the signal in light.

The digital modulation method described in the present invention also utilizes the orthogonality of multi-client communication of CDMA coding to achieve decoding and transmission capabilities for specific users. CDMA is used to construct and generate multi-frequency signals, and the spread spectrum mask in the CDMA coding can ensure that when multiple users communicate and transmit simultaneously, only their own transmission signals can be extracted and decoded, thereby ensuring the security and confidentiality of the overall transmission system for multiple customers' simultaneous communication and transmission.

BRIEF DESCRIPTION OF THE DRAWINGS

In order to make the content of the present invention more clearly understood, the present invention is further described in detail below according to specific embodiments of the present invention in conjunction with the accompanying drawings, wherein

FIG1 is a flow chart of a digital modulation method of the present invention;

FIG2 is a schematic diagram of the structure of a polarization control algorithm according to an embodiment of the present invention;

FIG3 is a block diagram of a coherent demodulation algorithm according to an embodiment of the present invention;

FIG4 is a block diagram of a fourth power frequency offset estimation algorithm according to an embodiment of the present invention;

FIG5 is a structural block diagram of a digital modulation device of the present invention.

DETAILED DESCRIPTION

The present invention will be further described below in conjunction with the accompanying drawings and specific embodiments so that those skilled in the art can better understand the present invention and implement it, but the embodiments are not intended to limit the present invention.

Embodiment 1

Referring to FIG. 1 , the present invention provides a digital modulation method, comprising:

At the transmitting end, binary data is input, multi-level digital modulation is performed, the binary data is digitally modulated and encoded, and sampling mapping and quantization are performed to obtain the initial digital signal; the initial digital signal is spread spectrum processed through the CDMA spread spectrum code to generate a mask; after the initial digital signal is transmitted through CPRI and processed by the digital intermediate frequency, the digital signal is converted into an analog signal using a digital-to-analog converter, and radio frequency modulation is performed to modulate the analog signal into a high-frequency carrier signal; the above processing enables all users to not only use the broadband carrier signal of the same frequency for transmission, but also the signals overlap in time.

The embodiment of the present invention utilizes the orthogonality of multi-client communication of CDMA coding to achieve decoding and transmission capabilities for specific users, and utilizes CDMA to construct and generate multi-frequency signals. The spread spectrum mask in the CDMA coding can ensure that when multiple users communicate and transmit simultaneously, only their own transmission signals can be extracted and decoded, thereby ensuring the security and confidentiality of the overall transmission system for simultaneous communication and transmission of multiple customers.

As shown in FIG2 , before the high-frequency carrier signal is input to the IQ modulator, the bias voltage of the IQ modulator is roughly adjusted using the polarization control algorithm in the single-chip microcomputer. The bias value is adjusted within a preset bias range according to the minimum and maximum values of the optical power so as to lock the bias point near the linear point. The specific steps include:

The single chip microcomputer quickly scans the bias voltages Vbias_XI and Vbias_XQ of the X polarization arm I and the X polarization arm Q respectively, obtains the bias voltages Vbias_XI and Vbias_XQ of the X polarization arm I and the X polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, and controls the signals of the X polarization arm I and the X polarization arm Q to be near the maximum power point;

After obtaining the bias voltages Vbias_XI and Vbias_XQ of the X polarization arm I and the X polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, the single chip microcomputer quickly scans the bias voltage Vbias_XP of the X polarization component to obtain the bias voltage Vbias_XP of the X polarization component that makes the DC component of the signal light power received by the built-in diode the average value of the maximum value and the minimum value, and controls the X polarization component signal to be located near the 90° phase point. At this time, the bias voltage Vbias_XP of the X polarization component is the initial bias voltage of the X polarization component;

After obtaining the bias voltage Vbias_XP of the X-polarization component that makes the DC component of the signal light power received by the built-in diode the average value of the maximum value and the minimum value, the single-chip microcomputer quickly scans the bias voltages Vbias_XI and Vbias_XQ of the X-polarization arm I and the X-polarization arm Q to obtain the bias voltages Vbias_XI and Vbias_XQ of the X-polarization arm I and the X-polarization arm Q that make the DC component of the signal light power received by the built-in diode minimum, and controls the X-polarization arm I and the X-polarization arm Q signals to be located near the optimal polarization point. At this time, the bias voltages Vbias_XI and Vbias_XQ of the X-polarization arm I and the X-polarization arm Q are the initial X-polarization arm I bias voltage and the initial X-polarization arm Q bias voltage;

The single chip microcomputer quickly scans the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q respectively, obtains the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, and controls the signals of the Y polarization arm I and the Y polarization arm Q to be near the maximum power point;

After obtaining the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, the single-chip microcomputer quickly scans the bias voltage Vbias_YP of the Y polarization component to obtain the bias voltage Vbias_YP of the Y polarization component that makes the DC component of the signal light power received by the built-in diode the average value of the maximum value and the minimum value, and controls the Y polarization component signal to be located near the 90° phase point. At this time, the bias voltage Vbias_YP of the Y polarization component is the initial bias voltage of the Y polarization component;

After obtaining the bias voltage Vbias_YP of the Y polarization component that makes the DC component of the signal light power received by the built-in diode the average of the maximum and minimum values, the single-chip microcomputer quickly scans the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q to obtain the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q that minimize the DC component of the signal light power received by the built-in diode, and controls the Y polarization arm I and the Y polarization arm Q signals to be located near the optimal polarization point. At this time, the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q are the initial Y polarization arm I bias voltage and the initial Y polarization arm Q bias voltage.

After the bias voltage is roughly adjusted, the high-frequency carrier signal is input to the IQ modulator, and the bias voltage is continuously tracked using the polarization control algorithm in the single-chip microcomputer. The bias voltage is first increased and then decreased with a step value to obtain two DC components of optical power of different sizes, and two AC components of optical power of different sizes. The bias voltages corresponding to the smaller DC component and AC component are respectively taken as the new bias voltages, and the iteration is continued until the bias voltage no longer changes. The specific steps include:

A high-frequency carrier signal is added for modulation to obtain the optical signals output by the X polarization arm and the Y polarization arm;

The single chip microcomputer controls the initial bias voltage of the X polarization component, and makes its step value increase first and then decrease, so as to obtain two AC components of optical power of different magnitudes, and takes the bias voltage corresponding to the smaller power AC component as the bias voltage of the new X polarization component;

After obtaining the bias voltage of the new X polarization component, the single chip microcomputer controls the initial X polarization arm I bias voltage and the initial X polarization arm Q bias voltage, so that the step value thereof is increased first and then decreased, to obtain two optical power DC components of different sizes before and after, and the bias voltage corresponding to the smaller power DC component is taken as the new X polarization arm I bias voltage and the new X polarization arm Q bias voltage;

The single chip microcomputer controls the initial bias voltage of the Y polarization component, and makes its step value increase first and then decrease, so as to obtain two AC components of optical power with different magnitudes, and the bias voltage corresponding to the smaller power AC component is taken as the bias voltage of the new Y polarization component;

After obtaining the bias voltage of the new Y polarization component, the single chip microcomputer controls the initial Y polarization arm I bias voltage and the initial Y polarization arm Q bias voltage, so that the step value thereof is increased first and then decreased, to obtain two DC components of optical power of different sizes before and after, and the bias voltage corresponding to the smaller power DC component is taken as the new Y polarization arm I bias voltage and the new Y polarization arm Q bias voltage;

Repeat the above steps and iterate continuously until the bias voltage of the X polarization arm I, the bias voltage of the X polarization arm Q, the bias voltage of the Y polarization arm I, and the bias voltage of the Y polarization arm Q no longer change, and determine them as the optimal linear point, and lock the bias voltage of the X polarization component of the bias voltage and the bias voltage of the Y polarization component to the 90° phase point to keep the orthogonal phase of the I and Q paths.

The high-frequency carrier signal is converted into an optical signal through an IQ modulator, and the power of the optical signal is amplified by a small signal-to-noise amplifier. After completion, the optical signal enters a 500km optical fiber link for ultra-long-distance transmission.

When the polarization control algorithm is not added, the signal will cause a surge in bit errors after ultra-long distance transmission due to the increase in signal-to-noise. However, after adding the polarization control algorithm, the signal-to-noise growth rate of the signal is alleviated, thereby enabling ultra-long distance transmission of the signal in the optical fiber.

At the receiving end, the optical signal after optical fiber transmission is first amplified by the backward Raman amplifier, and the coherent demodulation algorithm is used for adaptive equalization processing. The carrier recovery algorithm is used to recover the frequency offset and phase of the output result of the coherent demodulation algorithm and restore it to a high-frequency carrier signal. The following is a detailed introduction to the signal processing process at the receiving end.

Referring to FIG3 , the original signal sent by the signal source is , after a 500km optical fiber link, combined with the channel additive white Gaussian noise , the receiving equalizer receives the sequence The equalizer receives the sequence Get information from the The equalizer weight coefficient , so that the output of the equalizer after convergence The original signal best estimate of .

The model construction process of using filters to eliminate signal group delay is as follows:

When a group signal passes through a communication system, the delay caused by the communication system to the entire group signal wave group is called group delay, where the group signal refers to a complex signal or wave group with a frequency close to multiple frequency components and composed in a certain way, that is, Group delay distortion describes the effect of different frequency parts of a group signal passing through the channel at different times.

The concept and characteristics of signal group delay are as follows:

Assume that the communication channel frequency response characteristic function is:

,

in, and are the channel amplitude-frequency and phase-frequency characteristic functions respectively, then the group delay characteristic function Defined as:

,

The group delay frequency response characteristic function is equal to the system phase frequency characteristic function Diagonal frequency The negative sign indicates that the system output signal always lags behind its input signal. is a constant, that is and It is a linear relationship. At this time, the different frequency parts of the signal have the same group delay. Therefore, the signal will not be distorted when passing through the channel. On the contrary, if If it is not a constant, different frequency parts of the signal will produce group delay distortion when passing through the channel.

Assume that the group delay channel impulse response sequence is ,in is the channel filter length, then the discrete form of the channel frequency response function is:

;

From the above group delay characteristic function The definition of the group delay frequency response function is:

,

in, is the phase-frequency characteristic function of the system, represents the real part operation, Represents the imaginary part.

The expected amplitude-frequency response for a given channel and group delay frequency response , establish the weighted square error cost function of amplitude-frequency and group delay frequency response, and participate in the equalization calculation and coefficient update under the composite mode of the equalizer, then use the FIR filter to eliminate the signal group delay, and the weighted square error cost function of the group delay model is:

,

The embodiment of the present invention uses a FIR filter as an equalizer. is the passband of the FIR filter, is the stop band of the FIR filter, is the amplitude weight in the cost error function; is the expected amplitude-frequency response of the channel, is the expected group delay frequency response of the channel, is the angular frequency.

By discretizing the frequency, the weighted square error cost function becomes:

.

Set the group delay channel impulse response sequence The weighted square error cost function is minimized, and the expression of the group delay channel FIR filter model is obtained as follows:

.

According to the group delay channel FIR filter model, a set of coefficients can be obtained. , that is, the group delay channel impulse response sequence is calculated so that the weighted square error of the group delay model is minimized, thereby obtaining the group delay filter used in the embodiment of the present invention.

Use FIR filter to filter the sequence Multiple equalization processes are performed to eliminate the signal group delay, and then a decision device is used to make decisions and decode data on the signal after the group delay is eliminated, and the original transmitted signal without carrier recovery is output.

At the receiving end, the coherent demodulation algorithm is used to perform adaptive equalization processing on the received signal, which effectively reduces the bit error rate in signal transmission. By eliminating the group delay distortion generated during the transmission process, the channel noise and interference are suppressed, and the information in the original signal can be extracted to the maximum extent, reducing the errors and distortion introduced during the transmission process, thereby improving the reliability of the data.

Since the optical signal will be phase-skewed when it is transmitted over a long distance in the channel, a carrier recovery algorithm needs to be introduced.

Carrier recovery performs frequency offset estimation and phase recovery on the signal after decision and data decoding to obtain the constellation diagram of the corresponding modulation format. It corresponds to the frequency offset recovery and phase recovery in the digital signal processing flow, where the frequency offset recovery algorithm uses the M-th power frequency offset estimation algorithm to correct the residual frequency error, and the phase recovery algorithm uses the Viterbi-Viterbi (V-V) phase estimation algorithm to correct the residual phase error.

4, in the embodiment of the present invention, a quartic frequency offset estimation algorithm is used to correct the residual frequency error, and the specific steps include:

S1. Remove the phase damage introduced by the laser line width:

Set the received signal phase to ,in represents the information phase, represents the phase caused by the laser line width, represents the noise phase, Indicates the sequence number of the sample value, Indicates a cycle;

For the sampled signal The conjugate and initial signal Multiplying them, we get:

,

in It is a slow-changing signal, so the sample values before and after are zero. After this step, the laser line width can be eliminated. The phase damage caused by

S2, remove the residual modulation phase:

Ideally, the residual modulation phase The value of ,but In , can Remove;

Will Divided into real and imaginary parts, the X polarization state is:

,

The real part is:

,

The imaginary part is:

,

in, , They are the initial signals of the I and Q signals of X polarization, , They are sampling signals of I and Q signals in X polarization state respectively;

The X polarization state is divided into two square operations and then the fourth power operation is performed to obtain the X polarization state:

,

;

The calculation process of the Y polarization state is the same as that of the X polarization state:

Will Divided into real and imaginary parts, the Y polarization state is:

,

The real part is:

,

The imaginary part is:

,

in, , They are the initial signals of the I and Q signals of Y polarization respectively. , They are sampling signals of I and Q signals in Y polarization state respectively;

The Y polarization state is divided into two square operations and then subjected to a fourth power operation, and the Y polarization state is obtained as follows:

,

;

S3, remove noise phase:

In an ideal situation, that is, there is no noise phase When signal samples, calculate the arithmetic mean of the real and imaginary parts of the X polarization state and the Y polarization state after the above fourth power operation, and the phase of each result should be , that is, the value of each result is the same; but in reality there is a noise phase In the case of The signal samples are obtained, and the phase result is at the ideal phase value. Therefore, averaging the phase results gives the ideal phase value. ;

S4. Calculate the carrier frequency offset estimate:

The arithmetic mean of the real and imaginary parts of the X polarization state and the Y polarization state obtained in S3 is , , , ;

If the real part of the X polarization state And the imaginary part , then the angle ;

If the real part of the X polarization state And the imaginary part , then the angle ;

If the real part of the X polarization state , then the angle remains unchanged, ;

If the real part of the X polarization state , then when the imaginary part is greater than 0, the angle ; When the imaginary part is less than 0, the angle ;

The calculation of the amplitude angle of the Y polarization state is the same as that of the X polarization state:

If the real part of the Y polarization state And the imaginary part , then the angle ;

If the real part of the Y polarization state And the imaginary part , then the angle ;

If the real part of the Y polarization state , then the angle remains unchanged, ;

If the real part of the Y polarization state , then when the imaginary part is greater than 0, the angle ; When the imaginary part is less than 0, the angle ;

in , ;

Get frequency deviation value , ;

The frequency deviation values X polarization state and Y polarization state are used to make corrections respectively, and the real and imaginary parts of the corrected X polarization state are obtained as follows:

,

;

The real and imaginary parts of the corrected Y polarization state are:

,

;

After frequency deviation correction, the restored high-frequency carrier signal is output.

The carrier recovery algorithm is used to perform frequency offset recovery and phase recovery on the output results of the coherent demodulation algorithm, which can accurately recover the carrier frequency and phase of the transmitter at the receiving end, achieve frequency and phase synchronization between the receiving end and the transmitting end, thereby realizing signal demodulation and correct data recovery, and increasing the transmission distance of the signal in light.

The restored high-frequency carrier signal is subjected to broadband filtering, the in-band signal amplitude is adjusted and limited, and it is demodulated into an analog signal. The analog signal is converted into a digital signal through a digital-to-analog converter, and then after digital intermediate frequency processing and CPRI transmission, the digital signal is despread, the mask is filtered out, and the despread signal is baseband filtered to restore the digital baseband signal; each channel is then quantized and sampled, and the despread digital signal is restored to the original binary data through multi-level digital modulation decoding.

After being processed by the above digital modulation method, the optical signal can be transmitted over a longer distance on the original basis.

Embodiment 2

5, an embodiment of the present invention provides a digital modulation device, including:

The transmitting terminal comprises:

A spread spectrum module is used to superimpose the digital signals after spread spectrum processing of different users, convert the superimposed digital signals into analog signals, perform radio frequency modulation, and modulate the analog signals into high-frequency carrier signals;

An IQ modulator, whose input end is connected to the output end of the spectrum spreading module, and is used to convert the high-frequency carrier signal into an optical signal;

A single-chip computer connected to the IQ modulator, used to use a polarization control algorithm to roughly adjust the bias voltage of the IQ modulator before the high-frequency carrier signal is input to the IQ modulator, and adjust the bias value within a preset bias range according to the minimum and maximum values of the optical power so as to lock the bias point near the linear point; after the high-frequency carrier signal is input to the IQ modulator, the polarization control algorithm is used to continuously track the bias voltage, and the bias voltage that first increases and then decreases with a step value is used to obtain two DC components of optical power of different sizes, and two AC components of optical power of different sizes, and the bias voltages corresponding to the smaller DC component and the smaller AC component are respectively taken as new bias voltages, and the bias voltage is continuously iterated until the bias voltage no longer changes;

A small signal-to-noise amplifier, whose input end is connected to the IQ modulator and whose output end is connected to the optical fiber link, is used to amplify the power of the optical signal;

The receiving terminal comprises:

A backward Raman amplifier, the input end of which is connected to the optical fiber link, is used to amplify the optical signal after optical fiber transmission;

A carrier recovery module, whose input end is connected to the output end of the backward Raman amplifier, is used to perform adaptive equalization processing using a coherent demodulation algorithm, and use a carrier recovery algorithm to perform frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm to restore it to a high-frequency carrier signal;

The despreading module is used to demodulate the restored high-frequency carrier signal into an analog signal, then convert it into a digital signal, and then despread it.

Those skilled in the art will appreciate that the embodiments of the present application may be provided as methods, systems, or computer program products. Therefore, the present application may adopt the form of a complete hardware embodiment, a complete software embodiment, or an embodiment combining software and hardware. Moreover, the present application may adopt the form of a computer program product implemented on one or more computer-usable storage media (including but not limited to disk storage, CD-ROM, optical storage, etc.) containing computer-usable program codes.

The present application is described with reference to the flowcharts and/or block diagrams of the methods, devices (systems), and computer program products according to the embodiments of the present application. It should be understood that each process and/or box in the flowchart and/or block diagram, as well as the combination of the processes and/or boxes in the flowchart and/or block diagram, can be implemented by computer program instructions. These computer program instructions can be provided to a processor of a general-purpose computer, a special-purpose computer, an embedded processor, or other programmable data processing device to generate a machine, so that the instructions executed by the processor of the computer or other programmable data processing device generate a device for implementing the functions specified in one process or multiple processes in the flowchart and/or one box or multiple boxes in the block diagram.

These computer program instructions may also be stored in a computer-readable memory that can direct a computer or other programmable data processing device to operate in a specific manner, so that the instructions stored in the computer-readable memory produce a manufactured product including an instruction device that implements the functions specified in one or more processes in the flowchart and/or one or more boxes in the block diagram.

These computer program instructions may also be loaded onto a computer or other programmable data processing device so that a series of operational steps are executed on the computer or other programmable device to produce a computer-implemented process, whereby the instructions executed on the computer or other programmable device provide steps for implementing the functions specified in one or more processes in the flowchart and/or one or more boxes in the block diagram.

Obviously, the above embodiments are merely examples for clear explanation and are not intended to limit the implementation methods. For those skilled in the art, other different forms of changes or modifications can be made based on the above description. It is not necessary and impossible to list all the implementation methods here. The obvious changes or modifications derived from these are still within the protection scope of the invention.

Claims (9)

1. A digital modulation method, comprising: At the transmitting end, the digital signals after spread spectrum processing of different users are superimposed, the superimposed digital signals are converted into analog signals, radio frequency modulation is performed, and the analog signals are modulated into high-frequency carrier signals; Before inputting the high-frequency carrier signal into the IQ modulator, the bias voltage of the IQ modulator is roughly adjusted using the polarization control algorithm in the single-chip microcomputer. The bias voltage value is adjusted within a preset bias voltage range according to the minimum and maximum values of the optical power so as to lock the bias point near the linear point. After the bias voltage is roughly adjusted, the high-frequency carrier signal is input to the IQ modulator, and the bias voltage is continuously tracked using the polarization control algorithm in the single-chip microcomputer. The bias voltage is first increased and then decreased with a step value to obtain two DC components of optical power of different magnitudes, and two AC components of optical power of different magnitudes. The bias voltages corresponding to the smaller DC component and AC component are respectively taken as the new bias voltages, and the process is continuously iterated until the bias voltage no longer changes. The high-frequency carrier signal is converted into an optical signal through an IQ modulator, and the power of the optical signal is amplified by a small signal-to-noise amplifier. After completion, the optical signal enters the optical fiber for transmission; at the receiving end, the optical signal after optical fiber transmission is first amplified by a backward Raman amplifier, and adaptive equalization processing is performed using a coherent demodulation algorithm, and a carrier recovery algorithm is used to perform frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm; wherein The frequency offset of the signal is compensated using an M-th power frequency offset estimation algorithm, wherein the M-th power frequency offset estimation algorithm includes a quartic frequency offset estimation algorithm, and the specific steps include: S1. Remove the phase damage introduced by the laser line width: The received signal phase is set to θ _k =θ _s (k) + ΔωkT _i + θ _n + θ _ASE , where θ _s (k) represents the information phase, θ _n represents the phase caused by the laser line width, θ _ASE represents the noise phase, and T represents the period; Multiplying the conjugate of the sampled signal S(k-1) and the original signal S(k) yields: Where θ _n is a slow-varying signal, so the preceding and following sample values are zero. After this step, the phase damage caused by the laser line width θ _n can be eliminated; S2, remove the residual modulation phase: Ideally, the residual modulation phase Δθ _s is Then [S ^* (k-1)S(k)] ⁴ It is possible to remove Δθ _s ; Divide S ^* (k-1)S(k) into real and imaginary parts and calculate them separately. The X polarization state is: S _k,x S ^* _k-1,x = (X _I,k +jX _Q,k )·(X _I,k-1 -jX _Q,k-1 ) The real part is: S _{f_x_r} =X _I,k ·X _I,k-1 +X _Q,k ·X _Q,k-1 The imaginary part is: S _{f_x_i} =X _Q,k ·X _I,k-1 -X _I,k ·X _Q,k-1 Wherein, X _I,k and X _Q,k are initial signals of I and Q signals of X polarization, respectively, and X _I,k-1 and X _Q,k-1 are sampling signals of I and Q signals of X polarization, respectively; The X polarization state is divided into two square operations and then the fourth power operation is performed to obtain the X polarization state: S _{f_2_x} =S _{f_2_x_r} +jS _{f_2_x_i} =(S _{f_x_r} +jS _{f_x_i} ) ² =S ² _{f_x_r} -S ² _{f_x_i} +2jS _{f_x_r} S _{f_x_i} S _{f_4_x} =S _{f_4_x_r} +jS _{f_4_x_i} =(S _{f_2_x_r} +jS _{f_2_x_i} ) ² =S ² _{f_2_x_r} -S ² _{f_2_x_i} +2jS _{f_2_x_r} S _{f_2_x_i} The calculation process of the Y polarization state is the same as that of the X polarization state; S3, remove noise phase: In the presence of a noise phase θ _ASE , Nf signal samples are taken, and the arithmetic mean of the real and imaginary parts of the X polarization state and the Y polarization state after the fourth power operation is calculated respectively. The obtained phase results are all located near the ideal phase value ΔωT, and then the obtained phase results are averaged to obtain the ideal phase value ΔωT; S4. Calculate the carrier frequency offset estimate: The arithmetic mean of the real part and the imaginary part of the X polarization state and the Y polarization state obtained in S3 is ∑ _Nf S _{f_4_x_r} , ∑ _Nf S _{f_4_x_i} , ∑ _Nf S _{f_4_y_r} , ∑ _Nf S _{f_4_y_i} ; If the real part ∑ _Nf S _{f_4_x_r} of the X polarization state <0 and the imaginary part ∑ _Nf S _{f_4_x_i} <0, then the argument ΔωT _x.4 ＝ΔωT _x,1,4 -π; If the real part ∑ _Nf S _{f_4_x_r} of the X polarization state <0 and the imaginary part ∑ _Nf S _{f_4_x_i} >0, then the argument ΔωT _x.4 ＝ΔωT _x,1,4 +π; If the real part of the X polarization state ∑ _Nf S _{f_4_x_r} > 0, the argument remains unchanged, ΔωT _x.4 ＝ΔωT _x,1,4 ; If the real part of the X polarization state ∑ _Nf S _{f_4_x_r} = 0, then when the imaginary part is greater than 0, the angle When the imaginary part is less than 0, the angle The calculation of the argument of the Y polarization state is the same as that of the X polarization state; in Get frequency deviation value The X polarization state and the Y polarization state are corrected using the frequency deviation value, and the real and imaginary parts of the corrected X polarization state are: S _{fout_k_x_r} =X _I,k ·cos(Δω _x kT)+X _Q,k ·sin(Δω _x kT) S _{fout_k_x_i} =X _Q,k ·cos(Δω _x kT)-X _I,k ·sin(Δω _x kT) The real and imaginary parts of the corrected Y polarization state are: S _{fout_k_y_r} =Y _I,k ·cos(Δω _y kT)+Y _Q,k ·sin(Δω _y kT) S _{fout_k_y_i} =Y _Q,k ·cos(Δω _y kT)-Y _I,k ·sin(Δω _y kT) After frequency deviation correction, the restored high-frequency carrier signal is output; The restored high-frequency carrier signal is demodulated into an analog signal, then converted into a digital signal, and despread to separate and process the mathematical signals of each user from the mixed digital signals. 2. A digital modulation method according to claim 1, characterized in that the bias voltage of the IQ modulator is roughly adjusted by using a polarization control algorithm in a single-chip microcomputer connected to the IQ modulator, and the bias value is adjusted within a preset bias range according to the minimum and maximum values of the optical power so as to lock the bias point near the linear point, and the specific steps include: The single chip microcomputer quickly scans the bias voltages Vbias_XI and Vbias_XQ of the X polarization arm I and the X polarization arm Q respectively, obtains the bias voltages Vbias_XI and Vbias_XQ of the X polarization arm I and the X polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, and controls the signals of the X polarization arm I and the X polarization arm Q to be near the maximum power point; After obtaining the bias voltages Vbias_XI and Vbias_XQ of the X polarization arm I and the X polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, the single chip microcomputer quickly scans the bias voltage Vbias_XP of the X polarization component to obtain the bias voltage Vbias_XP of the X polarization component that makes the DC component of the signal light power received by the built-in diode the average value of the maximum value and the minimum value, and controls the X polarization component signal to be located near the 90° phase point. At this time, the bias voltage Vbias_XP of the X polarization component is the initial bias voltage of the X polarization component; After obtaining the bias voltage Vbias_XP of the X-polarization component that makes the DC component of the signal light power received by the built-in diode the average value of the maximum value and the minimum value, the single-chip microcomputer quickly scans the bias voltages Vbias_XI and Vbias_XQ of the X-polarization arm I and the X-polarization arm Q to obtain the bias voltages Vbias_XI and Vbias_XQ of the X-polarization arm I and the X-polarization arm Q that make the DC component of the signal light power received by the built-in diode minimum, and controls the X-polarization arm I and the X-polarization arm Q signals to be located near the optimal polarization point. At this time, the bias voltages Vbias_XI and Vbias_XQ of the X-polarization arm I and the X-polarization arm Q are the initial X-polarization arm I bias voltage and the initial X-polarization arm Q bias voltage; The single chip microcomputer quickly scans the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q respectively, obtains the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, and controls the signals of the Y polarization arm I and the Y polarization arm Q to be near the maximum power point; After obtaining the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q that maximize the DC component of the signal light power received by the built-in diode, the single-chip microcomputer quickly scans the bias voltage Vbias_YP of the Y polarization component to obtain the bias voltage Vbias_YP of the Y polarization component that makes the DC component of the signal light power received by the built-in diode the average value of the maximum value and the minimum value, and controls the Y polarization component signal to be located near the 90° phase point. At this time, the bias voltage Vbias_YP of the Y polarization component is the initial bias voltage of the Y polarization component; After obtaining the bias voltage Vbias_YP of the Y polarization component that makes the DC component of the signal light power received by the built-in diode the average of the maximum and minimum values, the single-chip microcomputer quickly scans the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q to obtain the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q that minimize the DC component of the signal light power received by the built-in diode, and controls the Y polarization arm I and the Y polarization arm Q signals to be located near the optimal polarization point. At this time, the bias voltages Vbias_YI and Vbias_YQ of the Y polarization arm I and the Y polarization arm Q are the initial Y polarization arm I bias voltage and the initial Y polarization arm Q bias voltage. 3. A digital modulation method according to claim 2, characterized in that the high-frequency carrier signal is input into the IQ modulator, the polarization control algorithm in the single-chip microcomputer continuously tracks the bias voltage, and obtains two DC components of optical power of different sizes and two AC components of optical power of different sizes by increasing the bias voltage with a step value first and then decreasing, and respectively taking the bias voltage corresponding to the smaller DC component and the smaller AC component as the new bias voltage, and continuously iterating until the bias voltage no longer changes, and the specific steps include: A high-frequency carrier signal is added for modulation to obtain the optical signals output by the X polarization arm and the Y polarization arm; The single chip microcomputer controls the initial bias voltage of the X polarization component, and makes its step value increase first and then decrease, so as to obtain two AC components of optical power of different magnitudes, and takes the bias voltage corresponding to the smaller power AC component as the bias voltage of the new X polarization component; After obtaining the bias voltage of the new X polarization component, the single chip microcomputer controls the initial X polarization arm I bias voltage and the initial X polarization arm Q bias voltage, so that the step value thereof is increased first and then decreased, to obtain two optical power DC components of different sizes before and after, and the bias voltage corresponding to the smaller power DC component is taken as the new X polarization arm I bias voltage and the new X polarization arm Q bias voltage; The single chip microcomputer controls the initial bias voltage of the Y polarization component, and makes its step value increase first and then decrease, so as to obtain two AC components of optical power with different magnitudes, and the bias voltage corresponding to the smaller power AC component is taken as the bias voltage of the new Y polarization component; After obtaining the bias voltage of the new Y polarization component, the single chip microcomputer controls the initial Y polarization arm I bias voltage and the initial Y polarization arm Q bias voltage, so that the step value thereof is increased first and then decreased, to obtain two DC components of optical power of different sizes before and after, and the bias voltage corresponding to the smaller power DC component is taken as the new Y polarization arm I bias voltage and the new Y polarization arm Q bias voltage; Repeat the above steps and iterate continuously until the bias voltage of the X polarization arm I, the bias voltage of the X polarization arm Q, the bias voltage of the Y polarization arm I, and the bias voltage of the Y polarization arm Q no longer change, and determine them as the optimal linear point, and lock the bias voltage of the X polarization component of the bias voltage and the bias voltage of the Y polarization component to the 90° phase point to keep the orthogonal phase of the I and Q paths. 4. A digital modulation method according to claim 1, characterized in that the adaptive equalization processing using a coherent demodulation algorithm includes: using an FIR filter to eliminate the signal group delay, and then using a decision maker to make decisions and data decoding on the signal after the group delay is eliminated, and outputting the original transmitted signal without carrier recovery. 5. A digital modulation method according to claim 4, characterized in that the FIR filter is used to eliminate the signal group delay, and the weighted square error cost function of the group delay model is: e＝α∫ _ω∈P,S ||H(e ^jω )|-|H _d (e ^jω )|| ² dω+∫ _ω∈P |τ(ω)-τ _d (ω)| ² dω Wherein, P is the passband of the FIR filter, S is the stopband of the FIR filter, α is the amplitude weight in the cost error function; H _d (e ^jω ) is the expected amplitude-frequency response of the channel, τ _d (ω) is the expected group delay frequency response of the channel, and ω is the angular frequency; H(e ^jω ) is the channel frequency response function: Wherein, h is the group delay channel impulse response sequence: h = [h(0), h(1), ..., h(N-1)] ^T , N is the channel filter length; τ(ω) is the group delay frequency response function: in, is the phase-frequency characteristic function of the system, Re[·] represents the real part operation; By discretizing the frequency, the weighted square error cost function becomes: The group delay channel impulse response sequence h is set to minimize the weighted square error cost function. The expression of the group delay channel FIR filter model is obtained as follows: 6. A digital modulation method according to claim 1, characterized in that the use of a carrier recovery algorithm to perform frequency offset recovery and phase recovery on an output result of a coherent demodulation algorithm comprises: The Viterbi-Viterbi phase estimation algorithm is used to compensate the phase of the signal. 7. A digital modulation method according to claim 1, characterized in that the initial digital signal is spread spectrum processed using a CDMA spread spectrum code to generate a mask code. 8. A digital modulation method according to claim 1, characterized in that, at the receiving end, each channel of the demodulated digital signal is quantized and sampled and mapped, and the digital signal is restored to the original binary data through multi-level digital modulation decoding. 9. A digital modulation device, comprising: The transmitting terminal comprises: A spread spectrum module is used to superimpose the digital signals after spread spectrum processing of different users, convert the superimposed digital signals into analog signals, perform radio frequency modulation, and modulate the analog signals into high-frequency carrier signals; An IQ modulator, whose input end is connected to the output end of the spectrum spreading module, and is used to convert the high-frequency carrier signal into an optical signal; A single-chip computer is connected to the IQ modulator and is used to use a polarization control algorithm to roughly adjust the bias voltage of the IQ modulator before the high-frequency carrier signal is input to the IQ modulator, and adjust the bias value within a preset bias range according to the minimum and maximum values of the optical power so as to lock the bias point near the linear point; after the high-frequency carrier signal is input to the IQ modulator, the polarization control algorithm is used to continuously track the bias voltage, and the DC components of two optical powers of different sizes are obtained by first increasing and then decreasing the bias voltage with a step value, As for the AC components of two optical powers of different magnitudes, the bias voltages corresponding to the smaller DC component and AC component are respectively taken as new bias voltages, and the process is iterated continuously until the bias voltage no longer changes; A small signal-to-noise amplifier, whose input end is connected to the IQ modulator and whose output end is connected to the optical fiber link, is used to amplify the power of the optical signal; The receiving terminal comprises: A backward Raman amplifier, the input end of which is connected to the optical fiber link, is used to amplify the optical signal after optical fiber transmission; A carrier recovery module, the input end of which is connected to the output end of the backward Raman amplifier, is used to perform adaptive equalization processing using a coherent demodulation algorithm, and use the carrier recovery algorithm to perform frequency offset recovery and phase recovery on the output result of the coherent demodulation algorithm; wherein an M-th power frequency offset estimation algorithm is used to compensate for the frequency offset of the signal, and the M-th power frequency offset estimation algorithm includes a quartic frequency offset estimation algorithm, and the specific steps include: S1. Remove the phase damage introduced by the laser line width: The received signal phase is set to θ _k =θ _s (k) + ΔωkT _i + θ _n + θ _ASE , where θ _s (k) represents the information phase, θ _n represents the phase caused by the laser line width, θ _ASE represents the noise phase, and T represents the period; Multiplying the conjugate of the sampled signal S(k-1) and the original signal S(k) yields: Where θ _n is a slow-varying signal, so the preceding and following sample values are zero. After this step, the phase damage caused by the laser line width θ _n can be eliminated; S2, remove the residual modulation phase: Ideally, the residual modulation phase Δθ _s is Then [S ^* (k-1)S(k)] ⁴ It is possible to remove Δθ _s ; Divide S ^* (k-1)S(k) into real and imaginary parts and calculate them separately. The X polarization state is: S _k,x S ^* _k-1,x = (X _I,k +jX _Q,k )·(X _I,k-1 -jX _Q,k-1 ) The real part is: S _{f_x_r} =X _I,k ·X _I,k-1 +X _Q,k ·X _Q,k-1 The imaginary part is: S _{f_x_i} =X _Q,k ·X _I,k-1 -X _I,k ·X _Q,k-1 Wherein, X _I,k and X _Q,k are initial signals of I and Q signals of X polarization, respectively, and X _I,k-1 and X _Q,k-1 are sampling signals of I and Q signals of X polarization, respectively; The X polarization state is divided into two square operations and then the fourth power operation is performed to obtain the X polarization state: S _{f_2_x} =S _{f_2_x_r} +jS _{f_2_x_i} =(S _{f_x_r} +jS _{f_x_i} ) ² =S ² _{f_x_r} -S ² _{f_x_i} +2jS _{f_x_r} S _{f_x_i} S _{f_4_x} =S _{f_4_x_r} +jS _{f_4_x_i} =(S _{f_2_x_r} +jS _{f_2_x_i} ) ² =S ² _{f_2_x_r} -S ² _{f_2_x_i} +2jS _{f_2_x_r} S _{f_2_x_i} The calculation process of the Y polarization state is the same as that of the X polarization state; S3, remove noise phase: In the presence of a noise phase θ _ASE , Nf signal samples are taken, and the arithmetic mean of the real and imaginary parts of the X polarization state and the Y polarization state after the fourth power operation is calculated respectively. The obtained phase results are all located near the ideal phase value ΔωT, and then the obtained phase results are averaged to obtain the ideal phase value ΔωT; S4. Calculate the carrier frequency offset estimate: The arithmetic mean of the real part and the imaginary part of the X polarization state and the Y polarization state obtained in S3 is ∑ _Nf S _{f_4_x_r} , ∑ _Nf S _{f_4_x_i} , ∑ _Nf S _{f_4_y_r} , ∑ _Nf S _{f_4_y_i} ; If the real part ∑ _Nf S _{f_4_x_r} of the X polarization state <0 and the imaginary part ∑ _Nf S _{f_4_x_i} <0, then the argument ΔωT _x.4 ＝ΔωT _x,1,4 -π; If the real part ∑ _Nf S _{f_4_x_r} of the X polarization state <0 and the imaginary part ∑ _Nf S _{f_4_x_i} >0, then the argument ΔωT _x.4 ＝ΔωT _x,1,4 +π; If the real part of the X polarization state ∑ _Nf S _{f_4_x_r} > 0, the argument remains unchanged, ΔωT _x.4 ＝ΔωT _x,1,4 ; If the real part of the X polarization state ∑ _Nf S _{f_4_x_r} = 0, then when the imaginary part is greater than 0, the angle When the imaginary part is less than 0, the angle The calculation of the argument of the Y polarization state is the same as that of the X polarization state; in Get frequency deviation value The X polarization state and the Y polarization state are corrected using the frequency deviation value, and the real and imaginary parts of the corrected X polarization state are: S _{fout_k_x_r} =X _I,k ·cos(Δω _x kT)+X _Q,k ·sin(Δω _x kT) S _{fout_k_x_i} =X _Q,k ·cos(Δω _x kT)-X _I,k ·sin(Δω _x kT) The real and imaginary parts of the corrected Y polarization state are: S _{fout_k_y_r} =Y _I,k ·cos(Δω _y kT)+Y _Q,k ·sin(Δω _y kT) S _{fout_k_y_i} =Y _Q,k ·cos(Δω _y kT)-Y _I,k ·sin(Δω _y kT) After frequency deviation correction, the restored high-frequency carrier signal is output; The despreading module, whose input end is connected to the output end of the carrier recovery module, is used to demodulate the restored high-frequency carrier signal into an analog signal, then convert it into a digital signal, and perform despreading.