CN114282340B - Method for analyzing and optimizing antenna based on non-port matching RCS - Google Patents
Method for analyzing and optimizing antenna based on non-port matching RCS Download PDFInfo
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Abstract
The invention discloses a method for optimizing an antenna based on non-port matching RCS analysis, and belongs to the technical field of radars. The method takes the matching condition of the incident plane wave and the antenna as a dividing standard, and divides the scattered field into two parts: 1) When any plane wave enters the antenna, the scattering generated by the antenna as a metal plate is used for bearing a scattered field2) When the incident plane wave is matched with the polarization of the antenna and the modes are matched, the corresponding resonant mode of the antenna is excited, and the generated scattering is an antenna scattering fieldAnd provides an expression of the antenna fringe field that is produced when an incident plane wave matches the antenna polarization and the pattern matches, exciting the corresponding resonant mode of the antenna. The method of the invention shows the scattering condition of the antenna more clearly, and interprets the scattering phenomenon of the antenna more accurately; thus, a low scattering antenna with more excellent performance can be designed according to the analysis method.
Description
Technical Field
The invention belongs to the technical field of radars, and particularly relates to a method for optimizing an antenna based on non-port matching RCS analysis.
Background
Stealth technology has been a very important place in modern electronic warfare. With the progressive maturity of radar cross-section reduction technology, antennas have become the most contributing part to radar cross-section on low visible platforms such as aircraft. Therefore, research on reducing the radar cross section of an antenna and ensuring the normal working performance of the antenna is also becoming more important. Before doing this, a clear understanding of the antenna scattering components is required.
Theoretical studies on antenna RCS were first traced back to document "Green R B.The general theory of antenna scattering[J].Southern Journal of Philosophy,1963,6(2):108-114.", where a general theory of antenna scattering based on antenna port conjugate matching conditions was proposed. The scattered field when the antenna port is connected with the conjugate matching is called a structural mode item scattered field, and the other part of the scattered field is the scattered field generated by re-radiation of energy reflected by the antenna port connected with the unconjugated matching load through the antenna.
The basic expression of antenna scattering is derived by using the scattering matrix in literature Shuxi, liu Ying, antenna radar cross section estimation and reduction [ M ], university of western electronic technology press, 2010. Based on the matching condition of the antenna ports as a condition, the scattered field when the antenna ports are connected with a matching load is called a structural mode item scattered field; the fringe field that varies with antenna loading is referred to as the antenna mode term fringe field.
However, applying these theoretical methods to analysis of antenna fringe fields does not explain well the fringe reduction phenomenon that occurs when matching loads are ported, and it was found that the antenna pattern item fringe fields considered in the above studies remain despite the antennas having been ported. That is, the analysis method of dividing the scattered field into the structural mode item scattered field and the antenna mode item scattered field is not practical with the antenna port matching condition as a standard.
Disclosure of Invention
The invention aims to overcome the problem that in the existing analysis method, antenna port matching condition is used as a division standard, so that antenna scattering analysis is not clear enough, and provides a method for optimizing an antenna based on non-port matching RCS analysis.
The technical scheme adopted by the invention is as follows:
A method for optimizing an antenna based on non-port matching RCS analysis, characterized in that an antenna fringe field is divided into two parts according to the following criteria:
1) When any plane wave enters the antenna, the scattering generated by the antenna as a metal plate is used for bearing a scattered field
2) When the incident plane wave is matched with the polarization of the antenna and the modes are matched, the corresponding resonant mode of the antenna is excited, and the generated scattering is an antenna scattering field
Superposing the bearing scattered field and the antenna scattered field to obtain an antenna total scattered field
The expression of the antenna fringe field is as follows:
Where c 1 and c 2 are constant coefficients for the overall antenna structure and satisfy c 1+c2＝1;Γl as the reflection coefficient at the port as seen towards the load; Γ a is the reflection coefficient at the port as seen towards the antenna; lambda is the wavelength of electromagnetic waves in free space; c is a constant coefficient with Guan Rushe plane wave amplitude; amplitude vectors of radiation fields for the antenna remote areas; is a direction vector; for radiating far field polarization vectors; Is the polarization vector of the incident wave; an antenna radiation field in the case of excitation of unit amplitude; in the whole of the process, the process is carried out, For the total received amplitude incident into the antenna interior,
And adjusting the antenna structure according to the analysis result of the total scattered field of the antenna, and optimizing the scattering condition of the antenna.
In the prior art, the matching condition of the antenna ports is used as a division standard, and the problem that part of incident energy is reflected still when the matching load of the antenna terminal is not considered exists. The invention takes the matching condition of the incident plane wave and the antenna as a dividing standard, and provides an antenna scattering field expression generated by exciting a corresponding resonance mode of the antenna when the incident plane wave is matched with the polarization of the antenna and the mode is matched. Equation (2) is derived from the theory of reception of the antenna and taking into account the partial reflected energy that would still exist if the antenna was terminated with a matching load. The analysis method shows the scattering condition of the antenna more clearly, and the scattering phenomenon of the antenna is explained more accurately; thus, a low scattering antenna with more excellent performance can be designed according to the analysis method.
Drawings
FIG. 1 is a top view of an example of the invention;
FIG. 2 is a side view of an example of the invention;
FIG. 3 is a reflectance simulation curve of an example of the invention;
FIG. 4 is a radiation pattern of the present invention illustrated at xoz and yoz planes at 1.143 GHz;
FIG. 5 is a graph of radiation surface current distribution at 1.143GHz for an example of the invention;
FIG. 6 is a graph of single station backscattering and forward scattering changes from 1GHz to 1.25GHz with an exemplary termination 50Ω matching load according to the present invention;
FIG. 7 is a graph of surface scattering current at 1.143GHz with an exemplary termination of the present invention at 50Ω matched load;
FIG. 8 is a plot of single station backscattering and forward scattering changes from 1GHz to 1.25GHz when an exemplary terminated infinite impedance load in accordance with the present invention;
FIG. 9 is a surface scattering current at 1.143GHz when an exemplary termination of the present invention is an infinite impedance load;
FIG. 10 is a surface scattering current at 1.75GHz for an example and a co-aperture PEC plate according to the invention;
FIG. 11 is a fringe field comparison of the present invention with the xoz and yoz faces of the same-aperture PEC plate at 1.75 GHz;
FIG. 12 is a surface scattering current at 2.3GHz for an example and a co-aperture PEC plate according to the invention;
FIG. 13 is a fringe field contrast of an example of the invention and a co-aperture PEC plate at xoz and yoz planes at 2.3 GHz;
FIG. 14 is a surface scattering current at 7GHz for an example and a co-aperture PEC plate according to the invention;
FIG. 15 is a fringe field comparison of an example of the invention and a common-caliber PEC plate at the xoz and yoz faces of 7 GHz;
FIG. 16 is a plot of single station backscattering and forward scattering variations of 3.3GHz-3.6GHz when an exemplary terminated infinite impedance load in accordance with the present invention;
FIG. 17 is a graph of surface scattering current at 3.439GHz when an exemplary termination of the present invention is with an infinite impedance load;
FIG. 18 is a reflectance simulation graph of examples 3.3GHz-3.6GHz according to the present invention;
FIG. 19 is a plot of single-station backscattering and forward scattering changes for 3.3GHz-3.6GHz with an exemplary termination of the present invention at 50Ω matched loading;
fig. 20 is a graph of surface scattering current at 3.439GHz with an exemplary termination of the present invention at 50Ω matching load.
Reference numerals illustrate: 1. the feed point, 2, rectangular metal paster, 3, dielectric plate, 4 ground plane.
Detailed Description
The invention will be further described with reference to the accompanying drawings and the most basic microstrip patch antenna embodiments presented.
The structure diagram of the embodiment is shown in fig. 1, the specification of the upper rectangular metal patch is 70mm×45mm, the specification of the lower grounding layer and the middle dielectric plate is 100mm×75mm, the middle dielectric plate is made of a material with a relative dielectric constant of 3.38, the thickness of the material is 1.524mm, and the feeding point of the 50 ohm coaxial line is 7.3mm away from the center of the antenna.
The thickness of the antenna meets the requirement of the thin microstrip antenna, the resonant frequency of each resonant mode can be accurately calculated by using a cavity model, and specific values corresponding to the antenna are shown in table 1. Fig. 3 is a reflection coefficient simulation curve of the present embodiment, in which it can be seen that the present embodiment has a good match at 1.143GHz, S11 reaches-35.6 dB, and Γ a of the present embodiment can be considered as 0. The real resonance frequency point value of the embodiment is very close to the resonance frequency 1.142GHz of the TM10 mode calculated in table 1. Fig. 4 is a radiation pattern of the present embodiment at xoz and yoz planes at 1.143GHz, with a gain maximum of 6.08dB. Fig. 5 is a radiation surface current distribution at 1.143GHz for this embodiment, which is a very typical TM10 mode surface current distribution. From the above three figures it can be seen that the TM10 mode of this embodiment is well excited when fed here.
TABLE 1
Fig. 6 is a graph of single-station back and front scattering variation under normal illumination of an x-polarized (same as TM10 mode radiation polarization) plane wave when the embodiment is terminated with a 50Ω matching load. According to the analysis method in the invention, the irradiated plane wave satisfies polarization matching, i.e.And pattern matching, i.e.The example corresponding TM10 resonant modes are excited to produce an antenna fringe field. Although the 50 Ω matching load is terminated, Γ l is 0 in the formula (2), the constant coefficient c 2 still exists in the formula (2) of the present invention to make the antenna fringe field not be 0. The loading conditions mainly affect the amplitude of the antenna fringe field that is excited.
Fig. 7 is a graph showing the surface scattering current at 1.143GHz when the current is terminated with a 50Ω matching load, and it can be seen that the scattering current distribution at the patch corresponds to the radiation current distribution in fig. 5, i.e. this part of the current mainly produces the antenna scattering field. And the carrier scattered field is mainly generated by the scattered current on the ground plane. It is also noted that the current flow on the patch is opposite to the current flow on both sides of the ground plane. The two have 180 DEG phase difference, and the generated bearing scattered field and the antenna scattered field are subjected to interference cancellation at a far field. The embodiment has very small back lobes when radiating in the TM10 mode, as can be seen in fig. 4, so that it is mainly observed in fig. 6 that there is a large reduction in back scattering at the resonance frequency point, while the forward scattering changes very little.
Fig. 8 is a graph of single station back and forward scattering variation under normal illumination of an x-polarized plane wave for the same situation when the present embodiment is terminated with an infinite impedance load. Fig. 9 is a graph of the surface scattering current at 1.143GHz when the present embodiment is terminated with an infinite impedance load. The current distribution at the patch is the same as in fig. 5, and the current direction on the patch is also opposite to the current direction on both sides of the ground layer. In formula (2), Γ l =1, the overall excited antenna fringe field amplitude is greater than in the case of terminating a 50Ω matching load. At this point, after the antenna fringe field is in the back-direction and the carrier fringe field is cancelled, there is still a substantial portion remaining, so that no significant reduction in back-scattering is observed at the resonant frequency point. In contrast, since the antenna of this embodiment has smaller back lobes when radiating in TM10 mode, the magnitude of the antenna fringe field in the forward direction is less than the magnitude of the carrier fringe field, and a significant reduction in the forward direction can be observed at this time with an increase in the overall magnitude of the antenna fringe field.
FIG. 10 is the surface scattering current of this example and a same caliber PEC plate at 1.75GHz under vertical irradiation by an x-polarized plane wave. Fig. 11 is a plot of fringe field comparisons for this example and the same-aperture PEC plates at xoz and yoz faces at 1.75 GHz. As can be seen from table 1, 1.75GHz is the resonance frequency of example TM01, but the polarization direction of TM01 mode is y-polarization. The x-polarized plane wave illumination does not satisfy polarization matching, i.e., in formula (2)So that the antenna fringe field cannot be excited at this time, only the carrier fringe field will be present in the total fringe field. It can be clearly observed in fig. 10 and 11 that the example in this case is identical to the current distribution of the PEC plate of the same caliber, concentrated on both sides of the floor and the resulting scattered fields are almost comparable.
FIG. 12 is the surface scattering current of this example and a same caliber PEC plate at 2.3GHz, also vertically illuminated by an x-polarized plane wave. Fig. 13 is a fringe field comparison of this example and a PEC plate of the same caliber at the xoz and yoz faces of 2.3 GHz. As can be seen from Table 1, 2.3GHz is the resonance frequency of the example TM20 mode, but since the vertical direction is the zero point of the TM20 mode pattern, the plane wave incident in this direction does not satisfy the mode matching, i.e., in formula (2)So the antenna fringe field corresponding to the TM20 mode of the example cannot be excited, and only the carrier fringe field is in the total fringe field. It can be clearly observed in fig. 12 and 13 that in this case the current distribution is the same as for the PEC plate of the same caliber and the generated scattering fields are almost equivalent. Similarly, fig. 14 and 15 are surface scattering currents at 7GHz for the example and same caliber PEC plates, respectively, under normal irradiation by an x-polarized plane wave, versus the scattering fields for the xoz and yoz faces. 7GHz is the resonance point of the mode of example TM04, and plane wave incidence in this case does not satisfy either polarization or mode matching, so the total fringe field also only carries fringe field, and current distribution and fringe field similar to those of the same caliber PEC plate are also observed in FIGS. 14 and 15.
Fig. 16 is a graph of single station back and forward scatter variation under vertical illumination of an x-polarized plane wave at 3.3GHz-3.6GHz when the present embodiment is terminated with an infinite impedance load. Fig. 17 is a graph of the surface scattering current at 3.439GHz when the present embodiment is terminated with an infinite impedance load. As can be seen from table 1, 3.439GHz is the resonance frequency point of the example TM30 mode. The exemplary TM30 mode is excited to produce an antenna fringe field because the x-polarized plane wave satisfies the mode matching and polarization matching when illuminated vertically. As shown in FIG. 17, there is a typical surface current distribution in TM30 mode on the patch and the direction is opposite to the current direction at both sides of the floor, so a near-35 dB reduction can be observed in FIG. 16 by backscattering at 3.439 GHz. It can be considered that in the backward direction the amplitude of the excited antenna fringe field is almost equal to the carrier fringe field amplitude.
FIG. 18 is a reflection coefficient simulation curve of the present embodiment from 3.3GHz to 3.6 GHz. Fig. 19 is a graph of single station backscattering and forward scattering changes for 3.3GHz-3.6GHz when the present embodiment is terminated with 50Ω matching loads. Fig. 20 is a graph of the surface scattering current at 3.439GHz when the present embodiment is terminated with a 50Ω matching load. Although there is no perfect match, the coefficients in the corresponding equation (2) decrease. The antenna mode field amplitude excited when the embodiment terminates 50Ω is reduced compared to the case where the impedance is terminated infinitely, so the degree of backscatter reduction observed in fig. 19 is correspondingly reduced.
The scattering conditions of the antenna obtained by applying the analysis method of the invention are described above, so that the scattering mechanism of the antenna is more clearly shown, and meanwhile, various scattering phenomena in the example are more clearly and completely explained. According to the analysis result, the reduced design of the antenna scattering can be well realized. The reduction for the in-band scattered field can be divided into two cases: co-polarized plane wave incidence and cross-polarized plane wave incidence. For incidence of the co-polarized plane wave, reasonably adjusting the antenna structure to control the amplitude of the antenna scattering field excited by the plane wave, so that the amplitude of the antenna scattering field is similar to that of the bearing scattering field, and interference cancellation is achieved to the greatest extent; for cross polarization plane wave incidence, only bearing scattered fields exist at the moment, and induced currents at the edges of the floor and the patch can be cut off, so that the effect of reducing the total scattered fields is achieved. The idea of reducing the out-of-band scattered field of the antenna is the same as that of the in-band scattered field in the incidence condition of cross polarization plane waves.
Claims (1)
1. A method for optimizing an antenna based on non-port matching RCS analysis, characterized in that an antenna fringe field is divided into two parts according to the following criteria:
1) When any plane wave enters the antenna, the scattering generated by the antenna as a metal plate is used for bearing a scattered field
2) When the incident plane wave is matched with the polarization of the antenna and the modes are matched, the corresponding resonant mode of the antenna is excited, and the generated scattering is an antenna scattering field
Superposing the bearing scattered field and the antenna scattered field to obtain an antenna total scattered field
The expression of the antenna fringe field is as follows:
Where c 1 and c 2 are constant coefficients for the overall antenna structure and satisfy c 1+c2＝1;Γl as the reflection coefficient at the port as seen towards the load; Γ a is the reflection coefficient at the port as seen towards the antenna; lambda is the wavelength of electromagnetic waves in free space; c is a constant coefficient with Guan Rushe plane wave amplitude; amplitude vectors of radiation fields for the antenna remote areas; is a direction vector; for radiating far field polarization vectors; Is the polarization vector of the incident wave; An antenna radiation field in the case of excitation of unit amplitude;
And adjusting the antenna structure according to the analysis result of the total scattered field of the antenna, and optimizing the scattering condition of the antenna.
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| CN112151969A (en) * | 2020-09-25 | 2020-12-29 | 电子科技大学 | In-band RCS Control Method for Strongly Coupled Broadband Phased Array Based on Generalized Scattering Matrix |
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| CN112751184B (en) * | 2020-12-29 | 2021-11-02 | 电子科技大学 | A Phased Array Antenna with High Radiation Efficiency and Low Scattering Properties |
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| CN111984911A (en) * | 2020-08-17 | 2020-11-24 | 西安电子科技大学 | A Comprehensive Optimization Method of Radiation and Scattering of Array Antenna Based on Electromagnetic Cancellation |
| CN112151969A (en) * | 2020-09-25 | 2020-12-29 | 电子科技大学 | In-band RCS Control Method for Strongly Coupled Broadband Phased Array Based on Generalized Scattering Matrix |
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