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CN113141121A - Current source type high-frequency isolation matrix type cascade converter and control method - Google Patents

Current source type high-frequency isolation matrix type cascade converter and control method Download PDF

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CN113141121A
CN113141121A CN202110437779.3A CN202110437779A CN113141121A CN 113141121 A CN113141121 A CN 113141121A CN 202110437779 A CN202110437779 A CN 202110437779A CN 113141121 A CN113141121 A CN 113141121A
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current source
matrix converter
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voltage
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CN113141121B (en
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王政
吴佳丽
徐阳
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Southeast University
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/02—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC
    • H02M5/04—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters
    • H02M5/22—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M5/275—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M5/293—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M5/00—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases
    • H02M5/02—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC
    • H02M5/04—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters
    • H02M5/22—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M5/225—Conversion of AC power input into AC power output, e.g. for change of voltage, for change of frequency, for change of number of phases without intermediate conversion into DC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode comprising two stages of AC-AC conversion, e.g. having a high frequency intermediate link
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/02—Conversion of AC power input into DC power output without possibility of reversal
    • H02M7/04—Conversion of AC power input into DC power output without possibility of reversal by static converters
    • H02M7/06—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes without control electrode or semiconductor devices without control electrode
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02P—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN THE PRODUCTION OR PROCESSING OF GOODS
    • Y02P80/00—Climate change mitigation technologies for sector-wide applications
    • Y02P80/20—Climate change mitigation technologies for sector-wide applications using renewable energy

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Abstract

本发明公开了一种电流源型高频隔离矩阵型级联变换器及控制方法，包括：三相交流电网经过LC滤波后，分别与两个电流源型矩阵变换器连接传输电能；高频隔离变压器原边侧输入端与矩阵变换器串联，副边输出端连接不控整流桥，两套电流源型高频隔离矩阵型级联变换器交流侧并联且公用LC滤波器，直流输出侧串联接阻感负载。在该系统中，工频交流经三相/单相直接式矩阵变换器变换至高频交流形式，去除直流母线电容，降低系统故障率，减少功率变换级数，大大地减小了系统体积。本发明的控制技术方案使得矩阵变换器器件开关柔性化，抑制了开关器件高频工作带来的电磁干扰问题，满足海上风力发电、电动汽车充电等工业需求，应用前景广阔。

Figure 202110437779

The invention discloses a current source type high-frequency isolation matrix type cascade converter and a control method, comprising: after a three-phase alternating current grid is filtered by LC, it is respectively connected with two current source type matrix converters to transmit electric energy; The input end of the primary side of the transformer is connected in series with the matrix converter, and the output end of the secondary side is connected to an uncontrolled rectifier bridge. Resistive load. In this system, the power frequency AC is converted to the high frequency AC form through the three-phase/single-phase direct matrix converter, the DC bus capacitance is removed, the system failure rate is reduced, the number of power conversion stages is reduced, and the system volume is greatly reduced. The control technical scheme of the invention makes the switching of the matrix converter device flexible, suppresses the electromagnetic interference problem caused by the high-frequency operation of the switching device, meets the industrial demands of offshore wind power generation, electric vehicle charging and the like, and has broad application prospects.

Figure 202110437779

Description

Current source type high-frequency isolation matrix type cascade converter and control method
Technical Field
The invention relates to the field of wind power generation, in particular to a current source type high-frequency isolation matrix type cascade converter and a control method.
Background
Wind energy (onshore and offshore) is one of the most important energy sources in renewable energy power generation systems, and at present, with the continuous development of offshore wind power generation, the power of an electric energy transmission system is continuously increased, and the size of a corresponding converter is also continuously enlarged. A power frequency transformer in a wind energy system of a traditional permanent magnet synchronous generator is large in size, and the traditional generator in the wind energy system mainly comprises a fan impeller, a three-level gear box, a synchronous generator, a bidirectional PWM rectifier and the like. However, the conventional transformer in such a system has a large weight and volume, a small power capacity, and poor power transmission controllability, and because the transformer has a solid iron core and copper windings, the maintenance cost is high, and the manufacturing cost is expensive, so that the modular production and assembly of the wind power system are not facilitated. The transmission power of the wind power system is continuously increased, and the requirements on the power electronic converter system are higher and higher. When a certain phase in the power circuit breaks down, the transmission of other bridge arms is influenced, and the normal operation of the whole system is further influenced. For the above reasons, research on power electronic converters has been a hot spot in offshore wind power generation high voltage direct current transmission systems.
The cascade structure is a promising offshore wind farm configuration and can replace expensive and huge offshore substations. The cascade structure based on the modular direct matrix converter is very suitable for a high-voltage direct-current transmission system with a current source converter due to the characteristics of high power output and high dynamic response. A high-frequency isolation transformer is adopted to replace a low-frequency input transformer, and an electrolytic capacitor which is large in size and easy to damage in a wind energy conversion system is eliminated. Therefore, the power density, reliability and transmission efficiency of the system can be improved. In addition, the direct matrix converter can also realize single-stage power conversion and soft switching, and further reduce the loss of the system. The modular design of the direct matrix converter facilitates the installation and maintenance of the system, which is of great importance to offshore wind farms, and the cost reduction is beneficial to the large-scale application of offshore wind farms. Finally, the series configuration eliminates an offshore substation and facilitates the pooling of multiple low voltage wind power generation systems into a medium voltage grid. And a proper power device can be selected, and the advantages of high reliability, controllable power factor, high waveform quality and the like are realized.
A great part of loss of the high-frequency isolation matrix type cascade converter comes from switching loss of a switching device, and the application of the current conversion soft switching technology of the bidirectional switch of the matrix converter can greatly reduce the switching loss of the device and improve the transmission efficiency of the converter. The traditional matrix converter soft switching technology is mostly specific to a voltage source type converter, the current source type soft switching technology is not researched much, and the defect of large loss of the current source type matrix converter can be effectively overcome by adopting the soft switching technology. An active damping harmonic suppression scheme is adopted in a control strategy, so that low-order harmonics in a cascade system are effectively suppressed, and harmonic loss of the system is reduced.
Disclosure of Invention
In order to solve the defects mentioned in the background technology, the invention aims to provide a current source type high-frequency isolation matrix type cascade converter and a control method, the input sides of two current source type high-frequency isolation matrix converters are connected in parallel and the output sides of the two current source type high-frequency isolation matrix converters are connected in series, the power conversion capacity of a system is improved, the reliability of the system is enhanced, and the volume of the system is reduced; by applying the soft switching technology, the switching loss of the current source type high-frequency isolation matrix converter is reduced, and the efficiency of the system is improved.
The purpose of the invention can be realized by the following technical scheme:
a current source type high-frequency isolation matrix type cascade converter comprises a three-phase power grid, an LC filter, a first current source type matrix converter, a second current source type matrix converter, a first leakage inductor, a second leakage inductor, a first high-frequency isolation transformer, a second high-frequency isolation transformer, a first uncontrolled rectifier bridge, a second uncontrolled rectifier bridge, a first inductance resistance load and a second inductance resistance load, wherein the three-phase power grid is connected with the LC filter; the LC filter is respectively connected with the first current source type matrix converter and the second current source type matrix converter in parallel; the alternating current output side of the first current source type matrix converter is connected with the first high-frequency isolation transformer in series, and the alternating current output side of the second current source type matrix converter is connected with the second high-frequency isolation transformer in series; the secondary output side of the first high-frequency isolation transformer is connected with the first uncontrolled rectifier bridge in series, and the secondary output side of the second high-frequency isolation transformer is connected with the second uncontrolled rectifier bridge in series, so that energy is transferred through a magnetic field; the first uncontrolled rectifier bridge output end and the second uncontrolled rectifier bridge output end are connected in series through a first resistance-inductance load and a second resistance-inductance load to form a closed loop.
Further, the power direction and the power magnitude of the current source type high-frequency isolation matrix type cascade converter are determined by the control modules of the first current source type matrix converter and the second current source type matrix converter; the first current source type matrix converter and the second current source type matrix converter are respectively connected with ports of the LC filter circuit and receive electric energy transmitted by the three-phase power grid; the current of the first resistance-inductance load and the current of the direct current bus inductor corresponding to the second resistance-inductance load output by the direct current side are controlled by a current loop.
Further, the control method adopted by the control module of the first current source type matrix converter comprises the following processes:
1) capacitor voltage U passing through three-phase LC filterabcAnd the network voltage VgElectric angle theta obtained by phase-locked loopeObtaining a capacitance voltage d-axis component U of the LC filter through coordinate transformationdAnd q-axis component Uq；
2) Component U under filter capacitor voltage dq coordinate systemdAnd UqObtaining the low-frequency component of the capacitor voltage through a low-pass filter, the electrical angle thetaeObtaining the electrical angular velocity omega after differentiationeAnd calculating to obtain the low-frequency capacitance current of the filter capacitor
Figure BDA0003033822870000021
And
Figure BDA0003033822870000022
3) given load side DC bus current Idc *And the actual DC bus current IdcThe error value between the two is obtained by a D-axis direct current component through a PI controller
Figure BDA0003033822870000023
To realize the transmission of unit power factor, let the reference value Q of reactive powerrefIs 0, QrefAnd the network voltage VgObtaining the q-axis current component after calculation
Figure BDA0003033822870000024
Is zero;
4) the dq axis component of the capacitor voltage is processed by a high-pass filter to obtain a high-frequency component U of the capacitor voltagehdAnd UhqThe high-frequency components of the capacitor voltage are respectively multiplied by a same virtual resistance coefficient kpvAnd kpvObtaining dq axis component values of the virtual current, and eliminating fifth and seventh harmonics of the system current through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Figure BDA0003033822870000031
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
Figure BDA0003033822870000032
And
Figure BDA0003033822870000033
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after polar coordinate conversion1dc *And phase angle thetaα1；
6) Phase current fundamental wave peak value I1dc *Divided by a given value of DC current Idc *Obtaining a modulation ratio m of the first current source type matrix converter1iPhase angle thetaα1Plus the electrical angle theta measured by the phase-locked loopeObtaining the switching pulse phase angle theta of the matrix converter1ωUsing the modulation ratio m1iAnd angle theta1ωAnd the switching period Ts generates twelve switching pulses of the first current source matrix converter.
Further, the control method adopted by the control module of the second current source type matrix converter comprises the following processes:
1) capacitor voltage U passing through three-phase LC filterabcAnd the network voltage VgElectric angle theta obtained by phase-locked loopeObtaining LC filter by coordinate transformationCapacitor voltage d-axis component U of wave filterdAnd q-axis component Uq；
2) Component U under filter capacitor voltage dq coordinate systemdAnd UqObtaining the low-frequency component of the capacitor voltage through a low-pass filter, the electrical angle thetaeObtaining the electrical angular velocity omega after differentiationeThe low-frequency capacitance current of the filter capacitor can be obtained after calculation
Figure BDA0003033822870000034
And
Figure BDA0003033822870000035
3) given load side DC bus current Idc *And the actual DC bus current IdcThe error value between the two is obtained by a D-axis direct current component through a PI controller
Figure BDA0003033822870000036
To realize the transmission of unit power factor, let the reference value Q of reactive powerrefIs 0, QrefAnd the network voltage VgObtaining the q-axis current component after calculation
Figure BDA0003033822870000037
Is zero;
4) the dq axis component of the capacitor voltage is processed by a high-pass filter to obtain a high-frequency component U of the capacitor voltagehdAnd UhqThe high-frequency components of the capacitor voltage are respectively multiplied by a same virtual resistance coefficient kpvAnd kpvObtaining dq axis component values of the virtual current, and eliminating fifth and seventh harmonics of the system current through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Figure BDA0003033822870000038
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
Figure BDA0003033822870000039
And
Figure BDA00030338228700000310
obtaining a given value of final current at the input side of the matrix converter through operation, and obtaining a phase current fundamental wave peak value I after polar coordinate conversion2dc *And phase angle thetaα2；
6) Phase current fundamental wave peak value I2dc *Divided by a given value of DC current Idc *Obtaining a modulation ratio m of the second current source type matrix converter2iPhase angle thetaα2Plus the electrical angle theta measured by the phase-locked loopeObtaining the switching pulse phase angle theta of the matrix converter2ωUsing the modulation ratio m2iAnd angle theta2ωAnd the switching period Ts generates twelve switching pulses of the second current source matrix converter.
Further, the modulation method adopted by the first current source type matrix converter and the second current source type matrix converter comprises the following steps:
three current vectors acting on the first current source type matrix converter in one switching period are respectively I11，I12，I10The output voltage of the corresponding first current source type matrix converter is U11，U12，U10By changing the order of action of the current vectors, U is made11>U12>U10(ii) a Similarly, the three current vectors acting on the second current source type matrix converter in one switching period are respectively I21，I22，I20Corresponding to the input voltage of the second current source type matrix converter as U21，U22，U20By changing the order of action of the current vectors, U is made21>U22>U20(ii) a Because the first current source matrix converter and the second current source matrix converter are connected in parallel, the output voltage waveforms of the matrix converters in a switching period are consistent, and the working states are consistent, taking the working state of the first current source matrix converter as an example, the specific working process of the soft switch in a switching period is as follows,
1) state 0: primary side commutation
When a switching period begins, the current vectors corresponding to the first current source type matrix converter and the second current source type matrix converter are zero vectors I7Output voltage v of matrix converterpIs reduced to 0;
2) state 1: conduction time of switch tube
The zero current vector still acts, the primary side input side current and the secondary side output side current of the first high-frequency isolation transformer are equal, and the primary side current of the first high-frequency isolation transformer flows through a switching tube S of the first current source type matrix converter11、S22And S14、S24The current of the secondary side rectifier bridge of the first high-frequency isolation transformer flows through D1、D4Three-phase capacitors provide current paths to three-phase inductors, this state being free of energy transmission, S11And S14Conducting at zero voltage;
3) state 2: conduction time of switch tube
Zero vector I of first current source type matrix converter7End of action, current vector I1In operation, the inductive current of the first uncontrolled rectifier bridge flows through the diode D1、D4According to the four-step commutation, the switching tube S of the first current source type matrix converter11And S14Is turned off because the voltage u is nowabGreater than 0, current to S16And S26The output capacitor of the first current source type matrix converter16And S26Zero voltage conduction is carried out, and the voltage of the input side of the first high-frequency isolation transformer is equal to uabEnergy flows from the grid;
4) state 3: conduction time of switch tube
Similar to the operating state of state 2, the effective current vector I of the first current source matrix converter1End of action, current vector I2In operation, the inductive current of the first uncontrolled rectifier bridge flows through the diode D1、D4According to the four-step commutation, the power switch tube S of the first current source type matrix converter16And S26Closing, S12And S22Zero voltage conduction is carried out, and the voltage of the input side of the first high-frequency isolation transformer is equal to uacEnergy flows from the grid;
5) and 4: commutation of uncontrolled rectifier bridge
The rectification of the first uncontrolled rectifier bridge is completed with the aid of the first current source matrix converter, at t3At the moment, the diode D of the first uncontrolled rectifier bridge2、D3Zero current conduction. The current flowing through the inductor in the first uncontrolled rectifier bridge decreases linearly through D2、D3Through D1、D4Is linearly decreased at the same rate, the commutation overlap time TdSelecting 100ns, and commutating current on the first uncontrolled rectifier bridge inductor before the mode is finished;
6) and state 5: conduction time of switch tube
Commutation overlap time TdAfter that, the power switch tube S of the first current source type matrix converter12And S22Off, S14And S24Zero voltage conduction, the voltage drop of the primary side input side of the first high-frequency isolation transformer is 0, and no direct current energy is transmitted under the mode;
7) and 6: conduction time of switch tube
First current source type matrix converter current vector I2End of action, zero vector I0In operation, a direct current flows through the diode D of the first uncontrolled rectifier bridge on the load side2And D3While the input side current of the first high-frequency isolation transformer flows through the power switch tube S of the first current source type matrix converter11、S21And S24、S14And the three-phase capacitor in the LC filter provides a current channel for the three-phase inductor.
The invention has the beneficial effects that:
(1) the direct conversion from power frequency alternating current to high frequency alternating current is realized through the current source type matrix converter, the direct current capacitor is removed, the failure rate of the converter is reduced, the power conversion stage number is reduced, and the reliability and the power density of system operation are greatly improved;
(2) a high-frequency isolation transformer is adopted to replace a low-frequency input transformer, so that the system volume is reduced;
(3) the capacity of the power converter is increased through the cascade structure of the plurality of direct matrix converters, the output voltage of the system is improved, and the collection of medium-high voltage direct current transmission is facilitated;
(4) the control strategy of the matrix converter can realize unit power factor operation, the soft switching technology reduces the loss of the system, the modularized matrix converter is easy to install and maintain, and the cost of the offshore wind power generation system is reduced.
Drawings
The invention will be further described with reference to the accompanying drawings.
FIG. 1 is an overall architecture diagram of the present invention;
FIG. 2 is a schematic block diagram of a control method for two current source matrix converters according to the present invention;
FIG. 3(a) is a schematic diagram of the output voltage corresponding to the current vector of the first current source type matrix converter according to the present invention (v)ab＞vac)；
FIG. 3(b) is a schematic diagram of the output voltage corresponding to the current vector of the second current source type matrix converter according to the present invention (v)ac＞vab)；
Fig. 4 is a schematic diagram of current flow paths of two current source matrix converters based on the space current vector modulation method in the operating mode of the present invention, wherein fig. 4(a) -fig. 4(f) are schematic diagrams of current flow paths in the operating modes of state 1-state 6, respectively;
FIG. 5.1(a) shows the input voltage V of the current source matrix converter of the present inventionpAnd an output voltage VsAn experimental oscillogram;
FIG. 5.1(b) is a diagram showing an input voltage V of the current source matrix converter of the present inventionpAnd current ipAn experimental oscillogram;
fig. 5.2 is a simulation waveform diagram of the current source type high frequency isolation matrix type cascaded converter of the present invention.
Detailed Description
The technical solutions in the embodiments of the present invention will be clearly and completely described below with reference to the drawings in the embodiments of the present invention, and it is obvious that the described embodiments are only a part of the embodiments of the present invention, and not all of the embodiments. All other embodiments, which can be derived by a person skilled in the art from the embodiments given herein without making any creative effort, shall fall within the protection scope of the present invention.
As shown in fig. 1, a current source type high frequency isolation matrix type cascaded converter includes a three-phase power grid 1.1, an LC filter 1.2, two sets of matrix converters (1.3, 1.4), leakage inductances (1.5, 1.6), high frequency isolation transformers (1.7, 1.8), two sets of uncontrolled rectifier bridges (1.9, 1.10), and resistance-inductance loads (1.11, 1.12).
The three-phase power grid 1.1 is connected with the LC filter 1.2;
the LC filter 1.2 is respectively connected with the first matrix converter 1.3 and the second matrix converter 1.4 in parallel; the matrix converters (1.3 and 1.4) are respectively connected with high-frequency isolation transformers (1.7 and 1.8); the secondary sides of the high-frequency isolation transformers (1.7 and 1.8) are connected with uncontrolled rectifier bridges (1.9 and 1.10) and transmit energy through magnetic fields; the uncontrolled rectifier bridges (1.9, 1.10) are respectively connected with a resistance-inductance load 1.11 and a resistance-inductance load 1.12; the inductance-resistance load 1.11 and the inductance-resistance load 1.12 on the direct current output side are mutually connected in series to form a closed loop;
the matrix converters 1.3 and 1.4 at the input side of the high-frequency isolation transformer are connected in parallel;
direct current bus inductors 1.11 and 1.12 at the output side of the high-frequency isolation transformer are mutually connected in series;
the power direction and the power magnitude of the current source type high-frequency isolation matrix type cascade converter are determined by the control modules of the matrix converters 1.3 and 1.4;
the current of the direct current side output resistance-inductance loads 1.11 and 1.12 corresponding to the direct current bus inductor is controlled by a current loop.
The current source type matrix converter comprises a first current source type matrix converter 1.3 and a second current source type matrix converter 1.4, the high-frequency isolation transformer comprises a first high-frequency isolation transformer 1.7 and a second high-frequency isolation transformer 1.8, the rectifying circuit comprises a first uncontrolled rectifier bridge 1.9 and a second uncontrolled rectifier bridge 1.10, wherein:
the alternating current output side of the first current source type matrix converter 1.3 is connected with a first high-frequency isolation transformer 1.7 in series;
the alternating current output side of the second current source type matrix converter 1.4 is connected with a second high-frequency isolation transformer 1.8 in series;
the alternating current input side of the first current source type matrix converter 1.3 is connected with the alternating current input side of the second current source type matrix converter 1.4 in parallel;
the output end of the secondary side of the first high-frequency isolation transformer 1.7 is connected with a first uncontrolled rectifier bridge 1.9 in series;
the output end of the secondary side of the second high-frequency isolation transformer 1.9 is connected with a second uncontrolled rectifier bridge 1.10 in series;
the output side of the first uncontrolled rectifier bridge 1.9 and the output side of the second uncontrolled rectifier bridge 1.10 are connected in series through two sets of inductance-resistance loads 1.11 and 1.12 to form a closed loop;
the first current source type matrix converter 1.3 and the second current source type matrix converter 1.4 are respectively connected with the port of the three-phase LC filter circuit 1.2 and receive electric energy transmitted by a three-phase power grid.
As shown in fig. 2, 5.1 and 5.2, the current closed-loop control method adopted by the control strategy of the first current source type matrix converter 1.3 comprises the following steps:
1) the capacitor voltage U passing through the three-phase LC filter 1.21abcAnd the network voltage VgElectric angle theta obtained by phase-locked loopeObtaining the capacitance voltage d-axis component U of the LC filter 1.2 through coordinate transformation 2.11dAnd q-axis component U1q；
2) Component U under filter capacitor voltage dq coordinate systemdAnd UqThe low-frequency component of the capacitor voltage, the electrical angle theta, is obtained through a low-pass filter 2.2eObtaining the electrical angular velocity omega after differentiating by 2.3eAnd 2.6, the low-frequency capacitance current of the filter capacitor can be obtained
Figure BDA0003033822870000071
And
Figure BDA0003033822870000072
3) given load side DC bus current Idc *And the actual DC bus current IdcThe error value between the two is processed by the PI controller 2.5 to obtain the d-axis direct current component
Figure BDA0003033822870000073
To realize the transmission of unit power factor, let the reference value Q of reactive powerrefIs 0, QrefAnd the network voltage VgThe q-axis current component is obtained after 2.4 calculation
Figure BDA0003033822870000074
Is zero;
4) the dq axis component of the capacitor voltage passes through a high-pass filter 2.7 to obtain a high-frequency component U of the capacitor voltagehdAnd UhqThe high-frequency components of the capacitor voltage are respectively multiplied by a same virtual resistance coefficient kpv(2.8) obtaining dq axis component values of the virtual current (2.9), wherein the fifth harmonic and the seventh harmonic of the system current can be eliminated through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Figure BDA0003033822870000075
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
Figure BDA0003033822870000076
And
Figure BDA0003033822870000077
obtaining the given value of the final current at the input side of the matrix converter through operation, and obtaining the phase current fundamental wave peak value I after the conversion of polar coordinates 2.171dc *And phase angle thetaα1；
6) Phase current fundamental wave peak value I1dc *Divided by a given value of DC current Idc *2.18 obtaining modulation ratio m of current source type matrix converter1iPhase angle thetaα1Plus the electrical angle theta measured by the phase-locked loope2.19 obtaining the switching pulse phase angle theta of the matrix converter1ωUsing the modulation ratio m1iAnd angle theta1ωAnd a switching period TsTwelve switching pulses of the first current source matrix converter 1.3 are generated.
As shown in fig. 2, the dc current control method adopted by the control strategy of the second current source type matrix converter 1.4 includes the following steps:
1) the capacitor voltage U passing through the three-phase LC filter 1.2abcAnd the network voltage VgElectric angle theta obtained by phase-locked loopeObtaining the capacitance voltage d-axis component U of the LC filter 1.2 through coordinate transformation 2.10dAnd q-axis component Uq；
2) Component U under filter capacitor voltage dq coordinate systemdAnd UqThe low-frequency component of the capacitor voltage, the electrical angle theta, is obtained through a low-pass filter 2.12eObtaining the electrical angular velocity omega after differentiating by 2.11eAnd 2.13 calculation is carried out to obtain the low-frequency capacitance current of the filter capacitor
Figure BDA0003033822870000078
And
Figure BDA0003033822870000079
3) given load side DC bus current Idc *And the actual DC bus current IdcThe error value between the two is processed by the PI controller 2.5 to obtain the d-axis direct current component
Figure BDA00030338228700000710
To realize the transmission of unit power factor, let the reference value Q of reactive powerrefIs 0, QrefAnd the network voltage VgThe q-axis current component is obtained after 2.4 calculation
Figure BDA00030338228700000711
Is zero;
4) the dq axis component of the capacitor voltage passes through a high pass filter 2.14 to obtain capacitor electricityHigh frequency component of voltage UhdAnd UhqThe high-frequency components of the capacitor voltage are respectively multiplied by a same virtual resistance coefficient kpv(2.15) (2.16) obtaining dq axis component values of the virtual current, wherein the fifth harmonic and the seventh harmonic of the system current can be eliminated through an active damping scheme of the harmonic of the virtual resistance suppression circuit;
5) d-axis DC component
Figure BDA0003033822870000081
Compensating the upper low frequency capacitor current by combining the dq component of the high frequency capacitor voltage flowing through the virtual resistor current
Figure BDA0003033822870000082
And
Figure BDA0003033822870000083
obtaining the given value of the final current at the input side of the matrix converter through operation, and obtaining the phase current fundamental wave peak value I after the conversion of polar coordinates 2.202dc *And phase angle thetaα2；
6) Phase current fundamental wave peak value I2dc *Divided by a given value of DC current Idc *2.21 obtaining modulation ratio m of Current Source type matrix converter2iPhase angle thetaα2Plus the electrical angle theta measured by the phase-locked loope2.22 obtaining the switching pulse phase angle theta of the matrix converter2ωUsing the modulation ratio m2iAnd angle theta2ωAnd a switching period TsTwelve switching pulses of the second current source matrix converter 1.4 are generated.
The space vector modulation method of the matrix converters 1.3 and 1.4 comprises the following steps:
for simplicity of analysis, the three current vectors acting on the first current source matrix converter 1.3 in a switching cycle are each I11，I12，I10Corresponding to the output voltage of the first current source type matrix converter 1.3 as U11，U12，U10By changing the order of action of the current vectors, U is made11>U12>U10(ii) a In the same way, the method for preparing the composite material,three current vectors acting on the second current source matrix converter 1.4 in one switching cycle are respectively I21，I22，I20Corresponding to the input voltage of the second current source type matrix converter 1.4 as U21，U22，U20By changing the order of action of the current vectors, U is made21>U22>U20. Since the first current source matrix converter 1.3 and the second current source matrix converter 1.4 are connected in parallel, the output voltage waveforms of the matrix converters in one switching period are consistent, and the operating states are consistent, taking the operating state of the first matrix converter as an example, the specific operating process of the soft switch in one switching period is as follows, it is not assumed that the first current source matrix converter 1.3 operates in the first sector, and at this time, the second current source matrix converter 1.4 also operates in the first sector.
As shown in fig. 3(a) (b) and fig. 4 (light color for off and dark color for on):
1) state 0: primary side commutation (t)0 -)
Just after the start of a switching cycle, the current vectors corresponding to the first current source matrix converter 1.3 and the second current source matrix converter 1.4 are both zero vectors I7Output voltage v of matrix converterpIs reduced to 0;
2) state 1 (3.1): switch tube on-time (t)0-t1)
The zero current vector still acts, the primary side input side current and the secondary side output side current of the first high-frequency isolation transformer 1.7 are equal, and the primary side current of the first high-frequency isolation transformer 1.7 flows through the switching tube S of the first current source type matrix converter 1.311、S22And S14、S24The current of the secondary side rectifier bridge of the first high-frequency isolation transformer 1.7 flows through D1、D4Three-phase capacitors provide current paths to three-phase inductors, this state being free of energy transmission, S11And S14Conducting at zero voltage;
3) state 2 (3.2): switch tube on-time (t)1-t2)
First current source type matrix converter 1Zero vector I of 37End of action, current vector I1In operation, the inductive current of the first uncontrolled rectifier bridge 1.9 flows through the diode D1、D4According to a four-step commutation, the switching tube S of the first current source matrix converter 1.311And S14Is turned off because the voltage u is nowabGreater than 0, current to S16And S26The output capacitor of the matrix converter is charged, and the matrix converter power switch tube S16And S26Zero voltage conduction, when the input side voltage of the first high-frequency isolation transformer 1.7 is equal to uabEnergy flows from the grid;
4) state 3 (3.3): switch tube on-time (t)2-t3)
Similar to the operating state of state 2, the effective current vector I of the first current source matrix converter 1.31End of action, current vector I2In operation, the inductive current of the first uncontrolled rectifier bridge 1.9 flows through the diode D1、D4According to a four-step commutation, the switching tube S of the first current source matrix converter 1.316And S26Closing, S12And S22Zero voltage conduction, when the input side voltage of the first high-frequency isolation transformer 1.7 is equal to uacEnergy flows from the grid;
5) state 4 (3.4): commutation of an uncontrolled rectifier bridge (t)3-t4)
The rectification of the first uncontrolled rectifier bridge 1.9 is completed with the aid of the first current source matrix converter 1.3, at t3At the moment, the first uncontrolled rectifier bridge 1.9 diode D2、D3Zero current conduction. The current flowing through the inductor of the first uncontrolled rectifier bridge 1.9 decreases linearly through D2、D3Through D1、D4Is linearly decreased at the same rate, commutation overlap time T in this contextdSelecting 100ns, and commutating current on an inductor 1.9 of the first uncontrolled rectifier bridge before the mode is finished;
6) state 5 (3.5): switch tube on-time (t)4-t5)
Commutation overlap time TdAfter the completion of the process, the operation,power switch tube S of first current source type matrix converter 1.312And S22Off, S14And S24Zero voltage conduction, the voltage drop of the primary side input side of the first high-frequency isolation transformer 1.7 is 0, and no direct current energy is transmitted under the mode;
7) state 6 (3.6): switch tube on-time (t)5-t6)
First current source matrix converter 1.3 current vector I2End of action, zero vector I0In operation, a direct current flows on the load side through the diode D of the first uncontrolled rectifier bridge 1.92And D3While the input side current of the first high-frequency isolation transformer 1.7 flows through the switching tube S of the first current source type matrix converter 1.311、S21And S24、S14And the three-phase capacitor in the LC filter 1.2 provides a current channel for the three-phase inductor.
The foregoing shows and describes the general principles, essential features, and advantages of the invention. It will be understood by those skilled in the art that the present invention is not limited to the embodiments described above, which are described in the specification and illustrated only to illustrate the principle of the present invention, but that various changes and modifications may be made therein without departing from the spirit and scope of the present invention, which fall within the scope of the invention as claimed.

Claims (5)

1.一种电流源型高频隔离矩阵型级联变换器，包括三相电网(1.1)、LC滤波器(1.2)、第一电流源型矩阵变换器(1.3)与第二电流源型矩阵变换器(1.4)、第一漏感(1.5)与第二漏感(1.6)、第一高频隔离变压器(1.7)与第二高频隔离变压器(1.8)、第一不控整流桥(1.9)与第二不控整流桥(1.10)、第一阻感负载(1.11)与第二阻感负载(1.12)，其特征在于，所述三相电网(1.1)与LC滤波器(1.2)相连；LC滤波器(1.2)分别与第一电流源型矩阵变换器(1.3)与第二电流源型矩阵变换器(1.4)并联连接；第一电流源型矩阵变换器(1.3)的交流输出侧与第一高频隔离变压器(1.7)串联，第二电流源型矩阵变换器(1.4)的交流输出侧与第二高频隔离变压器(1.8)串联；第一高频隔离变压器(1.7)副级输出侧与第一不控整流桥(1.9)串联，第二高频隔离变压器(1.8)副级输出侧与第二不控整流桥(1.10)串联，通过磁场传递能量；第一不控整流桥(1.9)输出端和第二不控整流桥(1.10)输出端通过第一阻感负载(1.11)和第二阻感负载(1.12)串联形成闭合回路。1. A current source type high frequency isolation matrix type cascade converter, comprising a three-phase power grid (1.1), an LC filter (1.2), a first current source type matrix converter (1.3) and a second current source type matrix Converter (1.4), first leakage inductance (1.5) and second leakage inductance (1.6), first high frequency isolation transformer (1.7) and second high frequency isolation transformer (1.8), first uncontrolled rectifier bridge (1.9 ) and the second uncontrolled rectifier bridge (1.10), the first resistance-inductive load (1.11) and the second resistance-inductive load (1.12), characterized in that the three-phase power grid (1.1) is connected to the LC filter (1.2) ; LC filters (1.2) are respectively connected in parallel with the first current source matrix converter (1.3) and the second current source matrix converter (1.4); the AC output side of the first current source matrix converter (1.3) connected in series with the first high-frequency isolation transformer (1.7), and the AC output side of the second current source matrix converter (1.4) is connected in series with the second high-frequency isolation transformer (1.8); the first high-frequency isolation transformer (1.7) has a secondary stage The output side is connected in series with the first uncontrolled rectifier bridge (1.9), and the secondary output side of the second high-frequency isolation transformer (1.8) is connected in series with the second uncontrolled rectifier bridge (1.10) to transfer energy through the magnetic field; the first uncontrolled rectifier bridge (1.9) The output end and the output end of the second uncontrolled rectifier bridge (1.10) are connected in series to form a closed loop through the first resistive-inductive load (1.11) and the second resistive-inductive load (1.12). 2.根据权利要求1所述的一种电流源型高频隔离矩阵型级联变换器，其特征在于，所述电流源型高频隔离矩阵型级联变换器的功率方向和功率大小由第一电流源型矩阵变换器(1.3)与第二电流源型矩阵变换器(1.4)的控制模块决定；第一电流源型矩阵变换器(1.3)和第二电流源型矩阵变换器(1.4)分别与LC滤波电路(1.2)端口连接并接收三相电网传输的电能；直流侧输出第一阻感负载(1.11)与第二阻感负载(1.12)对应直流母线电感的电流由电流环控制。2. The current source type high frequency isolation matrix cascaded converter according to claim 1, wherein the power direction and power size of the current source type high frequency isolation matrix cascade converter are determined by the A current source matrix converter (1.3) and a control module of the second current source matrix converter (1.4) are determined; the first current source matrix converter (1.3) and the second current source matrix converter (1.4) It is respectively connected to the port of the LC filter circuit (1.2) and receives the electric energy transmitted by the three-phase power grid; the current of the DC bus inductance corresponding to the output of the first resistance-inductive load (1.11) and the second resistance-inductive load (1.12) on the DC side is controlled by the current loop. 3.根据权利要求2所述的一种电流源型高频隔离矩阵型级联变换器，其特征在于，所述第一电流源型矩阵变换器(1.3)的控制模块所采用的控制方法包括以下过程：3. A current source type high frequency isolation matrix type cascaded converter according to claim 2, wherein the control method adopted by the control module of the first current source type matrix converter (1.3) comprises the following steps: The following process: 1)经过三相LC滤波器(1.2)的电容电压Uabc和电网电压Vg经过锁相环得到的电角度θe经过坐标变换(2.1)得到LC滤波器(1.2)的电容电压d轴分量Ud和q轴分量Uq；1) After the three-phase LC filter (1.2), the capacitor voltage U abc and the grid voltage V g are obtained through the phase-locked loop to obtain the electrical angle θ e. After coordinate transformation (2.1), the d-axis component of the capacitor voltage of the LC filter (1.2) is obtained U d and q-axis components U q ; 2)滤波电容电压dq坐标系下的分量Ud和Uq经过低通滤波器(2.2)得到电容电压的低频分量，电角度θe经过微分(2.3)后得到电角速度ωe，经过(2.6)计算后得到滤波电容的低频电容电流
Figure FDA0003033822860000011
和
Figure FDA0003033822860000012
2) The components U d and U q in the filter capacitor voltage dq coordinate system pass through the low-pass filter (2.2) to obtain the low-frequency components of the capacitor voltage, and the electrical angle θ e is differentiated (2.3) to obtain the electrical angular velocity ω e , after (2.6 ) is calculated to obtain the low-frequency capacitor current of the filter capacitor
Figure FDA0003033822860000011
and
Figure FDA0003033822860000012
 3)给定负载侧直流母线电流Idc *和实际直流母线电流Idc之间的误差值经过PI控制器(2.5)得到d轴直流电流分量
Figure FDA0003033822860000013
为了实现单位功率因数传输，令无功功率参考值Qref为0，Qref与电网电压Vg经(2.4)计算后得到q轴电流分量
Figure FDA0003033822860000014
为零；3) The error value between the given load-side DC bus current I dc * and the actual DC bus current I dc is obtained through the PI controller (2.5) to obtain the d-axis DC current component
Figure FDA0003033822860000013
In order to achieve unity power factor transmission, let the reactive power reference value Q ref be 0, and Q ref and grid voltage V g are calculated by (2.4) to obtain the q-axis current component
Figure FDA0003033822860000014
zero; 4)电容电压的dq轴分量经过高通滤波器(2.7)得到电容电压的高频分量Uhd和Uhq，电容电压高频分量分别乘以一个相同的虚拟电阻系数kpv(2.8)和kpv(2.9)得到虚拟电流的dq轴分量值，此通过虚拟电阻抑制电路谐波的主动阻尼方案消除系统电流的五、七次谐波；4) The high-frequency components U hd and U hq of the capacitor voltage are obtained through the high-pass filter (2.7) of the dq-axis component of the capacitor voltage, and the high-frequency components of the capacitor voltage are multiplied by the same virtual resistivity k pv (2.8) and k pv respectively. (2.9) Obtain the dq-axis component value of the virtual current, which eliminates the fifth and seventh harmonics of the system current through the active damping scheme of the virtual resistance to suppress the harmonics of the circuit; 5)d轴直流电流分量
Figure FDA0003033822860000021
结合高频电容电压流经虚拟电阻电流的dq分量再补偿上低频电容电流
Figure FDA0003033822860000022
和
Figure FDA0003033822860000023
经过运算得到矩阵变换器输入侧最终电流的给定值，再经过极坐标(2.17)转换后得到相电流基波峰值I1dc *和相角θα1；5) d-axis DC current component
Figure FDA0003033822860000021
Combined with the dq component of the high frequency capacitor voltage flowing through the virtual resistor current to compensate the upper low frequency capacitor current
Figure FDA0003033822860000022
and
Figure FDA0003033822860000023
The given value of the final current at the input side of the matrix converter is obtained through operation, and then the phase current fundamental wave peak value I 1dc * and the phase angle θ α1 are obtained after polar coordinate (2.17) transformation; 6)相电流基波峰值I1dc *除以直流电流的给定值Idc *(2.18)得到第一电流源型矩阵变换器(1.3)的调制比m1i，相角θα1加上锁相环测得的电角度θe(2.19)得到矩阵变换器开关脉冲相角θ1ω，利用调制比m1i和角度θ1ω以及开关周期Ts生成第一电流源型矩阵变换器(1.3)的十二路开关脉冲。6) Divide the phase current fundamental wave peak value I 1dc * by the given value of the DC current I dc * (2.18) to obtain the modulation ratio m 1i of the first current source matrix converter (1.3), add the phase angle θ α1 to the phase lock The measured electrical angle θ e (2.19) of the loop obtains the switching pulse phase angle θ 1ω of the matrix converter, and the modulation ratio m 1i and the angle θ 1ω and the switching period Ts are used to generate the first current source matrix converter (1.3) Twelve circuit switching pulse. 4.根据权利要求2所述的一种电流源型高频隔离矩阵型级联变换器，其特征在于，所述第二电流源型矩阵变换器(1.4)的控制模块所采用控制方法包括以下过程：4. A current source type high frequency isolation matrix type cascaded converter according to claim 2, characterized in that, the control method adopted by the control module of the second current source type matrix converter (1.4) comprises the following: process: 1)经过三相LC滤波器(1.2)的电容电压Uabc和电网电压Vg经过锁相环得到的电角度θe经过坐标变换(2.10)得到LC滤波器(1.2)的电容电压d轴分量Ud和q轴分量Uq；1) After the three-phase LC filter (1.2), the capacitor voltage U abc and the grid voltage V g are obtained through the phase-locked loop to obtain the electrical angle θ e. After coordinate transformation (2.10), the d-axis component of the capacitor voltage of the LC filter (1.2) is obtained U d and q-axis components U q ; 2)滤波电容电压dq坐标系下的分量Ud和Uq经过低通滤波器(2.12)得到电容电压的低频分量，电角度θe经过微分(2.11)后得到电角速度ωe，经过(2.13)计算后可以得到滤波电容的低频电容电流
Figure FDA0003033822860000024
和
Figure FDA0003033822860000025
2) The components U d and U q in the filter capacitor voltage dq coordinate system pass through the low-pass filter (2.12) to obtain the low-frequency component of the capacitor voltage, and the electrical angle θ e is differentiated (2.11) to obtain the electrical angular velocity ω e , after (2.13) ), the low-frequency capacitor current of the filter capacitor can be obtained after calculation
Figure FDA0003033822860000024
and
Figure FDA0003033822860000025
 3)给定负载侧直流母线电流Idc *和实际直流母线电流Idc之间的误差值经过PI控制器(2.5)得到d轴直流电流分量
Figure FDA0003033822860000026
为了实现单位功率因数传输，令无功功率参考值Qref为0，Qref与电网电压Vg经(2.4)计算后得到q轴电流分量
Figure FDA0003033822860000027
为零；3) The error value between the given load-side DC bus current I dc * and the actual DC bus current I dc is obtained through the PI controller (2.5) to obtain the d-axis DC current component
Figure FDA0003033822860000026
In order to achieve unity power factor transmission, let the reactive power reference value Q ref be 0, and Q ref and grid voltage V g are calculated by (2.4) to obtain the q-axis current component
Figure FDA0003033822860000027
zero; 4)电容电压的dq轴分量经过高通滤波器(2.14)得到电容电压的高频分量Uhd和Uhq，电容电压高频分量分别乘以一个相同的虚拟电阻系数kpv(2.15)和kpv(2.16)得到虚拟电流的dq轴分量值，此通过虚拟电阻抑制电路谐波的主动阻尼方案消除系统电流的五、七次谐波；4) The high-frequency components U hd and U hq of the capacitor voltage are obtained through the high-pass filter (2.14) of the dq-axis component of the capacitor voltage, and the high-frequency components of the capacitor voltage are multiplied by an identical virtual resistivity k pv (2.15) and k pv respectively. (2.16) Obtain the dq-axis component value of the virtual current, which eliminates the fifth and seventh harmonics of the system current through the active damping scheme of the virtual resistance to suppress the harmonics of the circuit; 5)d轴直流电流分量
Figure FDA0003033822860000028
结合高频电容电压流经虚拟电阻电流的dq分量再补偿上低频电容电流
Figure FDA0003033822860000029
和
Figure FDA00030338228600000210
经过运算得到矩阵变换器输入侧最终电流的给定值，再经过极坐标(2.20)转换后得到相电流基波峰值I2dc *和相角θα2；5) d-axis DC current component
Figure FDA0003033822860000028
Combined with the dq component of the high frequency capacitor voltage flowing through the virtual resistor current to compensate the upper low frequency capacitor current
Figure FDA0003033822860000029
and
Figure FDA00030338228600000210
The given value of the final current at the input side of the matrix converter is obtained through operation, and then the fundamental wave peak value I 2dc * of the phase current and the phase angle θ α2 are obtained after the polar coordinate (2.20) transformation; 6)相电流基波峰值I2dc *除以直流电流的给定值Idc *(2.21)得到第二电流源型矩阵变换器(1.4)的调制比m2i，相角θα2加上锁相环测得的电角度θe(2.22)得到矩阵变换器开关脉冲相角θ2ω，利用调制比m2i和角度θ2ω以及开关周期Ts生成第二电流源型矩阵变换器(1.4)的十二路开关脉冲。6) The phase current fundamental wave peak value I 2dc * is divided by the given value of the DC current I dc * (2.21) to obtain the modulation ratio m 2i of the second current source matrix converter (1.4), the phase angle θ α2 plus the phase lock The electrical angle θ e (2.22) measured by the loop can obtain the switching pulse phase angle θ 2ω of the matrix converter, and the modulation ratio m 2i and the angle θ 2ω and the switching period Ts are used to generate the second current source matrix converter (1.4) twelve circuit switching pulse. 5.根据权利要求1所述的一种电流源型高频隔离矩阵型级联变换器，其特征在于，所述第一电流源型矩阵变换器(1.3)和第二电流源型矩阵变换器(1.4)采用的调制方法包括以下步骤：5. A current source type high frequency isolation matrix type cascade converter according to claim 1, characterized in that the first current source type matrix converter (1.3) and the second current source type matrix converter are (1.4) The modulation method adopted includes the following steps: 一个开关周期内作用于第一电流源型矩阵变换器(1.3)的三个电流矢量分别为I11，I12，I10，对应第一电流源型矩阵变换器(1.3)输出电压为U11，U12，U10，通过改变电流矢量作用顺序，使得U11>U12>U10；同理，一个开关周期内作用于第二电流源型矩阵变换器(1.4)的三个电流矢量分别为I21，I22，I20，对应第二电流源型矩阵变换器(1.4)输入电压为U21，U22，U20，通过改变电流矢量作用顺序，使得U21>U22>U20；由于第一电流源型矩阵变换器(1.3)和第二电流源型矩阵变换器(1.4)并联，在一个开关周期内矩阵变换器输出电压波形一致，工作状态一致，以第一电流源型矩阵变换器(1.3)的工作状态为例，一个开关周期内软开关具体工作过程如下，The three current vectors acting on the first current source matrix converter (1.3) in one switching cycle are I 11 , I 12 , and I 10 respectively, and the corresponding output voltage of the first current source matrix converter (1.3) is U 11 , U 12 , U 10 , by changing the current vector action sequence, U 11 >U 12 >U 10 ; in the same way, the three current vectors acting on the second current source matrix converter (1.4) in one switching cycle are respectively are I 21 , I 22 , I 20 , corresponding to the input voltages of the second current source matrix converter (1.4) U 21 , U 22 , U 20 , by changing the current vector action sequence, U 21 >U 22 >U 20 ; Since the first current source matrix converter (1.3) and the second current source matrix converter (1.4) are connected in parallel, the output voltage waveforms of the matrix converters in one switching cycle are the same, and the working states are the same. Taking the working state of the matrix converter (1.3) as an example, the specific working process of soft switching in one switching cycle is as follows: 1)状态0：初级侧换流(t0 -)1) State 0: Primary side commutation (t 0 - ) 一个开关周期刚刚开始，第一电流源型矩阵变换器(1.3)和第二电流源型矩阵变换器(1.4)对应的电流矢量均为零矢量I7，矩阵变换器输出电压vp降为0；A switching cycle has just started, the current vectors corresponding to the first current source matrix converter (1.3) and the second current source matrix converter (1.4) are both zero vector I 7 , and the output voltage v p of the matrix converter drops to 0 ; 2)状态1(3.1)：开关管导通时间(t0-t1)2) State 1 (3.1): switch on time (t 0 -t 1 ) 零电流矢量仍然作用，第一高频隔离变压器(1.7)的原边输入侧电流和副边输出侧电流相等，第一高频隔离变压器(1.7)初级侧电流流过第一电流源型矩阵变换器(1.3)的开关管S11、S22和S14、S24，第一高频隔离变压器(1.7)副级侧整流桥电流流过D1、D4，三相电容给三相电感提供电流路径，这个状态无能量传输，S11和S14零电压导通；The zero current vector still works, the primary side input side current of the first high frequency isolation transformer (1.7) is equal to the secondary side output side current, and the primary side current of the first high frequency isolation transformer (1.7) flows through the first current source matrix transformation The switch tubes S 11 , S 22 and S 14 , S 24 of the transformer (1.3), the secondary side rectifier bridge current of the first high-frequency isolation transformer (1.7) flows through D 1 , D 4 , and the three-phase capacitor provides the three-phase inductance Current path, no energy transmission in this state, S 11 and S 14 are turned on at zero voltage; 3)状态2(3.2)：开关管导通时间(t1-t2)3) State 2 (3.2): switch on time (t 1 -t 2 ) 第一电流源型矩阵变换器(1.3)的零矢量I7作用结束，电流矢量I1开始作用，第一不控整流桥(1.9)的电感电流流经二极管D1、D4，根据四步换流，第一电流源型矩阵变换器(1.3)的开关管S11和S14关闭，因为此时电压uab＞0，电流给S16和S26的输出电容充电，第一电流源型矩阵变换器(1.3)的功率开关管S16和S26零电压导通，此时第一高频隔离变压器(1.7)的输入侧电压等于uab，能量从电网流出；The action of the zero vector I 7 of the first current source matrix converter (1.3) ends, the current vector I 1 begins to act, and the inductor current of the first uncontrolled rectifier bridge (1.9) flows through the diodes D 1 and D 4 . According to the four steps For commutation, the switches S11 and S14 of the first current source matrix converter (1.3) are turned off, because the voltage u ab > 0 at this time, the current charges the output capacitors of S16 and S26 , the first current source type The power switch tubes S 16 and S 26 of the matrix converter (1.3) are turned on at zero voltage, at this time the input side voltage of the first high-frequency isolation transformer (1.7) is equal to u ab , and the energy flows out from the grid; 4)状态3(3.3)：开关管导通时间(t2-t3)4) State 3 (3.3): switch on time (t 2 -t 3 ) 和状态2的工作状态类似，第一电流源型矩阵变换器(1.3)的有效电流矢量I1作用结束，电流矢量I2开始作用，第一不控整流桥(1.9)的电感电流流经二极管D1、D4，根据四步换流，第一电流源型矩阵变换器(1.3)的功率开关管S16和S26关闭，S12和S22零电压导通，此时第一高频隔离变压器(1.7)的输入侧电压等于uac，能量从电网流出；Similar to the working state of state 2, the effective current vector I 1 of the first current source matrix converter (1.3) ends, the current vector I 2 starts to work, and the inductor current of the first uncontrolled rectifier bridge (1.9) flows through the diode. D 1 , D 4 , according to the four-step commutation, the power switches S16 and S26 of the first current source matrix converter (1.3) are turned off, S12 and S22 are turned on at zero voltage, and the first high frequency The input side voltage of the isolation transformer (1.7) is equal to u ac , and the energy flows out from the grid; 5)状态4(3.4)：不控整流桥的整流换向(t3-t4)5) State 4 (3.4): rectification commutation of uncontrolled rectifier bridge (t 3 -t 4 ) 在第一电流源型矩阵变换器(1.3)的协助下第一不控整流桥(1.9)的整流完成，在t3时刻，第一不控整流桥(1.9)的二极管D2、D3零电流导通。第一不控整流桥(1.9)流过电感的电流线性下降，通过D2、D3的电流线性增大，通过D1、D4的电流以相同速率线性减小，换向重叠时间Td选择100ns，在模态结束前，第一不控整流桥(1.9)电感上的电流换向；The rectification of the first uncontrolled rectifier bridge (1.9) is completed with the assistance of the first current source matrix converter (1.3). At time t3 , the diodes D2 and D3 of the first uncontrolled rectifier bridge (1.9 ) are zero. current conducts. The current flowing through the inductor of the first uncontrolled rectifier bridge (1.9) decreases linearly, the current through D 2 and D 3 increases linearly, and the current through D 1 and D 4 decreases linearly at the same rate, and the commutation overlap time T d Select 100ns, before the end of the mode, the current commutation on the inductor of the first uncontrolled rectifier bridge (1.9); 6)状态5(3.5)：开关管导通时间(t4-t5)6) State 5 (3.5): switch on time (t 4 -t 5 ) 换向重叠时间Td结束后，第一电流源型矩阵变换器(1.3)的功率开关管S12和S22关断，S14和S24零电压导通，第一高频隔离变压器(1.7)原边输入侧电压降为0，在此模态下不传输直流能量；After the commutation overlap time T d ends, the power switches S12 and S22 of the first current source matrix converter (1.3) are turned off, S14 and S24 are turned on at zero voltage, and the first high-frequency isolation transformer (1.7 ) The voltage drop on the input side of the primary side is 0, and no DC energy is transmitted in this mode; 7)状态6(3.6)：开关管导通时间(t5-t6)7) State 6 (3.6): switch on time (t 5 -t 6 ) 第一电流源型矩阵变换器(1.3)电流矢量I2作用结束，零矢量I0开始作用，负载侧直流电流流经第一不控整流桥(1.9)的二极管D2和D3，同时第一高频隔离变压器(1.7)输入侧电流流经第一电流源型矩阵变换器(1.3)的功率开关管S11、S21和S24、S14，LC滤波器(1.2)中三相电容为三相电感提供电流通道。The current vector I 2 of the first current source matrix converter (1.3) ends, the zero vector I 0 starts to work, and the load-side DC current flows through the diodes D 2 and D 3 of the first uncontrolled rectifier bridge (1.9), and at the same time the first The current on the input side of a high-frequency isolation transformer (1.7) flows through the power switch tubes S 11 , S 21 and S 24 , S 14 of the first current source matrix converter ( 1.3 ), and the three-phase capacitors in the LC filter ( 1.2 ) Provides a current path for a three-phase inductor.
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