CN1130899C - Segment synchronous recovery network for HDTV receiver - Google Patents

Abstract

A system for processing a received VSB modulated signal containing high definition video data represented by clusters of vestigial sideband (VSB) symbols, said data having a data frame format consisting of a continuous data frame, the data frame Including a field sync component as the beginning of a plurality of data segments with associated segment sync components, the means comprising: a demodulator (22) for generating a demodulated signal in response to said received signal; means (430) for providing a correlated reference pattern during a segment sync interval; and a data correlator (420) for detecting said segment sync component responsive to said demodulated signal and said reference pattern; wherein said correlated reference pattern Occupies less than four symbol intervals.

Description

Apparatus and method for processing vestigial sideband modulated signal containing high-definition video data

field of invention

The present invention relates to a receiver system for processing high definition television (HDTV) signals of the vestigial sideband modulation type such as proposed by Major League Union.

Background technique

Data recovery from a modulated signal conveying digital information in symbols typically requires three functions in the receiver: timing recovery for symbol synchronization, carrier recovery (frequency demodulation to baseband), and channel equalization. Timing recovery is the process by which the receiver clock (time base) is synchronized with the transmitter clock. This allows the received signal to be sampled at optimal times to reduce clipping errors associated with decision-directed processing of received symbol values. Carrier recovery is the process by which a received RF signal is frequency shifted to baseband after downconversion to a lower IF passband (eg, closer to baseband) to allow recovery of modulated baseband information. Adaptive channel equalization is a process by which the effects of changing conditions and interference in a signal transmission channel are compensated. This process typically employs filters that remove amplitude and phase distortion due to the frequency-varying nature of the transmission channel, thereby providing improved symbol decision capability.

Contents of the invention

In accordance with the principles of the present invention, a system for processing received vestigial sideband (VSB) modulated signals containing high definition television information includes a segment responsive to an abbreviated (1,-1) correlated reference pattern Synchronize detection network.

According to a first aspect of the present invention there is provided an apparatus for processing a received vestigial sideband modulated signal containing high definition video data represented by clusters of vestigial sideband symbols having Data frame format, each data frame comprising a field sync component as the start of a plurality of data segments with associated segment sync components, characterized in that the means comprise:

- a demodulator for generating a demodulated signal in response to said received signal;

means for generating an associated reference pattern during segment sync intervals; and

- a data correlator for detecting said segment sync component in response to said demodulated signal and said reference pattern; wherein

The above correlation reference pattern occupies fewer symbol intervals than the number of symbols in the above segment sync component.

According to a second aspect of the present invention there is provided a method for processing a received vestigial sideband modulated signal containing high definition video data represented by clusters of vestigial sideband symbols having A data frame format, each data frame comprising a field sync component as the start of a plurality of data segments with associated segment sync components, characterized in that the method comprises the steps of:

demodulating the received vestigial sideband signal to generate a demodulated signal; and

correlating said demodulated signal of a segment sync component symbol pattern with an associated reference symbol pattern generated during a segment sync interval; wherein

Said correlated reference symbol pattern occupies fewer symbol intervals than the number of symbols in said segment sync component of said demodulated signal.

Brief description of the drawings

Fig. 1 is a block diagram of a portion of a high definition television receiver including an apparatus in accordance with the principles of the present invention.

Figure 2 depicts the data frame format for a VSB modulated signal according to the Major League HDTV system.

FIG. 3 shows the detailed structure of the digital demodulator/carrier recovery network in FIG. 1. FIG.

FIG. 4 shows the detailed structure of the segment sync detector and symbol clock recovery network in FIG. 1. FIG.

Figure 5 depicts signal waveforms that are helpful in understanding the operation of the network shown in Figure 4.

FIG. 6 shows the detailed structure of a compensation network for removing DC offset in the symbol data stream processed by the system of FIG. 1. FIG.

FIG. 7 shows the detailed structure of the NTSC co-channel interference detection network in the system of FIG. 1. FIG.

FIG. 8 shows the frequency spectrum associated with the operation of the network in FIG. 7 .

Detailed description of the preferred embodiment

In FIG. 1, a terrestrial broadcast analog input HDTV signal consists of an input network 14 comprising radio frequency tuning circuitry and comprising a double conversion tuner and appropriate automatic gain control (AGC) circuitry for producing an intermediate frequency (IF) passband output signal. IF processor 16 for processing. The received signal was a carrier-suppressed 8-VSB modulated signal as proposed by Grand Union and used in the United States. This VSB signal is represented by a one-dimensional constellation of data symbols, where only one axis contains quantized data to be recovered by the receiver. To simplify the figure, the signals used to clock the illustrated functional blocks are not shown.

As described in the Grand Alliance HDTV System Specification effective April 14, 1994, the VSB transmission system transmits data having a prescribed data frame format shown in FIG. 2 . A small pilot signal at the suppressed carrier frequency is added to the transmitted signal to facilitate carrier lock at the VSB receiver. Referring to FIG. 2, each data frame includes two fields, where each field includes 313 segments of 832 multi-level symbols. The first segment of each field is called the field sync segment, while the remaining 312 segments are called the data segment. Data segments typically comprise MPEG compliant packets. Each data segment consists of a four-symbol segment sync character followed by 828 data symbols. Each field segment consists of a four-symbol segment sync character, followed by a field sync component consisting of a predetermined 511-symbol pseudo-random number (PN) sequence and three predetermined 63-symbol PN sequences, where The middle one of the three predetermined 63-symbol PN sequences is inverted in successive fields. The VSB mode control signal (defining the size of the VSB symbol conformation) follows the last sequence of 63PN, followed by 96 saved symbols and 12 symbols copied from the previous field.

Continuing to refer to FIG. 1, the passband IF output signal from unit 16 is converted by an analog-to-digital converter (ADC) 19 into an oversampled digital symbol data stream. The output oversampled digital data stream of ADC19 is demodulated to baseband by all-digital demodulator/carrier recovery network 22 . This is done by an all digital phase locked loop in response to a small reference pilot carrier in the received VSB data stream. Unit 22 produces an output I-phase demodulated symbol data stream which will be described in more detail with reference to FIG. 3 .

The ADC19 oversamples the incoming 10.76 Msymbol/s VSB symbol stream by using a 21.52 MHz sampling clock that is twice the received symbol rate, thereby providing an oversampled 21.52 Msample/s stream of two samples per symbol . Using this sample-based processing of two sample values per symbol instead of symbol-by-symbol (one sample value per symbol) produces The associated advantageous operation has subsequent signal processing functions.

Associated with ADC 19 and demodulator 22 is segment synchronization and symbol clock recovery network 24 . The network 24 detects and separates the repetitive data segment sync components of each data frame from the random data. The segment sync is used to regenerate a properly phased 21.52 MHz clock which is used to control data stream symbol sampling by the analog-to-digital converter 19 . As discussed in connection with FIGS. 4 and 5, network 24 facilitates detection of segment sync using the abbreviated two-symbol correlation reference pattern and associated two-symbol data correlators.

As will be discussed with reference to FIG. 6, DC compensation unit 26 uses an adaptive tracking circuit to remove the DC offset component due to the pilot signal component from the demodulated VSB signal. Unit 28 detects the data field sync component by comparing each received data segment with an ideal field reference signal stored in the receiver's memory. The field sync signal also provides a training signal for the channel equalizer 34 in addition to the field sync.

NTSC interference detection and suppression is performed by unit 30 which will be discussed in more detail with reference to FIGS. 7 and 8 . The signal is then adaptively equalized by a channel equalizer 34 operating in a combination of blind, training and decision directional modes. Equalizer 34 may be of the type described in the Grand League HDTV System Specification and in the article "VSB Modem Subsystem Design for Grand League Digital Television Receivers" by W. Bretl et al., IEEE Transactions on Consumer Electronics, August 1995. Equalizer 34 may also be of the type described in co-pending US Patent Application Serial No. 09/102.885, filed June 23, 1998 by Shiue et al. The output data stream of detector 30 is down-converted prior to equalizer 34 to a sample/symbol (10.76 Msymbol/sec) data stream. This down-conversion may be accomplished by a suitable down-sampling network (not shown for simplicity of the figure).

Equalizer 34 corrects for channel distortion, but phase noise randomly rotates symbol clusters. Phase tracking network 36 removes residual phase and gain noise in the output signal of equalizer 34, including phase noise not removed by the preceding carrier recovery network by responding to the pilot signal. The phase corrected signal is then format decoded by unit 40 and deinterleaved by unit 42 . Reed-Solomon errors are corrected by unit 44 and descrambled (derandomized) by unit 46 . Afterwards, the decoded data stream is processed by unit 5 for audio, video and display.

Tuner 14, IF processor 16, field sync detector 28, equalizer 34, phase tracking loop 36, format decoder 40, deinterleaver 42, Reed-Solomon decoder 44, and descrambler 46 may employ Circuit types mentioned in the Major League HDTV System Specification, April 4, 1994, or in the aforementioned Bretl et al. article. Circuitry suitable for performing the functions of units 19 and 50 are well known.

Demodulation in unit 22 is performed by a fully digital automatic phase control (APC) loop for carrier recovery. The PLL uses the pilot component as a reference for the initial acquisition and uses a common phase detector for the phase acquisition. Embedded in the received data stream, the pilot signal contains data that exhibits a random, noise-like pattern. Random data is basically not considered by the filtering operation of the demodulator APC loop. The 10.76 Msymbol/s input signal to ADC 19 is a near-baseband signal with a VSB spectrum centered at 5.38 MHz and a pilot component at 2.69 MHz. At 21.52MHz, the input data stream is oversampled by ADC19 twice. In the demodulated data stream from unit 22, the pilot components have been frequency shifted down to DC.

FIG. 3 shows a detailed structure of the digital demodulator 22. As shown in FIG. The 8-VSB modulated and oversampled data symbol stream from ADC 19 , including the very low frequency pilot component, is applied to the input of Hilbert filter 320 and delay unit 322 . Filter 320 separates the incoming IF sample data stream into "I" (in-phase) and "Q" (90° out-of-phase) components. Delay 322 exhibits a delay that matches Hilbert filter 320 . The I and Q components are rotated to baseband by using complex multiplier 324 in the APC loop. Once the loop is synchronized, the output of multiplier 324 is the complex baseband signal. The data stream from the output I of the decoder 324 is used as the actual demodulator output and is also used to obtain the pilot component of the received data stream by using the low pass filter 326 . The data stream from the output Q of multiplier 324 is used to obtain the phase of the received signal.

In a phase control loop, the I and Q output signals from multiplier 324 are applied to low pass filters 326 and 328, respectively. Filters 326 and 328 are Nyquist low pass filters with a cutoff frequency of approximately 1 MHz and are used to reduce the bandwidth of the signal before 8:1 data downsampling is performed by units 330 and 332 . The downsampled Q signal is filtered by an automatic frequency control (AFC) filter 336 . After filtering, the Q signal is clipped by unit 338 to reduce the dynamic range requirements of phase detector 340 . Phase detector 340 detects and corrects the phase difference between the I and Q signals applied to its input and produces an output phase difference signal filtered by APC filter 344 such as a second stage low pass filter. The phase difference detected by unit 340 represents the frequency difference between the desired near DC pilot signal frequency and the received pilot signal frequency.

If the received pilot signal exhibits the desired near DC frequency, the AFC unit 336 produces no phase shift. The pilot components of the I and Q channels input to phase detector 340 will not deviate from a phase relationship of 90 degrees to each other, so phase detector 340 produces a zero or near zero phase error input signal. However, if the received pilot signal appears to be of the wrong frequency, the AFC unit 336 will produce a phase shift. This will cause an additional phase difference between the I and Q channel pilot signals applied to the input of phase detector 340 . Detector 340 produces an output error value in response to this phase difference.

The filtered phase error signal from filter 344 is upsampled 1:8 by interpolator 346 to the downsampling previously performed by computation units 330 and 332 such that NCO 348 operates at 21.52 MHz. The output of interpolator 346 is applied to the control input of NCO 348, which locally regenerates the pilot signal used to demodulate the received data stream. NCO 348 includes sine and cosine look-up tables for reproducing the pilot tone at the correct phase by responding to phase control signals from units 340, 344, and 346. The output of NCO 348 is controlled until the I and Q signal outputs of multiplier 324 are such that the phase error signal produced by detector 340 is effectively zero, indicating that a properly demodulated baseband I signal is present at the output of multiplier 324 .

In digital demodulator 22 , the main signal processor actually includes elements 336 , 338 , 340 and 344 . The 8:1 downsampling provided by units 330 and 332 facilitates savings in demodulator processing power and hardware, and by clocking APC loop elements 336, 338, 340, and 344 at a lower clock frequency, i.e. using 21.52 MHz/8 or 2.69MHz clocking instead of 21.52MHz clocking allows for efficient processing. The data transformation results in increased software efficiency when a digital signal processor (DSP) is used to implement the network 22, particularly the phase detection loop, for example by requiring proportionally fewer lines of instruction code. DSP loops can be used for other signal processing purposes. When an Application Specific Integrating Circuit (ASIC) is used to implement network 22, the data transformation results in reduced hardware and power requirements and a reduction in the surface area of the integrating circuit. It is advantageous for the demodulator to use the pilot component to achieve carrier recovery, and to use a feed-forward process rather than the more complex and time-consuming feedback process using a slicer to determine the data.

The demodulated I-channel data stream is applied to a segment sync and symbol clock recovery unit 24 which is shown in detail in FIGS. 4 and 5 . When the repeating data segment sync pulse is recovered from the random data pattern of the received data stream, the segment sync is used to control the sampling operation of the analog-to-digital converter 19 (FIG. 1) by regenerating twice the symbol rate for proper phasing. 21.52MHz sampling clock to achieve proper symbol timing. Figure 5 depicts eight levels (-7 to +7) of data segments with associated segment synchronization for an 8-VSB modulated ground station signal according to the Grand Alliance HDTV specification. Segment synchronization occurs at the beginning of each data segment and takes four symbol intervals. Segment syncs are defined by patterns 1, -1, -1, 1 corresponding to segment sync pulse amplitudes from +5 to -5.

The four symbol segments occur synchronously every 832 symbols, but are difficult to locate in the demodulated VSB digital data stream due to the random and noise-like nature of the data. To detect segment synchronization under such conditions, it is common to apply the demodulated I-channel data stream to one input of a data correlator, and apply a reference pattern with 1,-1,-1,1 characteristics to the correlator reference input of the register to compare with the demodulated data. The correlator produces an intensified coincidence with the reference pattern every 832 characters. Intensified data events are accumulated by an accumulator associated with the correlator. The interpolated random (reinforced) correlations disappear relative to the enhanced correlated segment-synchronous components. Networks for recovering segment sync data in this manner are known, for example, from the previously mentioned Major League HDTV specification and from the article by Bretl et al.

It can be seen here that while segment synchronization is generally difficult to locate, it is particularly difficult to detect when multipath ("ghosting") is present. Moreover, it can be seen from this that the last two properties (amplitude levels) (-1, 1) of the segment sync pattern are easily corrupted by transmission distortions such as multipath, but the first two properties of the segment sync pattern (1 , -1) are particularly hard to destroy. It can also be determined that even if the first two amplitude characteristics (1, -1) of the segment sync pattern are corrupted, they are generally corrupted in the same way, which makes the first two characteristics more easily detectable by correlation techniques. Thus, in the disclosed system, the reference pattern applied to the correlator for detecting segment synchronization is preferably composed of the first two pattern stages (1,-1) rather than all four pattern stages (1,- 1, -1, 1) form. Thus, the reference pattern of the correlator preferably only includes two symbol intervals.

In FIG. 4 , the oversampled output data stream from demodulator 22 ( FIGS. 1 and 3 ) is applied to one signal input of phase detector 410 and 832 symbol correlators 420 . The other signal input of the phase detector 410 receives input from a data correlation processing path comprising a correlator 420, an associated correlation reference pattern generator 430 connected to the reference input of the correlator 420, and a segment integrator and accumulator 424 Signal. The correlator 420 actually responds to symbol encoding data segment synchronization. The reference pattern generator 430 provides a simple abbreviated reference pattern 1,-1, thus allowing simpler correlator networks to be used. This simpler reference mode is less likely to cause confusion in the sync detection process, especially under poor signal conditions, because more stable and reliable information is used. The disclosed system is also unlikely to be confused if two of the four correlations are broken. Additionally, the computation time of correlator 420 is significantly reduced.

The output of correlator 420 is integrated and accumulated by unit 424 . A segment sync generator 428, including a comparator with a predetermined threshold, responds to the output of unit 424 by generating segment syncs at appropriate times in the data stream corresponding to data segment sync intervals. This condition occurs when the accumulation of intensified data events (segment synchronization extrinsic) exceeds a predetermined level. Phase detector 410 compares the phase of the segment sync produced by unit 428 with the phase of the segment sync appearing in the demodulated data stream from unit 22 and produces an output phase error signal. This error signal is low pass filtered by an automatic phase control (APC) filter 434 to produce a signal suitable for controlling a 21.52 MHz voltage controlled crystal oscillator (VCXO) 436 which provides a 21.52 MHz oversampling clock for ACD 19 . This sampling clock exhibits proper timing when the phase error signal contributed by the APC is substantially zero. Symbol timing (clock) recovery is done at this point. The segment sync generated by unit 428 is also applied to other decoder circuits including automatic gain control (AGC) circuits (not shown).

There is a DC offset in the demodulated output I symbol data from demodulator 22 due to the pilot component of the low frequency suppressed carrier in the received VSB signal. This DC offset is associated with each symbol and is removed by compensation network 26 (FIG. 1) before further processing. The cancellation of the DC component of the transmitted symbols facilitates the recovery of the symmetrical symbol values of the 8-VSB signal, namely ±7, ±5, ±3, ±1. Figure 6 shows the detailed structure of the network 26, which is actually a DC tracking feedback network. The arrangement of network 26 in FIG. 6 facilitates clocking at twice the symbol rate, thereby rapidly canceling the DC component. This operation facilitates rapid convergence of the receiver and its several independent subsystems to rapidly produce suitable operating conditions for processing received video data for display.

In FIG. 6, the oversampled demodulated data stream including unwanted DC bias is applied to one input of a subtractive combiner 610. One inverting input terminal (-) of the combiner 610 receives a DC compensation voltage from a DC voltage generator 616 through a control signal generated according to an output of the combiner 610 as follows. The DC offset in the output signal from combiner 610 is gradually attenuated by a feedback operation at a sampling rate twice the symbol rate. This DC offset is detected by unit 622 and compared to a reference by comparator 624 . The output of comparator 624 is indicative of the magnitude and polarity of the remaining DC bias and is used to generate control signals via control signal generator 626 . In turn, the control signal causes generator 616 to incrementally adjust the magnitude and polarity of the DC value combined with the demodulated data stream. This process continues until a steady state is reached in which DC value regulation is no longer provided by the feedback operating unit 616 . Generator 616 can provide positive and negative DC offset values, since transmission channel disturbances can cause changes in the (positive) DC bias added at the transmitter, so that both positive and negative offset values are required at the receiver.

FIG. 7 shows the detailed structure of the NTSC co-channel interference detection network 30 shown in FIG. 1. Referring to FIG. As explained in the Major League HDTV System Specification, the interference rejection performance of the VSB transmission system is based on the frequency location of the base components of the NTSC co-channel interfering signal within the 6 MHz television channel, and the baseband comb filter of the VSB receiver The periodic notch. These comb filter notches exhibit high attenuation (zeros) at frequencies where the interfering high-energy NTSC components are present. These components include a video carrier at 1.25MHz from the lower band edge, a color subcarrier at 3.58MHz above the video carrier frequency, and a sound carrier at 4.5MHz above the video carrier frequency.

NTSC interference is detected by the circuit shown in Figure 7, where the signal-to-interference plus noise of the field synchronous patterns is measured at the input and output of the comb filter network, and the patterns are compared with each other. The reference field sync pattern used for this purpose is a programmed and locally stored "ideal" version of the received VSB signal field sync pattern.

In FIG. 7, the oversampled demodulated I channel symbol data is applied to one input of NTSC rejection comb filter 710, a first input of multiplexer 745, and one input of difference combiner 720. Comb filter 710 includes a subtractor 712 which subtracts the samples delayed by delay element 714 from the input I data to produce a combed stream of I channel symbols. Comb filter 710 produces a significant amplitude attenuation or "null" at the aforementioned high energy interfering NTSC frequencies. The comb I data from filter 710 is applied to a second input of multiplexer 745 . The delay element 714 of the comb filter advantageously exhibits a 24-sample delay as will be described later.

A programmed 21.52 Msample/s (twice the symbol rate) reference field sync pattern is obtained from local memory during the field sync interval of the received data stream. The vertical sync reference pattern is applied to one input of NTSC band-stop comb filter 718, and to the inverting input (-) of combiner 720. Comb filter 718 is similar to comb filter 710 and also includes delay elements that advantageously exhibit a 24-sample delay. The network in Figure 7, in particular the comb filters 710, 718 and associated delay networks, are clocked at a frequency of 21.52 MHz.

A first error signal, produced at the output of combiner 720, represents the difference between the received field sync pattern and the reference field sync pattern in the input data stream. This error signal is squared by unit 722 and integrated by unit 724 . The second error signal produced at the output of combiner 730 represents the difference between the received field sync pattern after comb filtering by filter 710 and the reference field sync pattern after comb filtering by filter 718 . This second error signal is squared by unit 732 and integrated by unit 734 . The outputs of elements 722 and 732 represent the energy of the respective error signals. The integrated output signals of integrators 724 and 734 represent the signal-to-interference-plus-noise quantities of the uncomb-filtered and comb-filtered received field sync components, respectively. These integrated energy representative signals are applied to respective inputs of an energy detector (comparator) 740 which compares the magnitudes of the integrated first and second error signals. The output signal of the detector 740 is applied to the control input of the multiplexer 745 for providing the multiplexer 745 as a "data output" with one of its input signals exhibiting a higher quality, i.e. It has a good signal-to-noise plus interference ratio. Thus in the case of significant NTSC co-channel interference, the comb filtered output signal of filter 710 will be output from multiplexer 745, while the unfiltered received symbol data stream will be output in the absence of such interference.

The use of oversampled I channel data and field sync reference pattern data with 24-sample delays in comb filters 710 and 718 facilitates the generation of full spectral information about NTSC co-channel interference. This facilitates more accurate NTSC interference analysis and detection and better comb filtering. In particular, the 24-sample delay in comb filters 710 and 718 used with the oversampled input data and corresponding circuit timing produces a comb-filtered spectrum that is not provided by the 10.76 Msymbol/s symbol rate The input data stream is corrupted by phase and amplitude aliasing effects generated by comb filters 710 and 718 operating at a symbol rate of 10.76 Msymbols/sec. The resulting spectrum produced at the output of comb filters 710 and 718 is shown in FIG. 8 and includes two comb filtered full NTSC passband components centered around but separated from 10.76 MHz. An attenuation notch occurs at said high energy interfering NTSC frequency.

FIG. 7 shows one form of an NTSC co-channel interference detector comprising elements 722, 724, 732, 734, and 740. In FIG. However, other types of detectors may also be used. These elements can thus be represented by a four-input detector, a so-called "black box", which can be programmed to operate according to the requirements of a particular system. In this case, the four inputs are two oversampled (two samples/symbol) inputs connected to combiner 720, and two oversampled inputs connected to combiner 730, where the input to combiner 730 The output of the connected filter 710 is of particular importance.

As shown in Figure 8, the arrangement of Figure 7 produces a clean spectrum without the associated amplitude and phase corruption due to the frequency overlap of the upper band edge of the lower passband component with the lower band edge of the higher passband component ( Aliasing). Thus, elements 720, 722, 724, 730, 732, 734, and 740 perform better co-channel interference detection than a system employing a comb filter with a 12-sample delay processing input data at a symbol rate of 10.76 Msymbols/sec. The detection performed needs to be precise. In the latter case, amplitude and phase corruption may occur around 5.38MHz, where the upper and lower passband components overlap, at which point the passband components are not well matched and the overlap cannot be canceled. Such poor matching may occur in the case of signal channels that include, for example, multipath. This aliasing situation reduces the effectiveness of NTSC co-channel interference detection and is avoided by the disclosed system.

Claims (5)

1. An apparatus for processing a received vestigial-sideband modulated signal containing high-definition video data represented by clusters of vestigial-sideband symbols, said data having a data frame format consisting of a succession of data frames, each The data frame includes a field sync component as the start of a plurality of data segments with associated segment sync components, characterized in that the means comprise: - a demodulator (22) for generating a demodulated signal in response to said received signal; means (430) for generating a relative reference pattern during segment sync intervals; and - a data correlator (420) for detecting said segment sync component in response to said demodulated signal and said reference pattern; wherein The above correlation reference pattern occupies fewer symbol intervals than the number of symbols in the above segment sync component. 2. The device according to claim 1, characterized in that: the aforementioned segment sync intervals consist of four symbol intervals; and The above-mentioned relative reference pattern occupies the first two of the four symbol intervals. 3. The device according to claim 1, characterized in that: said segment sync component includes a four-symbol-spaced segment sync pattern 1, -1, -1, 1, where the pattern values 1 and -1 represent the relative amplitude levels of said segment sync component; and The correlation reference pattern described above is a 1,-1 pattern occupying two symbol intervals corresponding to the start of the segment sync interval. 4. A method for processing a received vestigial sideband modulated signal containing high definition video data represented by clusters of vestigial sideband symbols, said data having a data frame format consisting of a succession of data frames, each The data frame comprises a field sync component as the start of a plurality of data segments with associated segment sync components, characterized in that the method comprises the steps of: demodulating the received vestigial sideband modulated signal to generate a demodulated signal; and correlating said demodulated signal of a segment sync component symbol pattern with an associated reference symbol pattern generated during a segment sync interval; wherein Said correlated reference symbol pattern occupies fewer symbol intervals than the number of symbols in said segment sync component of said demodulated signal. 5. The method according to claim 4, characterized in that: said segment synchronization interval of said demodulated signal comprises four symbol intervals; The above-mentioned relative reference symbol pattern occupies the first two of the four symbol intervals of the demodulated signal.

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