CN112782687A

CN112782687A - Radar waveform generation circuit, method, radar, and computer storage medium

Info

Publication number: CN112782687A
Application number: CN202110292548.8A
Authority: CN
Inventors: 陈宏浩; 席厚金; 郧亚峰; 刘陈峰; 戚宇飞; 侯凡; 刘伟; 朱旻
Original assignee: Jiangsu Manku Technology Co ltd
Current assignee: Jiangsu Manku Technology Co ltd
Priority date: 2021-03-18
Filing date: 2021-03-18
Publication date: 2021-05-11
Anticipated expiration: 2041-03-18
Also published as: CN112782687B

Abstract

The present application provides a radar waveform generation circuit, method, radar and computer storage medium. In the radar waveform generation circuit, a frequency synthesizer sends a waveform completion signal according to the first enable signal sent by the second counter when the set waveform is completed; A counter sends a gating conversion signal to the gating device when the number of the waveform completion signal is counted; Counter; the second counter is used to send a second enabling signal to the frequency synthesizer when the time counting is completed after the time counting is performed according to the waveform completion signal and the time gap data. Through the difference in the time gap between the two set waveforms, the two sets of waveforms can be set to have different Doppler resolutions. After further processing of the echoes, the measurement interval can be effectively improved and the minimum measurement can be further refined. resolution.

Description

Radar waveform generation circuit, method, radar, and computer storage medium

Technical Field

The present application relates to the field of radar detection technologies, and in particular, to a radar waveform generation circuit, a radar waveform generation method, a radar, and a computer storage medium.

Background

Among millimeter wave radar products for vehicles, FMCW (frequency modulated continuous wave) radar is the most widely used one. The radar generates a frequency domain slope type signal (Chirp) with continuously changing frequency through a waveform generator and a Voltage Controlled Oscillator (VCO), the frequency domain slope type signal is transmitted to an external detection space through an antenna, and a slightly delayed (proportional to distance) obstacle echo signal and a transmitting signal are subjected to frequency mixing demodulation to obtain a baseband signal with constant frequency.

FMCW radars generally use a Fast Fourier Transform (FFT) to analyze a time-domain baseband signal once, and each FFT peak obtained by the analysis represents one or more obstacle targets.

In the FMCW radar, generally, FFT is used to combine the spectral lines of the first distance fourier transform (RFFT) targets into a virtual sampling sequence according to the time of each Chirp, FFT is performed, and each of the FFT spectral line peaks obtained through analysis represents one or more obstacle targets having the same distance but different doppler velocities.

In the radar signal processing method generally used above, the dynamic measurement range in the range dimension (maximum measured distance divided by range resolution) is determined by the number of spectral lines (bin) of the range FFT, and the dynamic measurement range in the doppler velocity dimension (maximum measured doppler velocity ÷ doppler velocity resolution) is determined by the number of spectral lines of the doppler FFT. At present, due to the obvious improvement of the system complexity of automatic driving and auxiliary driving, the market demand on the dynamic measurement range of vehicle-mounted radar products is increased day by day, and further the demand on the scale of the FFT spectral line number of each dimension is increased day by day. In engineering implementation, a larger dot size FFT basically means a higher memory (RAM) occupancy and also means a higher hardware cost. While a larger dot-size FFT also requires more time on the baseband signal to ADC sample, thus requiring a higher duty cycle and higher average transmit power. Generally, due to the limitation of engineering implementation, the dynamic measurement range of the vehicle-mounted radar product is difficult to be improved by expanding the point scale of each stage of FFT without limit, and the maximum measurement value and the minimum measurement resolution are difficult to be improved.

Disclosure of Invention

An object of the application is to provide a radar waveform generation circuit, a method, a radar and a computer storage medium, so as to solve the problem that the maximum measurement value and the minimum measurement resolution are difficult to be improved by a vehicle-mounted millimeter wave radar.

In one aspect, the present application provides a radar waveform generation circuit, in particular, the radar waveform generation circuit includes a gate, a first memory storing a first time gap, a second memory storing a second time gap, a first counter, a second counter, and a frequency synthesizer, wherein:

the frequency synthesizer is connected with the first counter and the second counter, generates a set waveform of a frequency modulation continuous wave according to a first-time enabling signal sent by the second counter, and sends a waveform completion signal to the first counter and the second counter when the set waveform is completed;

the first counter is used for counting the number of the waveform completion signals and sending a gating conversion signal to the gate when the number counting is completed;

the gate is used for gating the first time gap to the second time gap according to the gating conversion signal, so that the second time gap is used as time gap data to be sent to the second counter;

and the second counter is used for reading the time gap data according to the waveform completion signal, counting time according to the time gap data, and sending the enabling signal to the frequency synthesizer for the second time when the time counting is completed.

Optionally, the gate includes a first group of data input pins, a second group of data input pins, a selection pin, and a data output pin, the first group of data input pins is connected to the first memory, the second group of data input pins is connected to the second memory, the selection pin is connected to the first counter, and the data output pin is connected to the second counter.

Optionally, the first counter includes a first counting chip, a second counting chip, and a first not gate, clock input ends of the first counting chip and the second counting chip are respectively connected to the frequency synthesizer, a carry output pin of the first counting chip is connected to a count control pin of the second counting chip, a lowest data output pin of the second counting chip is connected to the gate, and the first not gate is connected in series between a second lowest data output pin and a clear pin of the second counting chip.

Optionally, the frequency synthesizer includes a second and gate, a clock source and a frequency synthesis chip, a first input end of the second and gate is connected to the clock source, a second input end of the second and gate is connected to the second counter to receive the enable signal, and an output end of the second and gate is connected to the frequency synthesis chip.

Optionally, the second counter includes a not gate array, a first and gate, a third counting chip, a second not gate, and a third not gate, where the not gate array is connected between a data input pin of the third counting chip and the gate, the first and gate is connected between a data output pin of the third counting chip and an input end of the second not gate, an output end of the first and gate is connected to the frequency synthesizer to output the enable signal, an output end of the second not gate is connected to a count control pin of the third counting chip, an output end of the third not gate is connected to a count control pin of the third counting chip, and an input end of the third not gate is connected to the frequency synthesizer to receive the waveform completion signal.

Optionally, the second counter further includes a first nand gate, a first input end of the first nand gate is connected to an output end of the second not gate, a second input end of the first nand gate is connected to an output end of the third not gate, and an output end of the first nand gate is connected to the frequency synthesizer to output the enable signal.

In another aspect, the present application also provides a radar, in particular, including an antenna and a radar waveform generation circuit as described above, which are connected to each other.

Optionally, the radar further comprises a main controller for controlling the frequency modulated continuous wave, and the main controller is connected with a frequency synthesizer in the radar waveform generating circuit.

On the other hand, the present application further provides a radar waveform generation method, specifically, the radar waveform generation method includes:

acquiring a first enabling signal;

generating a set waveform of frequency-modulated continuous waves according to the first enabling signal, and generating a waveform completion signal when the set waveform is completed;

acquiring time gap data according to the waveform completion signal;

counting time according to the time gap data, and generating the enabling signal for the second time when the time counting is finished;

the step of acquiring time gap data from the waveform completion signal comprises:

and counting the number of the waveform completion signals, and generating a gating conversion signal when the number counting is completed so as to replace a first time interval to a second time interval from gating as the time interval data.

In another aspect, the present application further provides a computer storage medium, in particular, a computer program stored on the computer storage medium, which when executed by a processor, can implement the radar waveform generation method as described above.

The radar waveform generation circuit, the radar waveform generation method, the radar and the computer storage medium can set different Doppler resolution between two groups of waveforms by setting different time gaps between the waveforms in two sections, and can effectively improve a measurement interval and further refine the minimum measurement resolution by further processing echoes.

Drawings

Fig. 1 is a flowchart of a radar waveform design method according to an embodiment of the present application.

Fig. 2 is a block diagram of a radar waveform design apparatus according to an embodiment of the present application.

Fig. 3 is a schematic diagram of a radar wave according to an embodiment of the present application.

Fig. 4 is a flowchart of a radar wave processing method according to an embodiment of the present application.

Fig. 5 is a block diagram of a radar wave processing apparatus according to an embodiment of the present application.

Fig. 6 is a flowchart of a subdivided doppler velocity processing method according to an embodiment of the present application.

Fig. 7 is a schematic diagram illustrating a sampling sequence of a speech observation vector according to an embodiment of the present application.

Fig. 8 is a block diagram of a radar waveform generation circuit according to an embodiment of the present application.

Fig. 9 is a circuit connection diagram of a gate according to an embodiment of the present application.

Fig. 10 is a circuit diagram of a first counter according to an embodiment of the present application.

Fig. 11 is a circuit diagram of a frequency synthesizer according to an embodiment of the present application.

FIG. 12 is a block diagram of a radar according to an embodiment of the present application.

Fig. 13 is a flowchart of a radar waveform generation method according to an embodiment of the present application.

Detailed Description

The following detailed description of embodiments of the present application will be described in conjunction with the accompanying drawings and examples. The following examples are intended to illustrate the present application but are not intended to limit the scope of the present application.

In one aspect of the present application, a radar waveform design method is provided. Fig. 1 is a flowchart of a radar waveform design method according to an embodiment of the present application.

As shown in fig. 1, in an embodiment, a radar waveform design method includes:

s110: setting a first center wavelength, a first waveform duration, a first waveform number, a first waveform gap, a second center wavelength, a second waveform duration, a second waveform number and a second waveform gap.

The specific multiple waveforms with different Doppler resolution are realized by presetting the waveform parameters. The waveforms with different doppler resolutions can be realized by setting different center wavelengths, different waveform durations, different numbers of waveforms, and different waveform gaps, for example, by setting different waveform slopes or different radio frequency bandwidths to obtain different total observation durations to realize different doppler resolutions.

S120: and outputting the frequency-modulated continuous waves of the first sound part in the radar wave beams with the first waveform number according to the first center wavelength and the first waveform duration, wherein a first waveform gap is formed between adjacent frequency-modulated continuous wave waveforms of the first sound part.

A first tone part of the radar beam is output by the parameter of the first waveform.

S130: and outputting the frequency-modulated continuous waves of the second sound part in the radar wave beams with the second waveform number according to the second center wavelength and the second waveform duration, wherein a second waveform gap is formed between adjacent frequency-modulated continuous wave waveforms of the second sound part.

And outputting a second sound part of the radar beam through the parameters of the second waveform. Wherein, under different waveform parameters, the second sound part can have different Doppler resolution from the first sound part.

The arrangement of the plurality of sound parts with different Doppler resolution in the radar beam facilitates not only upward expansion of the measurement range of the Doppler velocity but also downward subdivision of the measurement resolution of the Doppler velocity in the echo processing.

In one embodiment, in performing S110: the steps of setting the first center wavelength, the first waveform duration, the first waveform number, the first waveform interval, the second center wavelength, the second waveform duration, the second waveform number and the second waveform interval comprise:

set according to the following formula:

[λ1*N2*(t21+t22)]/[λ2*N1*(t11+t12)]≠A

wherein λ 1 is a first center wavelength, N1 is a first waveform number, t11 is a first waveform duration, t12 is a first waveform gap, λ 2 is a second center wavelength, N2 is a second waveform number, t21 is a second waveform duration, t22 is a second waveform gap, and a is any positive integer.

In order to extend the measurement range of the doppler velocity upward, the ratio of the doppler resolution of the first sound component to the doppler resolution of the second sound component is controlled not to be an integer when setting the waveform parameters.

set according to the following formula:

[λ1*N2*(t21+t22)]/[λ2*N1*(t11+t12)]＝(M+1)/M

wherein λ 1 is a first center wavelength, N1 is a first waveform number, t11 is a first waveform duration, t12 is a first waveform gap, λ 2 is a second center wavelength, N2 is a second waveform number, t21 is a second waveform duration, t22 is a second waveform gap, and M is a positive integer greater than 1.

When the waveform parameters are set, the ratio of the Doppler resolution of the first sound part to the Doppler resolution of the second sound part is controlled to be the ratio of two adjacent integers, which is beneficial to reducing the calculation amount of a radar system.

In one embodiment, in performing S130: outputting the frequency-modulated continuous waves of the second tone part in the radar wave beams with the second waveform number according to the second center wavelength and the second waveform duration, wherein the step of spacing a second waveform gap between adjacent frequency-modulated continuous wave waveforms of the second tone part comprises the following steps:

in one beam, the output of the second tone part is started immediately after the output of the first tone part is completed.

The two sound parts are close to each other in time as much as possible, so that blank zones can be avoided in the measuring range, and particularly the analysis blind zone of the secondary Doppler Fourier transform is reduced.

On the other hand, this application still provides a radar waveform design device. Fig. 2 is a block diagram of a radar waveform design apparatus according to an embodiment of the present application.

As shown in fig. 2, in an embodiment, the radar waveform design apparatus includes a memory 11, a processing module 12 and a frequency synthesis module 13 connected in sequence, wherein:

the memory 11 is configured to store and send the first center wavelength, the first waveform duration, the first waveform number, the first waveform gap, the second center wavelength, the second waveform duration, the second waveform number, and the second waveform gap to the processing module 12;

the processing module 12 is configured to control the frequency synthesis module 13 to output the frequency modulated continuous waves of the first tone part of the radar beams with the first waveform number according to the first center wavelength and the first waveform duration, where a first waveform gap is formed between adjacent frequency modulated continuous wave waveforms of the first tone part; the processing module 12 controls the frequency synthesis module 13 to output the frequency modulated continuous waves of the second tone part in the radar waveforms with the second waveform number according to the second center wavelength and the second waveform duration, and a second waveform gap is formed between adjacent frequency modulated continuous wave waveforms of the second tone part.

And outputting the first sound part and the second sound part of the radar beam through the parameters of the preset waveform. Wherein the second sound part has a different Doppler resolution from the first sound part.

In one embodiment, the memory 11 performs the setting storage according to the following formula:

[λ1*N2*(t21+t22)]/[λ2*N1*(t11+t12)]≠A

In one embodiment, the number of the first waveforms output by the frequency synthesis module 13 is the same as the number of the second waveforms.

The number of the waveforms of the two sound parts is the same, and the calculation amount of the radar system is favorably reduced.

In one embodiment, the waveform of the frequency modulated continuous wave output by the frequency synthesis module 13 is selected from at least one of a triangular wave, a sawtooth wave, a step wave, and a sine wave.

In the radar application practice, various different waveforms can be selected and used according to the actual application environment so as to meet the requirements of radar distance measurement and speed measurement.

In one embodiment, the frequency synthesis module 13 outputs a plurality of scan frames in each radar scan period, and outputs a plurality of radar beams in different directions in each scan frame.

By transmitting the beams in a plurality of different directions, echo signals in various different directions can be acquired, and a detection target can be identified more accurately.

Referring to fig. 3, in an embodiment, a waveform transmitted by a radar is named as "dui-Duo" (Duo), and is a multi-frame multi-beam multi-level complex Frequency Modulated Continuous Wave (FMCW) waveform, which is basically characterized as follows:

each radar detection Cycle (Cycle) is equally divided in time into a plurality of scan frames (Frame #1, Frame #2 … …).

Beams (Duo Beam #1, Duo Beam #2, Duo Beam #3 … …) with different directions, different distance resolutions and different doppler resolutions are transmitted in each scanning frame.

Each beam is a Group of duet (Duo) ramps (Chirp) comprising at least two groups of high-resolution (H-tone) and low-resolution (L-tone) ramps (Chirp Group) which have slightly different doppler resolutions, are always present in close proximity, and are combined at appropriate times to the final expansion result, referred to in this application as the two "tones" of the duet.

The two sound parts of H and L are closely adjacent to each other in time (refer to a time-frequency diagram in a figure) as far as possible, namely, after the H sound part is finished, the L sound part is immediately started to be transmitted, the number of the slopes of the two groups of sound parts is the same, and each slope is half of the total number of the slopes of the located wave beam.

In another aspect, the present application further provides a computer storage medium.

The computer storage medium has stored thereon a computer program which, when executed by a processor, can implement the radar waveform design method as described above.

According to the radar waveform design method, the radar waveform design device and the computer storage medium, the plurality of sound parts with different Doppler resolution powers are arranged in the radar beam, so that the dynamic measurement range of a product can be effectively enlarged, and the maximum measurement value and the minimum measurement resolution power are improved.

In another aspect, the present application also provides a radar wave processing method. Fig. 4 is a flowchart of a radar wave processing method according to an embodiment of the present application.

As shown in fig. 4, the radar wave processing method includes:

s210: fourier transform of baseband signals is performed on echoes of a plurality of sound parts in a radar beam respectively to acquire a plurality of corresponding frequency domain data.

In engineering practice, the echo of the time domain signal can be subjected to fast Fourier transform and discrete Fourier transform, so that frequency domain spectrum data can be obtained, and a detection target can be identified. The fourier transform may include a distance fourier transform and a doppler fourier transform.

S220: and respectively acquiring a plurality of corresponding original signal-to-noise ratio matrixes according to the plurality of frequency domain data.

The frequency domain data of the plurality of sound parts can respectively obtain distance-Doppler matrix data of the plurality of sound parts as an original signal-to-noise ratio matrix.

S230: and merging and accumulating the plurality of original signal-to-noise ratio matrixes to the extended signal-to-noise ratio matrix.

For a plurality of original snr matrices formed by a plurality of voice component echoes with different doppler resolutions, the snr matrices need to be further integrated and extended to accumulate snr data.

S240: and screening a target point according to the extended signal-to-noise ratio matrix.

And a target point can be screened in a larger Doppler measurement range through an extended signal-to-noise ratio matrix formed by the accumulated signal-to-noise ratio data.

By processing the echo including a plurality of sound parts and further combining and accumulating the echo to an extended signal-to-noise ratio matrix through a mapping relation on the basis of an original signal-to-noise ratio matrix, a Doppler measurement interval can be extended upwards, and the measurement range is effectively expanded.

In one embodiment, in performing S210: the step of performing fourier transform of the baseband signal on the echoes of the plurality of tone parts in the radar beam to obtain a plurality of corresponding frequency domain data includes:

the baseband signals of the echoes of the plurality of sound parts are subjected to Doppler Fourier transform and/or distance Fourier transform.

In one embodiment, in performing S220: the step of obtaining a plurality of corresponding original signal-to-noise ratio matrixes according to the plurality of frequency domain data respectively comprises the following steps:

obtaining logarithmic signal intensities at all coordinate positions in an original signal-to-noise ratio matrix to generate a screening signal-to-noise ratio threshold;

carrying out distribution statistics on the logarithmic signal intensity of the peripheral coordinate position of the first coordinate position so as to obtain the logarithmic noise intensity of the first coordinate position according to a preset noise algorithm;

and subtracting the value of the screening signal-to-noise ratio threshold value from the difference between the logarithmic signal intensity and the logarithmic noise intensity of the first coordinate position to serve as the original signal-to-noise ratio of the first coordinate position.

constant false alarm calculation is carried out on the frequency domain data of the first voice part echo to obtain a first original signal-to-noise ratio matrix;

and performing constant false alarm calculation on the frequency domain data of the second tone part echo to obtain a second original signal-to-noise ratio matrix.

In one embodiment, in performing S230: the step of merging and accumulating the plurality of original signal-to-noise ratio matrixes to the extended signal-to-noise ratio matrix comprises the following steps:

respectively mapping the coordinates of the extended signal-to-noise ratio matrix to a plurality of original signal-to-noise ratio matrices;

interpolating the mapping coordinate positions in the original signal-to-noise ratio matrixes to obtain an interpolation result;

selecting an original signal-to-noise ratio minimum value in an interpolation result of mapping coordinate positions in a plurality of original signal-to-noise ratio matrixes;

and taking the sum of the signal-to-noise ratio accumulated value of the corresponding coordinate position in the extended signal-to-noise ratio matrix and the original signal-to-noise ratio minimum value as the combined signal-to-noise ratio accumulated value.

In an embodiment, the step of performing the mapping of the coordinates of the extended snr matrix to the plurality of original snr matrices includes:

acquiring the sequence number of a scanning frame in which an original signal-to-noise ratio matrix is positioned in a measurement period, and acquiring mapping coordinate positions mapped to a plurality of original signal-to-noise ratio matrices according to the following formula:

[i,j]＝[(Ei*ERres+k*Tf*(Ej–Edz)*EDres)/Rres,((Ej–Edz)*EDres)/Dres)％NDFFT]

wherein, i is the value of the coordinate horizontal axis of the mapped original signal-to-noise ratio matrix; j is a coordinate longitudinal axis value of the mapped original signal-to-noise ratio matrix, Ei is a coordinate transverse axis value of the expanded signal-to-noise ratio matrix, Ej is a coordinate longitudinal axis value of the expanded signal-to-noise ratio matrix, ERres is a distance resolution of the expanded signal-to-noise ratio matrix, EDres is a doppler resolution of the expanded signal-to-noise ratio matrix, EDz is a doppler zero point of the expanded signal-to-noise ratio matrix, k is a serial number of a scanning frame in a measurement period in which the original signal-to-noise ratio matrix is located, Tf is a period duration of the scanning frame, Rres is a distance resolution of a sound part in which the original signal-to-noise ratio matrix is located, Dres is a doppler resolution of a sound part in which the original signal-to-noise.

In one embodiment, the step of performing interpolation on the mapping coordinate positions in the plurality of original signal-to-noise ratio matrixes to obtain an interpolation result includes;

obtaining adjacent integer coordinates [ il, jb ] of which the horizontal axis coordinate is reduced by no more than one coordinate position and the vertical axis coordinate is reduced by no more than one coordinate position in the mapping coordinates [ i, j ], and interpolating an original signal-to-noise ratio of the mapping coordinate [ i, j ] position according to the following formula:

So[i，j]＝So[il，jb]*(il+1-i)*(jb+1-j)+

So[il，jb+1]*(il+1-i)*(j-jb)+

So[il+1，jb]*(i-il)*(jb+1-j)+

So[il+1，jb+1]*(i-il)*(j–jb)

and So is the original signal-to-noise ratio value of the corresponding coordinate position in the original signal-to-noise ratio matrix.

On the other hand, the application also provides a radar wave processing device. Fig. 5 is a block diagram of a radar wave processing apparatus according to an embodiment of the present application.

As shown in fig. 5, in one embodiment, the radar wave processing apparatus includes a radar receiver 21 and a processor 22 connected to each other.

The radar receiver 21 is used to receive and transmit radar echoes to the processor 22.

The processor 22 is used for executing the radar wave processing method as described above.

For example, in one embodiment, the radar-generic baseband signal processing unit provided by the processor 22 includes a range Fast Fourier Transform (FFT) unit, a doppler fast fourier transform (doppler FFT) unit, a constant false alarm detection (CFAR) unit, and so on. The baseband signal processing unit composed of the above subunits should perform the H and L two-tone parts once in each beam, and output an original constant false alarm detection signal-to-noise ratio matrix through the constant false alarm detection subunit. Preferably, the above units can all be implemented by using pure hardware, and the whole processing process does not need software intervention. Preferably, although the baseband signal processing means processes separately for both H and L tones, sufficient resource savings should be made to consider time-division multiplexing using the same device instance, rather than two duplicate devices performing in parallel.

The processor 22 is provided with an extended range-doppler (RD) signal-to-noise ratio storage unit, which is a two-dimensional matrix, and each cell of the matrix stores a signal-to-noise ratio of a range-doppler coordinate (range value, doppler value) corresponding to the cell, where the signal-to-noise ratio represents the probability that a real reflector target exists at the range-doppler coordinate. The number of Doppler layers stored in the signal-to-noise ratio storage unit is W times of the number of Doppler layers (usually the same as the number of Doppler fast Fourier transform points) of an original constant false alarm detection signal-to-noise ratio matrix in the baseband signal processing device, wherein W is a positive number not less than 1, so that the measurement range of Doppler is expanded upwards (higher measurement value) by W times.

The processor 22 is provided with an extended range-doppler snr computing unit, which increases or decreases the snr cumulative value in the extended range-doppler snr storage unit according to the baseband signal processing result (here, mainly referred to as the original constant false alarm detection snr) from each of the two repetitions, and the specific computing method is described in detail in the following description of the signal processing method. The calculation operation should be performed once when the processing of the baseband signal of each tone part of each beam of each period in the three-level waveform of the present application is completed.

The processor 22 is provided with an extended range-doppler signal-to-noise ratio filtering unit, that is, an extended constant false alarm detection unit, and when each radar detection cycle starts, a range-doppler coordinate corresponding to an extended range-doppler cell until the accumulated signal-to-noise ratio is higher than the target determination signal-to-noise ratio threshold value at this time is screened out on the extended range-doppler signal-to-noise ratio matrix by a constant false alarm detection method and used as a radar target point cloud of the current detection cycle.

The detected target points are identified by the processor 22 through a final screening of the radar target point cloud.

In an embodiment, each measurement period is equally divided into a plurality of measurement frames, each measurement frame performs a complete baseband signal processing, and each measurement frame needs to update a global extended range-doppler signal-to-noise ratio matrix according to a baseband signal processing result.

Before the baseband signal processing of the first measurement frame of each period starts, performing once extended constant false alarm detection, filtering out range-doppler coordinates with signal-to-noise ratio higher than a set threshold value from the currently accumulated extended range-doppler signal-to-noise ratio matrix, outputting the range-doppler coordinates as target points of the measurement period, and further performing subsequent operations, such as quadratic doppler Discrete Fourier Transform (DFT) and azimuth solution.

In the last measurement frame of each period, there is still a chance to further obtain a subdivided doppler measurement result with higher doppler resolution based on a group of tone part observation vector sequences generated by each tone part of all frames of the measurement period, the measurement result replaces the doppler value of the target extended distance-doppler coordinate obtained from extended constant false alarm detection in the first measurement frame of the period, and the target is split into a plurality of pieces according to different values of subdivided doppler, thereby realizing finer doppler resolution.

In one embodiment, the stages of the fast fourier transform operation may add a preceding window function, such as a hanning window.

In one embodiment, the constant false alarm detection algorithm may add a preceding Digital Beam Forming (DBF) algorithm, so as to virtually focus the beam in a certain direction, thereby improving the signal-to-noise ratio in the focusing direction.

In one embodiment, in order to obtain the range-doppler snr matrix with multiple extended intervals to achieve higher doppler measurement values, it is necessary to "merge" the original constant false alarm detection snr matrices output by two constant false alarm detections to each cell of the extended range-doppler snr matrix after the constant false alarm detections of the H and L two tones of each measurement frame are completed.

Based on the value of the extended distance-Doppler signal-to-noise ratio matrix, before the baseband signal processing of the first measurement frame of each period starts, one-time 'extended constant false alarm detection' is executed on the currently accumulated extended distance-Doppler signal-to-noise ratio matrix, namely, distance-Doppler coordinates with the signal-to-noise ratio higher than the set extended constant false alarm detection signal-to-noise ratio threshold are screened out, and the distance-Doppler coordinates are output as target point clouds of the measurement period.

In one embodiment, the "merging and accumulating" process to the extended range-doppler snr matrix is generally to accumulate the accumulated value of the last frame of the extended range-doppler snr by taking the lower of the raw constant false alarm detection snr interpolation results of the "corresponding positions" (detailed below) of the H-tone and the L-tone:

the current frame integrated value of extended distance-Doppler signal-to-noise ratio [ Ei, Ej ] + is the previous frame co-location integrated value +

MIN{

The original constant false alarm detection signal-to-noise ratio [ H map (Ei, Ej) ],

original constant false alarm detection signal-to-noise ratio [ L mapping (Ei, Ej) ]

}

For example, when the constant false alarm detection result of the j frame of the ith measurement period is being processed, the interpolation result of the original constant false alarm detection signal-to-noise ratio of the H tone part corresponding to a certain extended distance-doppler cell is 123, the interpolation result of the original constant false alarm detection signal-to-noise ratio of the L tone part is-456, and the extended distance-doppler signal-to-noise ratio that has been accumulated for this cell at this time is 789, then after this update, the new accumulated signal-to-noise ratio value of this cell is: 789+ min (123, -456) 333 ═ 333

In one embodiment, the method for calculating the original constant false alarm rate detection signal-to-noise ratio matrix is as follows: assuming that the logarithmic signal intensity of a certain cell is S, the logarithmic noise intensity obtained by statistics according to the intensity distribution of the peripheral cells is No, and the signal-to-noise ratio threshold used by the target generated by constant false alarm detection screening is T in the execution process of the constant false alarm detection algorithm, the original constant false alarm detection signal-to-noise ratio at the position is recorded as:

original constant false alarm detection signal-to-noise ratio (S-No-T)

That is, for a cell that can just be screened as the original constant false alarm detection target, the obtained original signal-to-noise ratio value is just 0; the signal-to-noise ratio of higher quality targets is higher than 0; while the signal-to-noise ratio of the cells that are not enough to be the original target is below 0.

The method for calculating the noise of each cell for constant false alarm detection belongs to the known field, and is not described in detail in the application. In an embodiment, in order to improve the operation efficiency of the engineering implementation, a constant false alarm detection noise calculation method supported by a radar hardware platform (e.g., an SoC chip), such as a simple CFAR-CA (including CFAR-CASO or CFAR-CAGO), or a CFAR-OS with better effect, should be considered. Generally speaking, in most implementations of constant false alarm detection algorithms, noise at a cell is obtained by counting (averaging, accumulating probability values, etc.) signal values of a plurality of neighboring cells around the cell, which generally represents the background signal strength, i.e. noise strength, in the neighborhood of the cell.

Due to the difference of the doppler resolution between H and L and the influence of the moving speed of the target, the "corresponding position" of the extended range-doppler coordinate (Ei, Ej) in the coordinate mapping method is often a non-integer original range-doppler coordinate, so the "original constant false alarm detection signal-to-noise ratio of the corresponding position" is usually interpolated from the non-integer original range-doppler coordinate mapping in the original constant false alarm detection signal-to-noise ratio matrix.

In one embodiment, the snr interpolation method is as follows:

assuming that the cell coordinates of a certain extended range-doppler signal-to-noise ratio matrix are (Ei, Ej), the H coordinates after mapping the coordinates to the original constant false alarm detection signal-to-noise ratio matrix corresponding to the H part are (ih, jh), there are 4 integer range-doppler coordinates closely adjacent to the non-integer coordinates, and it is noted that the leftmost lower corner (R and D are both minimum) in the 4 coordinates is [ ihl, jhb ], then the original constant false alarm detection signal-to-noise ratio of the H part at (ih, jh) obtained by interpolation based on the original signal-to-noise ratios of the 4 neighborhood cells is:

original constant false alarm rate detection signal-to-noise ratio [ ih, jh ] ═

Original constant false alarm detection signal-to-noise ratio [ ihl, jhb ] (ihl +1-ih) (jhb +1-jh) +

Original constant false alarm detection signal-to-noise ratio [ ihl, jhb +1] (ihl +1-ih) (jh-jhb) +

Original constant false alarm detection signal-to-noise ratio [ ihl +1, jhb ] (ih-ihl) (jhb +1-jh) +

Original constant false alarm detection signal-to-noise ratio [ ihl +1, jhb +1] (ih-ihl) (jh-jhb)

Similarly, the coordinate mapping method of the L-tone part is consistent with that of the H-tone part, and is not described herein again.

For example, if a certain extended range-doppler snr matrix cell (23, 4) is mapped to the L-tone part with the coordinates of (23.1, 3.6), the original constant false alarm detection snr at the L-tone part can be obtained by the above interpolation method:

l-tone original constant false alarm detection signal-to-noise ratio [23.1,3.6] ═

Original constant false alarm detection signal-to-noise ratio [23,3] + 0.9 + 0.4+ of L tone part

Original constant false alarm detection signal-to-noise ratio [23,4] + 0.9 + 0.6+ of L tone part

Original constant false alarm detection signal-to-noise ratio [24,3] + 0.1 + 0.4+ of L tone part

Original constant false alarm detection signal-to-noise ratio [24,4 ]. 0.1. 0.6 of L sound part

In one embodiment, the mapping strategy from the "extended range-doppler signal-to-noise ratio matrix" to the "original constant false alarm detection signal-to-noise ratio matrix" is as follows:

the doppler interval represented by constant false alarm detection needs to be spread in multiple continuous non-overlapping ways, which is one of the core operations of the present application and is intended to extend the measurement range of doppler. For example, the original doppler interval represented by the constant false alarm detection has 32 layers (the corresponding doppler fast fourier transform has 32 output spectral lines), and 8-fold expansion is performed, so that the expanded range-doppler matrix can reach 256 layers. Considering the aliasing characteristic of the doppler fast fourier transform, the doppler spectral line coordinate j in the extended range-doppler signal-to-noise ratio matrix after expansion needs to be periodically and circularly mapped to the original effective expression range of the constant false alarm detection. In the above example, a plurality of extended doppler coordinates, such as an extended doppler coordinate Ej of 2, an extended doppler coordinate Ej of 34, an extended doppler coordinate Ej of 66, and an extended doppler coordinate Ej of 98, are mapped to the same constant false alarm detection original doppler coordinate j of 2.

Considering that the target is moving, the mapping relationship should be kept synchronous with the moving speed of the target at the beginning of a plurality of different frames of each measurement period, so that the original constant false alarm detection signal-to-noise ratios between the plurality of different frames can be reasonably superposed at the same distance position, otherwise, the staggered superposition cannot accumulate the signal-to-noise ratio at the real target to a higher value.

Based on the above strategy, the following coordinate mapping method can be obtained:

suppose that in the 0 th frame of a certain measurement period, the cell coordinates of the extended range-doppler signal-to-noise ratio matrix corresponding to a certain real target are [ Ei, Ej ], the range resolution of the entire extended range-doppler signal-to-noise ratio matrix is ERres, the doppler resolution is EDres, and the doppler zero point (the extended range-doppler cell doppler coordinate representing zero velocity) is EDz.

Meanwhile, assuming that the number of doppler layers (the number of spectral lines of doppler fast fourier transform) of instantaneous constant false alarm detection of a certain tone part (H or L) is NDFFT, the distance resolution is Rres, the doppler resolution is Dres, and the time interval (frame duration) between two adjacent measurement frames is Tf, then:

in the 0 th frame of each measurement period, the physical coordinates corresponding to the extended coordinates [ Ei, Ej ] are: [ Ei ERres, (Ej-Edz) EDres ]

At the k-th frame of each measurement period, the distance values in the physical coordinates change due to the object motion (a uniform radial motion can be assumed for a short time between frames):

Ei*ERres+k*Tf*(Ej–Edz)*EDres

in the kth frame, according to the distance resolution and the doppler resolution of the original constant false alarm rate detection signal-to-noise ratio matrix, the original cell coordinate [ i, j ] is:

[i,j]＝[

(Ei*ERres+k*Tf*(Ej–Edz)*EDres)/Rres,

((Ej–Edz)*EDres)/Dres)％NDFFT

]

in the above equation, if the original distance coordinate after mapping exceeds the effective distance measurement range, it indicates that the target has moved outside the view field in the k-th frame, and it should be regarded as an invalid mapping, and the corresponding instantaneous signal-to-noise ratio should be set to an extremely low value, so as to ensure that such a target does not obtain a sufficient value in the final extended distance-doppler signal-to-noise ratio matrix to generate an erroneous target.

The operation of taking the remainder of the original mapped Doppler coordinates to the number of spectral lines in the above equation is designed according to the aliasing characteristic of the Doppler fast Fourier transform. Its aliasing properties determine where the k-th original spectral line that is out of the measurement range will be aliased to the k% NDFFT original spectral line. For example, when NDFFT is 16, the 18 th, 34 th, 66 th, 98 th original spectral lines that cannot be directly measured will be aliased to the 2 nd original spectral line, and the 2 nd original spectral line is measurable.

The Doppler resolution of each sound part is controlled not to be integral multiple relation. The key of the upward extension Doppler measurement interval is that two slightly different Doppler resolution forces are achieved through two H and L tone parts, two slightly different Doppler aliasing ranges are achieved, and then two different original constant false alarm detection measurable Doppler coordinates are obtained when the same extension Doppler coordinate Ej is mapped to the H and the L: if the target is actually present, both of the original constant false alarm detections at the doppler coordinate should exhibit a sufficiently high measured signal-to-noise ratio, otherwise at least one constant false alarm detection would correspond to a relatively low measured signal-to-noise ratio. By taking the lower of the two original constant false alarm detection signal-to-noise ratios to participate in the accumulation of the extended range-doppler signal-to-noise ratio matrix, such targets which do not exist can be eliminated at the stage of finally executing extended constant false alarm detection (screening targets according to the extended range-doppler signal-to-noise ratio).

In the above embodiment, the doppler measurement range is extended upward by matching the H + L duet waveform with the multi-layer extended range-doppler signal-to-noise ratio matrix to resist doppler aliasing.

Note that a plurality of sound portions in one beam are emitted in close proximity, and the highest (fine) resolution sound portion is not limited to the front or the rear.

In other embodiments, the design of upgrading the two H and L sections to multiple sections can improve the extended snr with increased time overhead and transmission power consumption, which is essentially the same as the method of doppler extension for two sections listed in this application for illustrating the design principle, and still falls within the protection scope of this application.

In one embodiment, the present application also provides a computer storage medium.

The computer storage medium has stored thereon a computer program which, when executed by a processor, can implement the radar wave processing method as described above.

According to the radar wave processing method, the radar wave processing device and the computer storage medium, the plurality of sound parts with different Doppler resolutions are arranged in the radar beam, so that the dynamic measurement range of a product can be effectively enlarged, and the maximum measurement value can be effectively enlarged.

On the other hand, the application also provides a method for processing the subdivided Doppler velocity. Fig. 6 is a flowchart of a subdivided doppler velocity processing method according to an embodiment of the present application.

As shown in fig. 6, in an embodiment, the method for processing the subdivided doppler velocity includes:

s310: and respectively carrying out Fourier transformation on the baseband signals of the echoes of the plurality of sound parts in the radar beam comprising the first beam so as to screen target points, wherein the Fourier transformation comprises first Doppler Fourier transformation.

And performing frequency domain conversion on echoes of a plurality of sound parts of the radar, performing Doppler Fourier transform for the first time, and screening out cell coordinates which possibly contain a plurality of targets with similar speeds.

S320: and acquiring a plurality of sound part observation vectors of the target point according to the first Doppler Fourier transform.

And calculating and storing the instantaneous observation vector of each part of the coordinate where each target point is located.

S330: and extracting a plurality of voice part observation vectors of the first beam in each scanning frame in at least one measuring period, and forming a voice part observation vector sequence according to the time domain sequence.

A beam includes a plurality of tones, each beam being distributed within each scan frame. By extracting a plurality of tone observation vectors of the same beam in each scanning frame, a virtual sampling sequence of the time domain can be sequentially constructed.

S340: and performing secondary Doppler Fourier transform on the sound part observation vector sequence to acquire amplitude distribution data.

And performing secondary Doppler Fourier transform on the virtual sampling sequence to obtain the amplitude distribution condition of the frequency domain.

S350: and screening the subdivision targets in the target points according to the amplitude distribution data.

And through a proper screening algorithm, subdivision targets with similar speeds in the same target point can be screened out.

By calculating the secondary Doppler Fourier transform, a subdivided Doppler spectrum is obtained, and a plurality of targets which are close in speed and cannot be resolved by the primary Fourier transform and are positioned in the same output spectral line of the primary Doppler Fourier transform can be resolved.

In one embodiment, in performing S310: performing Fourier transform of baseband signals on echoes of a plurality of sound parts in a radar beam including a first beam to screen a target point, wherein the step of performing Fourier transform including first Doppler Fourier transform comprises the following steps:

acquiring an extended signal-to-noise ratio matrix according to echoes of a plurality of sound parts;

before the first scanning frame of the measuring period, constant false alarm calculation is carried out on the extended signal-to-noise ratio matrix so as to screen out a target point.

As in the above-described embodiment, the target point is screened by expanding the snr matrix, so that the measurement range of the doppler velocity can be expanded, and the target point is screened for the subdivided quadratic doppler fourier transform.

In one embodiment, in performing S320: the step of acquiring a plurality of acoustic observation vectors of a target point based on the first doppler fourier transform includes:

mapping the target point to an original signal-to-noise ratio matrix to obtain an original Doppler coordinate of the current voice part; acquiring an original complex vector of the initial Doppler Fourier transform of an original Doppler coordinate; acquiring an estimated azimuth angle of a target point; and according to the estimated azimuth angle, performing antenna vector combination with reserved phase on the original complex vector to obtain a voice part observation vector of the current voice part.

In one embodiment, the step of obtaining the original complex phasor of the first doppler fourier transform at the original doppler coordinate comprises:

when the original Doppler coordinate is a non-integer coordinate, acquiring a first complex vector of a smaller integer coordinate and a second complex vector of a larger integer coordinate in two nearest non-integer coordinates;

and carrying out interpolation combination on the first complex vector and the second complex vector to obtain an original complex vector with non-integer coordinates.

For example, in one embodiment, if a second doppler discrete fourier transform is performed on a target from a cell of the extended range-doppler signal-to-noise ratio matrix (Ei, Ej) to subdivide the doppler value of the target and generate a plurality of subdivided doppler targets (if any), a sequence of "tone observation vectors" is first constructed, and the "tone observation vector" from the first doppler fast fourier transform result for each tone in the sequence can be generated by:

first, after the first doppler fast fourier transform of each tone part is completed, according to the velocity of the extended range-doppler target and the doppler resolution of the current tone part, (Ei, Ej) extended range-doppler coordinates are mapped to the original coordinates of the initial doppler fast fourier transform original complex vector matrix of the tone part, as described above:

{ H | L mapping (Ei) ((Ej-Edz). times. EDres)/Dres)% NDFFT }

In order to simplify the calculation in the engineering implementation, the distance resolution of the extended range-doppler signal-to-noise ratio matrix is often the same as the distance resolution of each tone part, so that the original distance coordinate i after mapping is completely consistent with the Ei before mapping. If the two resolutions are different, then the floating point number translation (round) operation is generally taken to approximate to integer coordinates.

For the doppler dimension coordinates, as mentioned above, the original doppler coordinates mapped to the current part are:

Dj＝((Ej–Edz)*EDres)/Dres)％NDFFT

since the above coordinates are not integer values in general, and the initial doppler fft original mv matrix can only take values at integer cells, two nearest integer coordinates (i.e. floor (Dj) and floor (Dj) +1) of the non-integer coordinates can be taken, and then at these two nearest integer coordinates, two sets of mv formed by receiving antennas corresponding to the non-integer coordinates roughly are taken from the original mv matrix, one set with doppler slightly higher than Dj is recorded as RXt, and one set with doppler slightly lower than Dj is recorded as RXb, for a radar with NRX receiving antennas, there are:

RXt-first doppler fast fourier transform [ i, floor (dj)) +1] { RXt (1), RXt (2), …, rxt (nrx)) }

RXb first doppler fast fourier transform [ i, floor (dj)) ] { RXb (1), RXb (2), …, rxb (nrx)) }

Then, according to the actual non-integer value of j, the original complex vectors of corresponding receiving antennas of RXt and RXb are combined by interpolation to obtain the original complex vector RX of the first doppler fast fourier transform which is exactly matched with j:

RX ═ interpolation (RXt (1), RXb (1)), interpolation (RXt (2), RXb (2)), …, interpolation (rxt (n), rxb (nrx)) }

Finally, according to the estimated azimuth (DoA) of the target obtained in other units of the radar baseband signal processing device, correlation (coherent) digital beam forming is performed, that is, antenna vector combination operation with reserved phase is performed, RX vectors are combined into a single total vector, and the total vector is used as a "tone observation vector" V of the current tone:

V＝Σ(RX(g)*Twiddle(g))

the rotation vector (Twiddle) of digital beam forming is calculated according to the position of the receiving antenna of the radar device, and when the receiving antenna g with absolute position p (g) (unit is lambda) is used for beam forming in the DoA direction, the rotation vector is as follows:

Twiddle(DoA,g)＝exp(sin(DoA)*2*Π*p(g))

in one embodiment, in the step of performing an interpolation combining of the first complex phasor and the second complex phasor to obtain the original complex phasor at the non-integer coordinate, the exact fft output vector at j may be interpolated from the fft output vectors of the RXt and RXb two nearest neighbor cells.

The original complex phasor rx (g) at the non-integer line position j is obtained by interpolating the first complex phasor rxb (g) and the second complex phasor rxt (g) of the receiving antenna g by:

β＝|RXb(g)|/|RXt(g)|

α＝1/(1+β)

A＝|RXb(g)|*Π*α/sin(Π*α)

P＝∠RXb(g)-Π*α

after obtaining a and P, rx (g) is calculated according to the following trigonometric function:

RX(g)＝A*cos(P)+A*sin(P)*J

wherein g is the serial number of the receiving antenna, and the value is a positive integer greater than 0, rx (g) is the original complex vector, rxb (g) is the first complex vector, rxt (g) is the second complex vector, and J is the unit complex vector.

By adopting a mature engineering algorithm, the method can have sufficient selectivity in the signal processing process, can improve the efficiency of radar design and maintenance, and greatly reduces the cost.

In one embodiment, the step of performing phase-preserving antenna vector combination on the original complex vectors according to the estimated azimuth to obtain the observation vector of the current part includes:

the acoustic observation vector is calculated according to the following formula:

V＝Σ(RX(g)*exp(sin(DoA)*2*Π*p(g)))

wherein, V is a sound observation vector, g is a positive integer serial number of the receiving antenna, the value is a positive integer greater than 1, rx (g) is an original complex vector, DoA is an estimated azimuth angle, pi is a circumferential ratio, and p (g) is an absolute position of the receiving antenna.

In one embodiment, in performing S330: the step of extracting a plurality of voice observation vectors of the first beam in each scanning frame in at least one measuring period and forming a voice observation vector sequence according to the time domain sequence comprises the following steps:

acquiring a sound part observation vector of a first sound part and a sound part observation vector of a second sound part of each scanning frame in at least one measuring period;

and forming a sound part observation vector sequence in the time domain according to the following formula sequence according to the sound part observation vector of the first sound part and the sound part observation vector of the second sound part:

V(Z)＝[VH(1),VL(1),VH(2),VL(2),……,VH(Z),VL(Z)]

wherein Z is a positive integer greater than 1, V (Z) is a sequence of vocal tract observation vectors, VH (Z) is a vocal tract observation vector for the first vocal tract, and VL (Z) is a vocal tract observation vector for the second vocal tract.

No matter the first Doppler fast Fourier transform or the second Doppler discrete Fourier transform is executed, effective sampling points are needed to construct a virtual time domain sampling value sequence, then the Fourier transform is executed to complete the mapping from the time domain to the frequency domain, and the obtained virtual frequency domain data is Doppler distribution information. As known in the art, for the first doppler fast fourier transform, the sequence of sampling points is a sequence of phasors from each ramp in the fast fourier transform on the same range line, and the sequence represents the initial phase of the target on the range line at the start of each ramp.

Referring to fig. 7, a similar manner for constructing a secondary doppler discrete fourier transform virtual sampling sequence is proposed in the present application, that is, after the primary doppler fast fourier transform of each tone part of the duet is completed, digital beam synthesis is performed according to the estimated azimuth where the target point is located, so as to obtain a total tone part observation complex vector that combines all receiving antennas, where the complex vector represents the initial phase of the target at the start time of the current tone part. By storing the 'sound part observation vector' of each sound part in the measurement period, a virtual sampling sequence with non-equal time distribution can be constructed and used for secondary Doppler discrete Fourier transform.

In one embodiment, in performing S340: the step of performing a second doppler fourier transform on the sequence of observation vectors of the vocal part to obtain amplitude distribution data includes:

acquiring observation time corresponding to each value in the voice observation vector sequence according to the voice observation vector sequence;

according to the observation time, obtaining a rotation vector of discrete Fourier transform corresponding to each sample in each spectral line in the secondary Doppler Fourier transform;

and obtaining amplitude distribution data of secondary Doppler Fourier transform according to the rotation vector of the discrete Fourier transform and the sound part observation vector sequence.

For example, referring to fig. 7 and 3 together, each radar detection Cycle (Cycle) is equally divided in time into a plurality of scan frames (Frame #1, Frame #2 … …).

Each beam is a Duo (Duo) ramp (Chirp) Group, which includes at least two sets of ramps (Chirp groups) of high resolution (H-tone part) and low resolution (L-tone part).

Since the "acoustic observation vector" sequence is sampled non-isochronously, it cannot be realized by a fast fourier transform algorithm, and a discrete fourier transform is required. The fast fourier transform with a time complexity of 2NLog2(N) is generally considered faster than the discrete fourier transform with a time complexity of N × N, but in engineering implementations, the hierarchical NDDFT2 of the quadratic doppler discrete fourier transform is generally not too much, and thus the execution time is within an acceptable range. In contrast, discrete fourier transforms have certain advantages over fast fourier transforms. The discrete fourier transform is pipelined and therefore not truncated by a fixed length measurement window. By virtue of this feature, the quadratic doppler discrete fourier transform of the present application can perform target tracking across measurement cycles, calculating the energy distribution of each subdivided doppler tier indefinitely over a longer time dimension, resulting in higher resolution and more stable results. In order to illustrate the method of the present application, the calculation process of the quadratic doppler discrete fourier transform is still constrained within one measurement period:

assuming that Z measurement frames are included in a measurement period duration Tc, where the tone part observation vectors of H and L two tone parts of a beam in the ith (1 ≦ i ≦ Z) measurement frame are VHi and VLi, where the measurement duration of each H tone part is Th, and the L tone part immediately follows, for the following sequence of tone part observation vectors:

V(Z)＝[VH(1),VL(1),VH(2),VL(2),……,VH(Z),VL(Z)]

the observation time corresponding to the ith value is (counted from the start time of the measurement period):

t(i)＝Tc/Z*(i/2)+Th*(i％2)

that is, the minimum time interval between the observation vector sequences of the sound parts is Th (because the L sound parts are the H sound parts in the close accompanying frame), and the periodic time interval that the whole observation vector sequence can cover is Tc (i.e. NDDFT × frame duration Tf), then the doppler measurement capability under the condition can be known according to the definition of the radar doppler resolution:

finest resolution Vres ═ λ/(2 × Tc)

Maximum measurement value Vmax ═ λ/(2 × Th)

With the finest resolution Vres as the output spectral line division value of the second Doppler discrete Fourier transform, the total number of output spectral lines can reach:

NDDFT＝Vmax/Vres＝Tc/Th

in the output spectral lines of a plurality of secondary discrete Fourier transforms of all the spectral lines, the discrete Fourier transform rotation vector corresponding to the ith sample of the kth spectral line is as follows:

Twiddle(k,i)＝exp(-1j*k*t(i)*(2*Π/Tc))

then, according to the mathematical meaning of discrete fourier transform, the amplitude distribution calculation method of the quadratic doppler discrete fourier transform corresponding to the present application can be obtained:

DFT(k)＝|Σ(Twiddle(k,i)*V(i))|

in the output spectral line of the quadratic doppler discrete fourier transform, each local maximum is possibly the result of a sub-doppler. Assuming that the current part observation vector sequence is obtained by interpolating the jth spectral line of the first doppler fast fourier transform, if the amplitude of the kth output spectral line of the second doppler discrete fourier transform is a local maximum, the final subdivided doppler velocity corresponding to the local maximum is:

Vd＝λ/(2*Tc)*j+λ/(2*Th)*k

where λ is the center wavelength. In any case, the doppler resolution is λ/(2 × total observation duration), the first doppler fast fourier transform is not sufficient in resolution, or the second doppler extends the observation duration by one period Tc because the duration of the successive observations is too short (one frame duration Th or Tl). Based on the method provided by the application, after the second doppler discrete fourier transform, the doppler resolution can be successfully refined from the resolution [ λ/(2 Th) and the lower of λ/(2 Tl) of the first doppler fast fourier transform to λ/(2 Tc), and a special point of the application is that, because the two sound parts H and L are seamlessly connected, the upper doppler measurement limit of the second doppler discrete fourier transform is also seamlessly connected with the lower doppler measurement limit (i.e. resolution) of the first doppler fast fourier transform, and the doppler value at the connection is λ/(2 Tc) j, where j is the output spectral line number of the first doppler fast fourier transform. Therefore, the method provided by the application can completely realize the downward expansion of the Doppler measurement range without continuous blind areas.

In the above embodiment, after performing the first doppler measurement on each tone part of the same beam multiframe in the measurement period to obtain the initial phase of the signal, the initial phase is used as the first virtual time domain sample to perform the second doppler fourier transform. Unlike the conventional quadratic doppler method, which only performs virtual sampling once in each period, here, due to the special design of the waveform of the duet, H and L are continuous, so the maximum measurement speed of the quadratic doppler discrete fourier transform is just equal to the resolution of the first doppler measurement, that is, the measurable range of the first doppler and the measurable range of the second doppler are seamless, and there is substantially no measurement blind area.

Also because the time intervals between the virtual samples are not exactly the same, a discrete fourier transform is used instead of a fast fourier transform, which is streamable compared to the fast fourier transform and therefore not truncated by a fixed length measurement window, and which can be measured over a longer time dimension across a measurement period, resulting in more stable and reliable results.

In one embodiment, in performing S350: the step of screening the subdivided ones of the target points based on the magnitude distribution data may be followed by:

acquiring the maximum output spectral line number of the secondary Doppler according to the minimum measurement time length in all the sound parts and the total time length of at least one measurement period;

selecting a plurality of local maximum values formed by the same real target due to side lobe interference from the output amplitude values of the subdivided targets according to the maximum output spectral line number and the period duration of the scanning frame;

and selecting the spectral line position where the highest amplitude value in the local maximum values is as a real target.

Since the observation vector sequence used in the second-order doppler discrete fourier transform is non-isochronously sampled, it is necessary to suppress side lobe interference if necessary, so that these side lobes can be prevented from generating false local maxima, and the radar can be prevented from generating false targets of false doppler velocity.

For example, in an embodiment, when the H-tone duration Th/measurement frame duration Tf is 1/m, then in each output range of the second doppler discrete fourier transform, there are m aliasing intervals, that is, a target with the velocity Vd of the kth output spectral line in the second doppler discrete fourier transform, which will result in a total of m local maxima in the whole output range of the second doppler discrete fourier transform, and the spectral line positions of these local maxima are:

[k,k+NDDFT/m,k+2*NDDFT/m,k+3*NDDFT/m,…,k+floor(m)*NDDFT/m]

or, for a target with Vd of the speed of any output spectral line in [ k, k + NDDFT/m, k +2 NDDFT/m, k +3 NDDFT/m, …, k + floor (m) NDDFT/m ] of the secondary doppler DFT, a total of m local maxima will be generated in the whole output range of the secondary doppler discrete fourier transform, and the spectral line positions of these local maxima are:

[k,k+NDDFT/m,k+2*NDDFT/m,k+3*NDDFT/m,…,k+floor(m)*NDDFT/m]

the method for avoiding the side lobe interference is provided while the method for the secondary Doppler discrete Fourier transform of non-isochronous sampling is provided, so that the anti-aliasing effect of the secondary Doppler discrete Fourier transform is realized. Theoretical analysis can prove that the amplitude of the k-th spectral line of the multiple local maximum values caused by aliasing effects is the highest, and the amplitudes of other positions are lower than that of the k-th spectral line. The present application therefore proposes a method of selecting, from the above-mentioned plurality of mutually aliased local maxima, only one of the highest-amplitude local maxima as a unique representation of these mutually aliased local maxima for output, thereby avoiding aliasing leading to the generation of false targets.

It should be noted that this multi-choice aliasing suppression method does not significantly affect the subdivision doppler resolution between measurable multiple targets in engineering implementation, and as long as there is no one of m aliasing spectral lines between multiple subdivision targets that happen to fall on each other, it can be output as an independent "local maximum" true target in the amplitude-frequency distribution of the quadratic doppler discrete fourier transform without being affected at all. For example, when the duty cycle of the radar waveform is about 67%, if a dual-beat waveform is used, where Th and Tl each account for about 33% of the duration Tf of a single measurement frame, that is, Th/Tf is 1/3, and if each measurement period includes NF 32 measurement frames, the output spectral lines of the double-doppler discrete fourier transform of the method total 96, and for each output true target, the aliasing position thereof is only Th/Tf 3 (including itself), and the blind area spectral lines are only (3-1)/96 and only about 2%, which is far lower than that of the conventional method of performing fast fourier transform by connecting a plurality of measurement frames of isochronous samples together (the blind area under the isochronous sampling method is at least the waveform dead time ratio, that is, about 33%).

On the other hand, the application also provides a device for processing the subdivided Doppler velocity.

Referring to fig. 5, in an embodiment, the subdivided doppler velocity processing apparatus includes a radar receiver 21 and a processor 22 connected to each other, wherein:

the radar receiver 21 is used for receiving and sending radar echo to the processor 22; the processor 22 is configured to perform the subdivided doppler velocity processing method described above.

It should be noted that, on the basis of some calculation formulas of the present application, some common optimization methods are used for performing transformations, such as fix-point operation, raising or lowering the order of an approximate polynomial, combining polynomials based on multiply-add operation, and the like, which are all within the protection scope of the present application.

According to the subdivided Doppler velocity processing method and device, the multiple sound parts with different Doppler resolution forces are arranged in the radar beam, primary Doppler Fourier transform and secondary Doppler Fourier transform are performed, and the minimum measurement resolution force can be refined.

In order to implement the foregoing radar waveform, in one aspect of the present application, a radar waveform generating circuit is also provided. Fig. 8 is a block diagram of a radar waveform generation circuit according to an embodiment of the present application.

As shown in fig. 8, in an embodiment, the radar waveform generating circuit includes a gate 30, a first memory 10, a second memory 20, a first counter 40, a second counter 50, and a frequency synthesizer 60.

The gate 30 is connected to the first memory 10 and the second memory 20, respectively, wherein the first memory 10 stores data of a first time interval, and the second memory 20 stores data of a second time interval. The gate 30 is also connected to the first counter 40 and the second counter 50, and transmits the gated memory data to the second counter 50 under the control of the signal of the first counter 40.

The second counter 60 is connected to the first counter 40 and the second counter 50, and the second counter 60 generates a setting waveform of the frequency modulated continuous wave according to the first time enable signal transmitted from the second counter 50, and transmits a waveform completion signal to the first counter 40 and the second counter 50 when the setting waveform is completed.

The first counter 40 is used to count the number of the waveform completion signal and transmit the gate transition signal to the gate 30 when the count of the number is completed.

The gate 30 is configured to change the gate from the first time slot to the second time slot according to the gate transition signal to transmit the second time slot as time slot data to the second counter 50.

The second counter 50 is configured to read the time gap data according to the waveform completion signal, perform time counting according to the time gap data, and send a second enable signal to the second counter 60 when the time counting is completed.

The difference of the time gap between the two sections of set waveforms is set, so that different Doppler resolution forces can be achieved between the two groups of waveforms, and the measurement interval can be effectively improved and the minimum measurement resolution force can be further refined through further processing of echoes.

As shown in fig. 9, in an embodiment, the strobe 30 employs a chip 74LS157D, and the first memory 10 and the second memory 20 respectively store four bits of time slot data. The gate 30 includes a first group of data input pins AI, a second group of data input pins BI, a selection pin a/B, and a data output pin Y.

Referring to fig. 9, a first group AI of data input pins is connected to the first memory 10, and a second group BI of data input pins is connected to the second memory 20. The selection pin a/B is connected to the first counter 40, and the data output pin YO is connected to the second counter 50.

Under the gating control of the first counter 40, the gate 30 gates four bits of time-gap data to the second counter 50 for time counting, thereby controlling the time gap required between the two waveforms. In other embodiments, other numbers of bits of time gap data may be selected as desired, such as eight or sixteen bits. Accordingly, other sizes of chips may be used for the gate 30.

As shown in fig. 10, in one embodiment, the first counter 40 includes a first count chip 41, a second count chip 42, and a first not gate M1. Wherein, the counter chip adopts 74LS 161D.

The clock inputs CP of the first counter chip 41 and the second counter chip 42 are connected to the frequency synthesizer 50, respectively. The carry output pin RCO of the first counting chip 41 is connected to the count control pin EP/ET of the second counting chip 42, and the count control pin EP/ET and the clear pin CR of the first counting chip 41 are connected to the high level terminal VCC. The gate 30 is connected to the lowest data output pin Q0 of the second counter chip 42, and the first not gate M1 is connected in series between the second lowest data output pin Q1 of the second counter chip 42 and the clear pin CR. In the present embodiment, the setting control pin LD and the clear pin CR of the first counting chip 41 are connected to each other, the setting control pin LD and the clear pin CR of the second counting chip 42 are connected to each other, and the data input pins D0, D1, D2, D3 of the first counting chip 41 and the data input pins D0, D1, D2, D3 of the second counting chip 42 are all grounded.

In the present embodiment, the first counter chip 41 is configured as a counter of a first specific number (for example, 16), and generates a carry signal to send to the second counter chip 42 for enabling each time a first specific number of waveform completion signals sent by the frequency synthesizer 60 are received. The second counter chip 42 is configured as a second specific digital counter that, when enabled, issues a strobe transition signal to strobe the designated memory every time it receives a second specific number (e.g., 1) of waveform completion signals from the frequency synthesizer 60. It should be noted that, in order to briefly explain the waveform implementation principle of the present application, the present embodiment provides a simplified circuit diagram. Further conventional variations of the circuit based on the principles set forth herein are within the scope of the present application. For example, in other embodiments, counters of other bitscales may be set according to the requirements of different waveform numbers.

As shown in fig. 11, in an embodiment, the frequency synthesizer 60 includes a second and gate M2, a clock source T1, and a frequency synthesizing chip 61, wherein the frequency synthesizing chip 61 employs ADF 4169.

Referring to fig. 11, a first input terminal of the second and gate M2 is connected to the clock source T1, a second input terminal of the second and gate M2 is connected to the second counter 50 for receiving the enable signal, and an output terminal of the second and gate M2 is connected to the STEP signal input terminal STEP of the frequency synthesizing chip 61.

Through the control of the enable signal, the clock signal sent by the clock source T1 can be sent to the frequency synthesis chip 61 within the set time, so as to achieve the purpose of setting the specific time gap.

With continued reference to fig. 11, in an embodiment, the second counter 50 includes a not gate array T2, a first and gate M3, a third counter chip 51, a second not gate M4, and a third not gate M5.

The not gate array T2 is connected between the data input pins D0, D1, D2, D3 of the third counting chip 51 and the gate 30, the first and gate M3 is connected between the data output pins Q0, Q1, Q2, Q3 of the third counting chip 51 and the input terminal of the second not gate M4, and the output terminal of the first and gate M3 is further connected to the frequency synthesizer 60 to output the enable signal. The output terminal of the second not gate M4 is connected to the count control pin EP/ET of the third count chip 51.

The output end of the third not gate M5 is connected to the setting control pin LD of the third counting chip 51, and the clear pin CR of the third counting chip 51 is connected to the high level terminal VCC. An input terminal of the third not gate M5 is connected to the waveform completion signal output terminal COMP of the frequency synthesizing chip 61 to receive the waveform completion signal.

The third counting chip 51 receives a clock signal via a clock input CP. In the present embodiment, the clock input CP of the third counting chip 51 is connected to the clock source T1. When the second counter 50 receives the waveform completion signal of the frequency synthesizer 60, the time slot data sent from the gate 30 is read by the setting control of the third counter chip 51, and the count is performed after the inversion by the not gate array T2, which corresponds to the time countdown according to the clock signal. In other embodiments, the second counter 50 may be configured by an appropriate circuit to count the read time slot data in the forward direction.

When the time gap countdown is completed, all of the data output pins Q0, Q1, Q2, and Q3 of the third counting chip 51 output a high level, and the first and gate M3 and the second not gate M4 output a low level to the count control pin EP/ET of the third counting chip 51, so that the third counting chip 51 stops counting. At the same time, the count completion signal of high level is sent to the frequency synthesizer 60 as an enable signal for outputting the next set waveform.

The frequency synthesizer 60 sends a high waveform completion signal when a set waveform is completed. The low level is sent to the setting control pin LD of the third counting chip 51 through the third not gate M5, so that the third counting chip 51 can read the time slot data again and start the time counting again. In the time gap between the two set waveforms, the frequency synthesizer 60 may stop outputting or may output at a specific frequency during the time counting of the second counter 50.

In some chip arrangements, the frequency synthesizing chip 61 outputs a waveform completion signal of a high level in the last clock cycle of the set waveform. The high waveform completion signal pulls the output signal of the first and gate M3 low. Referring to fig. 11, in an embodiment, the second counter 50 further includes a first nand gate M6, a first input of the first nand gate M6 is connected to the output of the second not gate M4, a second input of the first nand gate M6 is connected to the output of the third not gate M5, and an output of the first nand gate M6 is connected to the frequency synthesizer 60 for outputting the enable signal.

The connection of two NOT gates and one NAND gate is equivalent to the formation of one OR gate circuit. That is, in the output signal of the first and gate M3 or the waveform completion signal of the frequency synthesizer 60, as long as any one of them is at a high level, the effect of the enable signal is achieved, so that the frequency synthesizing chip 61 can receive the last clock signal again in the last clock cycle of the setting waveform, thereby finally ending the output of the current setting waveform.

The setting waveform of the frequency modulated continuous wave output from the frequency synthesizer 60 may be at least one selected from a triangular wave, a sawtooth wave, a step wave, and a sine wave.

On the other hand, the application also provides a radar. FIG. 12 is a block diagram of a radar according to an embodiment of the present application.

Referring to fig. 12, the radar includes an antenna 1 and a radar waveform generating circuit 2 as described above, which are connected to each other.

When radar waveforms having different doppler resolutions are generated by the radar waveform generation circuit 2, the radar waveforms are transmitted to an external space through the antenna 1 to detect a detection target. By further processing the echoes with different Doppler resolutions, the measurement interval can be effectively increased and the minimum measurement resolution can be further refined.

With continued reference to fig. 12, in one embodiment, the radar further includes a master controller 3 for controlling the frequency modulated continuous wave. The main controller 3 is connected to the frequency synthesizer 60 in the radar waveform generating circuit 1.

The waveform parameters of the frequency synthesizer 60 can be flexibly set by the main controller 3, and a flexible means is provided for further improving the measurement interval and further refining the minimum measurement resolution.

On the other hand, the application also provides a radar waveform generation method.

In one embodiment, a radar waveform generation method includes:

acquiring a first enabling signal; generating a set waveform of the frequency-modulated continuous wave according to the first enabling signal, and generating a waveform completion signal when the set waveform is completed; acquiring time gap data according to the waveform completion signal; and counting time according to the time gap data, and generating a second enabling signal when the counting of the time is completed.

and counting the number of the waveform completion signals, and generating a gating conversion signal when the counting of the number is completed so as to replace the first time interval to the second time interval from gating as time interval data.

Sawtooth is a frequency modulated continuous wave commonly used in radar detection. As shown in fig. 13, in an embodiment, the steps of the radar waveform generation method include:

s410: acquiring a first enabling signal; proceeding to step S20;

s420: generating a sawtooth waveform according to the first-time enable signal, and generating a waveform completion signal when one sawtooth waveform is completed; proceeding to step S30;

s430: judging whether the waveform finishing signal reaches the set number of times, if not, entering the step S431; if yes, go to step S432;

s431: acquiring stored first time gap data as time gap data; entering step S440;

s432: acquiring the stored second time gap as time gap data; entering step S440;

s440: and counting time according to the time gap data, and generating a second enabling signal when the counting of the time is completed.

By setting a number of sawtooth waves (for example, 16 sawtooth waves) to form one sound part, in at least two sound part echoes with different Doppler resolution, the measurement interval can be effectively improved, and the minimum measurement resolution can be further refined.

In an embodiment, a computer storage medium has stored thereon a computer program which, when executed by a processor, may implement the radar waveform generation method as described above.

For the method for generating a radar waveform implemented by a processor when executing a computer program, please refer to the above embodiments, which are not described herein again.

In the above embodiments of the radar waveform generation, waveforms of two sound parts are generated with different waveform gaps in order to realize that waveforms of a plurality of sound parts in close proximity have different doppler resolution, but for the sake of simplicity of description, waveform durations, the number of waveforms, and center wavelengths of the two sound parts are set to be the same. In fact, in other embodiments, it may be achieved by setting the waveforms of immediately adjacent sound sections to have different center wavelengths, different waveform durations, different numbers of waveforms, and different waveform gaps, for example, by setting different waveform slopes or different radio frequency bandwidths to obtain different total observed durations, to achieve different doppler resolutions for each sound section.

In order to explain the core technical features of the present application, the radar waveform generating circuit and method provided by the present application use a simple chip and a simple gate circuit to realize two types of time-gap frequency modulated continuous waves, so as to obtain two types of sound parts with different doppler resolutions. In other cases, other software and hardware methods may be used, and through simple transformation and expansion of the core technical features of the present application, for example, a logic circuit is used instead to implement counting, and at least three types of time gap data are used to obtain more than three types of sound portions with different doppler resolutions, which all belong to the protection scope of the present application.

In this document, unless expressly stated or limited otherwise, the terms "mounted," "connected," and "connected" are to be construed broadly, e.g., as meaning either a fixed connection, a removable connection, or an integral connection; can be mechanically or electrically connected; they may be connected directly or indirectly through intervening media, or they may be interconnected between two elements. The specific meaning of the above terms can be understood in a specific case to those of ordinary skill in the art.

As used herein, the ordinal adjectives "first", "second", etc., used to describe an element are merely to distinguish between similar elements and do not imply that the elements so described must be in a given sequence, either temporally, spatially, in ranking, or in any other manner.

As used herein, the meaning of "a plurality" or "a plurality" is two or more unless otherwise specified.

It will be understood by those skilled in the art that all or part of the steps of implementing the above method embodiments may be implemented by hardware associated with program instructions, and the program may be stored in a computer readable storage medium, and when executed, performs the steps including the above method embodiments. The foregoing storage medium includes: various media that can store program codes, such as ROM, RAM, magnetic or optical disks.

The technical features of the embodiments described above may be arbitrarily combined, and for the sake of brevity, all possible combinations of the technical features in the embodiments described above are not described, but should be considered as being within the scope of the present specification as long as there is no contradiction between the combinations of the technical features.

As used herein, the terms "comprises," "comprising," or any other variation thereof, are intended to cover a non-exclusive inclusion, including not only those elements listed, but also other elements not expressly listed.

The above description is only a specific embodiment of the present application, but the scope of the present application is not limited thereto. Any person skilled in the art can easily think of changes or substitutions in the technical scope disclosed in the present application, and all the changes or substitutions are covered in the protection scope of the present application. Therefore, the protection scope of the present application shall be subject to the protection scope of the claims.

Claims

1. A radar waveform generating circuit comprising a gate, a first memory storing a first time gap, a second memory storing a second time gap, a first counter, a second counter, and a frequency synthesizer, wherein:

2. The radar waveform generation circuit of claim 1, wherein the gate includes a first set of data input pins connected to the first memory, a second set of data input pins connected to the second memory, a selection pin connected to the first counter, and a data output pin connected to the second counter.

3. The radar waveform generating circuit according to claim 1, wherein the first counter includes a first counter chip, a second counter chip, and a first not gate, clock inputs of the first counter chip and the second counter chip are respectively connected to the frequency synthesizer, a carry output pin of the first counter chip is connected to a count control pin of the second counter chip, a lowest data output pin of the second counter chip is connected to the gate, and the first not gate is connected in series between a second lowest data output pin and a clear pin of the second counter chip.

4. The radar waveform generation circuit of claim 1 wherein the frequency synthesizer includes a second and gate, a clock source, and a frequency synthesizing chip, a first input of the second and gate being connected to the clock source, a second input of the second and gate being connected to the second counter to receive the enable signal, an output of the second and gate being connected to the frequency synthesizing chip.

5. The radar waveform generation circuit of claim 1, wherein the second counter comprises an array of NOT gates, a first AND gate, a third count chip, a second NOT gate, and a third NOT gate, wherein the array of NOT gates is connected between a data input pin of the third count chip and the gate, the first AND gate is connected between a data output pin of the third count chip and an input terminal of the second NOT gate, an output terminal of the first AND gate is connected to the frequency synthesizer to output the enable signal, an output terminal of the second NOT gate is connected to a count control pin of the third count chip, an output terminal of the third NOT gate is connected to a count control pin of the third count chip, and an input terminal of the third NOT gate is connected to the frequency synthesizer to receive the waveform completion signal.

6. The radar waveform generating circuit of claim 5 wherein the second counter further comprises a first NAND gate, a first input of the first NAND gate being connected to an output of the second NAND gate, a second input of the first NAND gate being connected to an output of the third NAND gate, an output of the first NAND gate being connected to the frequency synthesizer to output the enable signal.

7. A radar comprising an antenna and a radar waveform generating circuit as claimed in any one of claims 1 to 6 connected to each other.

8. The radar of claim 7 further comprising a master controller for controlling the frequency modulated continuous wave, said master controller being connected to a frequency synthesizer in said radar waveform generating circuit.

9. A method of radar waveform generation, comprising:

acquiring a first enabling signal;

acquiring time gap data according to the waveform completion signal;

10. A computer storage medium, characterized in that the computer storage medium has stored thereon a computer program which, when executed by a processor, can implement the radar waveform generation method of claim 9.