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CN111786920A - Multi-system CSS signal demodulation method and device - Google Patents

Multi-system CSS signal demodulation method and device Download PDF

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CN111786920A
CN111786920A CN201910265013.4A CN201910265013A CN111786920A CN 111786920 A CN111786920 A CN 111786920A CN 201910265013 A CN201910265013 A CN 201910265013A CN 111786920 A CN111786920 A CN 111786920A
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cfo
signal
value
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CN111786920B (en
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周海然
杨利
鲍东山
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Shanghai Nufront Electronic Technology Co ltd
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    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2668—Details of algorithms
    • H04L27/2669—Details of algorithms characterised by the domain of operation
    • H04L27/2671—Time domain
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0014—Carrier regulation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2657—Carrier synchronisation
    • H04L27/2659—Coarse or integer frequency offset determination and synchronisation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/26—Systems using multi-frequency codes
    • H04L27/2601—Multicarrier modulation systems
    • H04L27/2647—Arrangements specific to the receiver only
    • H04L27/2655—Synchronisation arrangements
    • H04L27/2657—Carrier synchronisation
    • H04L27/266—Fine or fractional frequency offset determination and synchronisation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
    • H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
    • H04L27/38—Demodulator circuits; Receiver circuits
    • H04L27/3845—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier
    • H04L27/3854—Demodulator circuits; Receiver circuits using non - coherent demodulation, i.e. not using a phase synchronous carrier using a non - coherent carrier, including systems with baseband correction for phase or frequency offset
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00—Modulated-carrier systems
    • H04L27/0014—Carrier regulation
    • H04L2027/0024—Carrier regulation at the receiver end
    • H04L2027/0026—Correction of carrier offset

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Abstract

The application provides a method and a device for demodulating multi-system CSS signals, comprising the following steps: performing analog-to-digital conversion on the multi-system CSS modulation signal to obtain a digital baseband signal; carrying out down-sampling processing on the digital baseband signals; carrying out data detection on the down-sampled signal to detect whether data come or not; when data comes, performing time domain synchronization on input data, and outputting a synchronized signal; searching the synchronized signal for the position of the frame separator SFD; judging to obtain an integer value CFO _ Int of the carrier frequency offset CFO according to the output signal after time domain synchronization and the position of the frame separator SFD; estimating a decimal value CFO _ Frac of a carrier frequency offset CFO; and demodulating the effective data, and performing carrier frequency offset compensation on the demodulated data to obtain final demodulated data.

Description

Multi-system CSS signal demodulation method and device
Technical Field
The invention belongs to the technical field of wireless communication, and particularly relates to a multi-system CSS signal demodulation method and device.
Background
In modern communication systems, communication technologies applied to the internet of things are various, and a multilevel Chirp Spread Spectrum (CSS) (Chirp Spread spectrum) modulation transmitter has the characteristic of ultralow power consumption, so that the CSS (Chirp Spread spectrum) modulation transmitter is superior to the internet of things communication technology. However, the multilevel CSS receiver has not been implemented with an explicit and good technology, and it is difficult for the existing receiver to consider both power consumption and performance. The current industry solution for implementing multilevel CSS receivers is: receiving by adopting a coherent and matched filtering method, but the power consumption and the area are extremely large; the method of differential frequency solving is adopted for receiving, but the performance is poor, and the SNR of the channel is required to be high; the direct FFT method is adopted for synchronization and receiving, the power consumption and the area are extremely large, and the scheme does not meet the characteristics of low power consumption, small area and high performance of the Internet of things.
Disclosure of Invention
In view of the above, the present invention provides a method and an apparatus for demodulating a multilevel CSS signal, which perform synchronization of a time domain and a frequency domain by using a correlation method after down-sampling of a cascaded-integration comb filter CIC, and perform demodulation by using an FFT method, and have low power consumption and area on the premise of high performance.
The application provides a multi-system CSS signal demodulation method, which comprises the following steps:
performing analog-to-digital conversion on the multi-system CSS modulation signal to obtain a digital baseband signal;
performing down-sampling processing on the digital baseband signal to obtain a down-sampled signal;
further, the down-sampling the digital baseband signal includes:
the digital baseband signal is subjected to preliminary down-sampling through a cascade integral comb filter, and high-frequency noise is filtered, so that a preliminary down-sampling signal with the high-frequency noise filtered is obtained;
and the preliminary down-sampling signal is further down-sampled after passing through the compensation filter, the in-band attenuation of the cascade integrator comb filter is compensated, and the down-sampling signal is output.
Carrying out data detection on the down-sampling signal, detecting whether data arrive or not, setting a threshold value of energy detection, calculating a related energy value of a current input data frame, and indicating that the data arrive when the related energy value is greater than the threshold value;
when data comes, performing time domain synchronization on input data, and outputting a synchronized signal;
further, the time domain synchronization of the data includes coarse synchronization and fine synchronization;
the time domain coarse synchronization is to perform time domain shift on the received data according to the frame label and the sampling point label;
and the time domain fine synchronization is to further synchronize the data after the coarse synchronization, find a position index corresponding to the maximum value of the correlation energy value and shift the time domain of the data according to the position index corresponding to the maximum value of the correlation energy value.
Searching the position of a frame separator SFD in the synchronized signal; specifically, the searching for the location of the frame delimiter SFD includes:
searching the position of the frame separator SFD by adopting a correlation method, and calculating a correlation energy value p '(n, k) of the output x' (n, k) after fine synchronization;
find the position index fine _ idx ' corresponding to the maximum value of the correlation energy value p ' (n, k), fine _ idx ' being the position of the frame separation symbol SFD.
Judging to obtain an integer value CFO _ Int of the carrier frequency offset CFO according to the output signal after time domain synchronization and the position of the frame separator SFD;
estimating a decimal value CFO _ Frac of a carrier frequency offset CFO;
the estimating the decimal value of the carrier frequency offset CFO includes:
calculating a correlation value cor (n, k) of the leader sequence x' (n, k);
fourier transform is carried out on the correlation value cor (n, k), amplitude values are solved, and three points with the maximum amplitude values and labels where the three points are located are found;
calculating an approximate signal-to-noise ratio (SNR) of the correlation value cor (n, k) of the preamble sequence;
delaying the leader sequence x '(n, k) by one sampling point, and calculating the approximate signal-to-noise ratio SNR' of the delayed leader sequence;
and obtaining a decimal value CFO _ Frac of the CFO according to the integer values CFO _ Int, SNR and SNR' of the CFO.
Demodulating the effective data to obtain demodulated data; and performing carrier frequency offset CFO compensation on the demodulated data, and adjusting the demodulated data according to the carrier frequency offset integer value CFO _ Int and the carrier frequency offset decimal value CFO _ Frac to obtain final demodulated data.
The present application also provides a multilevel CSS signal demodulating apparatus, including:
the analog-to-digital conversion module is used for performing analog-to-digital conversion on the multi-system CSS modulation signal to obtain a digital baseband signal;
the down-sampling module is used for performing down-sampling processing on the digital baseband signal to obtain a down-sampled signal;
the down-sampling module comprises:
the first down-sampling unit is used for performing preliminary down-sampling on the digital baseband signal through a cascade integration comb filter and filtering high-frequency noise to obtain a preliminary down-sampling signal with the high-frequency noise filtered;
and the second down-sampling unit is used for further down-sampling the preliminary down-sampled signal after passing through the compensation filter, compensating the in-band attenuation of the cascade integrator comb filter and outputting the down-sampled signal.
The detection module is used for detecting data by adopting a relevant method and detecting whether the data comes or not;
the detection module comprises:
a determination unit for setting a threshold for energy detection,
a calculating unit for calculating the correlation energy value of the current input data frame,
and the comparison unit is used for comparing the correlation energy value of the current input data frame with a threshold value of energy detection, and when the correlation energy value is greater than the threshold value, the arrival of data is indicated.
The synchronous processing module is used for carrying out time domain synchronization on input data when the data arrives and outputting a synchronized signal;
the synchronous processing module comprises:
the first synchronization unit is used for carrying out time domain coarse synchronization and carrying out time domain shift on the received data according to the frame label and the sampling point label;
and the second synchronization unit is used for carrying out time domain fine synchronization on the data after the coarse synchronization, searching a position index corresponding to the maximum value of the correlation energy value and carrying out time domain shift on the data according to the position index corresponding to the maximum value of the correlation energy value.
The searching module is used for searching the position of the frame separator SFD in the synchronized signal;
the searching module is further configured to search the position of the frame separator SFD by using a correlation method, and calculate a correlation energy value of the output after the fine synchronization;
and searching a position index corresponding to the maximum value of the correlation energy value, wherein the position index corresponding to the maximum value is the position of the frame separation symbol.
The calculation module is used for calculating carrier frequency offset CFO;
the calculation module comprises:
the integer calculating unit is used for judging to obtain an integer value CFO _ Int of the carrier frequency offset CFO according to the output signal after time domain synchronization and the position of the frame separator SFD;
a decimal calculation unit for estimating a decimal value CFO _ Frac of the carrier frequency offset CFO;
further, the decimal calculating unit is configured to:
fourier transform is carried out on the correlation value cor (n, k), amplitude values are solved, and three points with the maximum amplitude values and labels where the three points are located are found;
calculating an approximate signal-to-noise ratio (SNR) of the correlation value cor (n, k) of the preamble sequence;
delaying the leader sequence by a sampling point, and calculating the approximate signal-to-noise ratio (SNR') of the delayed leader sequence;
and calculating to obtain a decimal value CFO _ Frac of the CFO according to the integer values CFO _ Int and SNR of the CFO.
The device also comprises an adjusting module used for demodulating the effective data; and performing carrier frequency offset CFO compensation on the demodulated data, and adjusting the demodulated data according to the carrier frequency offset integer value CFO _ Int and the carrier frequency offset decimal value CFO _ Frac.
Compared with the scheme in the prior art, the scheme provided by the application has the following advantages:
1. the invention performs down-sampling on the signals after the analog-digital conversion by adopting the cascade integration comb filtering and compensating filter, reduces the areas of synchronization and demodulation and simultaneously reduces the power consumption.
2. The energy detection correlation method provided by the invention can effectively detect the arrival of data, and has the advantages of small required resources and high precision compared with the traditional Schmidl-Cox detection method.
3. The carrier frequency offset detection and compensation method provided by the invention can effectively compensate the carrier frequency offset, and has high speed and high resource utilization rate compared with the traditional methods for solving the signal frequency and the like.
For the purposes of the foregoing and related ends, the one or more embodiments include the features hereinafter fully described and particularly pointed out in the claims. The following description and the annexed drawings set forth in detail certain illustrative aspects and are indicative of but a few of the various ways in which the principles of the various embodiments may be employed. Other benefits and novel features will become apparent from the following detailed description when considered in conjunction with the drawings and the disclosed embodiments are intended to include all such aspects and their equivalents.
Drawings
FIG. 1 is a flow chart of the generation of a multilevel CSS modulated signal provided by the present invention;
fig. 2 is a flowchart of a method for demodulating multilevel CSS modulated signals according to an embodiment of the invention;
FIG. 2a is a flowchart of a down-sampling process according to an embodiment of the present invention;
fig. 2b is a flowchart of a time domain synchronization process according to an embodiment of the present invention;
fig. 2c is a flowchart of a fractional method for estimating carrier frequency offset CFO according to an embodiment of the present invention;
fig. 3 is a system block diagram of a multilevel CSS modulated signal demodulating apparatus according to a second embodiment of the present invention.
Detailed Description
The following description and the drawings sufficiently illustrate specific embodiments of the invention to enable those skilled in the art to practice them. Other embodiments may incorporate structural, logical, electrical, process, and other changes. The examples merely typify possible variations. Individual components and functions are optional unless explicitly required, and the sequence of operations may vary. Portions and features of some embodiments may be included in or substituted for those of others. The scope of embodiments of the invention encompasses the full ambit of the claims, as well as all available equivalents of the claims. Embodiments of the invention may be referred to herein, individually or collectively, by the term "invention" merely for convenience and without intending to voluntarily limit the scope of this application to any single invention or inventive concept if more than one is in fact disclosed.
Example one
Chirp Spread spectrum css (chirp Spread spectrum) is a way of Spread spectrum transmission, done by the chirp signal itself, without frequency spreading with a pseudo-random Sequence compared to direct Sequence Spread spectrum dsss (direct Sequence Spread spectrum). The Chirp signal can be generally expressed as:
Figure BDA0002016497520000061
wherein A is amplitude;
Figure BDA0002016497520000062
is the phase; μ is the slope of its frequency curve.
A signal waveform under CSS modulation can be defined as
Figure BDA0002016497520000063
The multilevel CSS signal in this application is a variation of the CSS signal, and specifically, the instantaneous frequency of the multilevel CSS signal is represented as:
Figure BDA0002016497520000064
fithe starting frequencies of different chirp symbols are related to the data to be transmitted by:
Figure BDA0002016497520000065
d (k) is the data to be transmitted, k being the index of the data. A single Chirp symbol can only transmit 1 bit data (+) under basic CSS modulation1 or-1), and a single Chirp symbol can transmit s-bit data under multilevel CSS modulation.
The multilevel CSS modulation transmitter has the characteristics of low power consumption and high performance, however, the multilevel CSS receiver is not realized by a clear and good technology, and the application provides a method and a device for demodulating multilevel CSS modulation signals aiming at the multilevel CSS receiver.
As shown in fig. 1, a multilevel CSS modulated signal is first acquired: an antenna receives a radio frequency signal, the radio frequency signal is amplified by a Low Noise Amplifier (LNA) (low noise amplifier), a carrier is generated by a phase-locked loop (PLL) (phase locked loops), a Variable Gain Amplifier (VGA) (variable gain amplifiers) adjusts the gain of the signal, a high frequency signal is filtered after a low pass filter (LP) (low pass), and multi-system CSS modulation signals Din _ I and Din _ Q can be obtained.
The application provides a multilevel CSS signal demodulation method, as shown in fig. 2, including:
s101, performing analog-to-digital conversion on the multi-system CSS modulation signal to obtain a digital baseband signal;
s102, performing down-sampling processing on the digital baseband signal to obtain a down-sampled signal;
further, as shown in fig. 2a, the down-sampling the digital baseband signal includes:
s102a, performing preliminary down-sampling on the digital baseband signal through a cascade integration comb filter CIC (Cascade IntegralComb), and filtering high-frequency noise to obtain a preliminary down-sampling signal with the high-frequency noise filtered;
and S102b, the primary down-sampling signal is further down-sampled after passing through a compensation filter Comp (Compensator filter), in-band attenuation of a CIC (Cascade integrator comb filter) is compensated, and a down-sampling signal x (n, k) is output.
S103, carrying out data detection on the down-sampling signal x (n, k) to detect whether data comes or not,
specifically, a correlation method is adopted for detection, and a correlation energy value p (n, k) of the down-sampled signal is calculated:
Figure BDA0002016497520000071
wherein k is a label of an input data frame, M is a length of a frame, n is a label of a sampling point in a frame, and Up _ c (n) is a reference signal generated by a lookup Table (Loop Up Table) Chirp _ LUT;
setting threshold TH of energy detection1The correlation energy value p (n, k) of the current input data frame is compared with the threshold value TH1By comparison, when p (n, k) > TH1Indicating the arrival of data.
S104, when data arrive, performing time domain synchronization on input data, and outputting a synchronized signal;
specifically, the time domain synchronization of the data, as shown in fig. 2b, includes coarse synchronization and fine synchronization,
s104a, time domain coarse synchronization, namely performing time domain shift on received data according to a frame label and a sampling point label;
when the data arrival is detected in S103, that is, the correlation energy value p (n, k) of the current input data frame is detected to be larger than the threshold TH1And at the moment, the frame label k 'and the sampling point label n' perform time domain shift on the received data, and move k 'frames and n' sampling point periods to obtain the data after coarse synchronization.
S104b, time domain fine synchronization, further synchronizing the data after coarse synchronization,
specifically, a position index [ -, fine _ idx ] ═ max (p (n, k)) corresponding to the maximum value of the correlation energy value is found, time domain shifting is performed on the data according to the index, that is, the data after coarse synchronization is moved by (fine _ idx-1) sampling point periods, wherein fine _ idx is a sampling point label of fine synchronization, and fine synchronization is performed according to the label to obtain a synchronized output signal x' (n, k).
S105, searching the position of a frame separator SFD (StartFrame Delimiter) in the synchronized signal x' (n, k);
searching the position of the frame separator SFD by adopting a correlation method, and calculating the correlation energy value p' (n, k) of the output after fine synchronization;
further, in the above-mentioned case,
Figure BDA0002016497520000081
x' (n, k) is the output after time domain fine synchronization, k is the index of the input data frame, M is the length of one frame, n is the index of the sampling point in one frame, and dn _ c (n) is the reference signal generated by the Chirp _ LUT;
the position index [ -, fine _ idx '] ═ max (p (n, k)') corresponding to the maximum value of the correlation energy value p '(n, k) is found, and the position index fine _ idx' is the position of the frame separation symbol SFD.
S106, judging to obtain an integer value CFO _ Int of Carrier Frequency Offset (CFO) (Carrier frequency offset) according to the sampling point mark fine _ idx in the time domain synchronization and the position fine _ idx' where the frame separator SFD is located;
specifically, according to the time domain synchronization and the position fine _ idx' of the frame separator SFD, the integer value of the carrier frequency offset CFO can be calculated:
Figure BDA0002016497520000082
s107, estimating a decimal value CFO _ Frac of the carrier frequency offset CFO;
estimating that data to be transmitted by a transmitter comprises a Preamble sequence (Preamble), wherein symbols in the Preamble sequence are up _ c (n); as shown in fig. 2c, includes:
s107a, calculating a correlation value cor '(n, k) of a leader sequence of data x' (n, k) after time domain fine synchronization; performing dot multiplication on the preamble sequence of the data x' (n, k) after time domain fine synchronization and the reference signal up _ c (n): obtaining a correlation value cor (n, k) of the leader sequence by using cor (n, k) ═ x' (n, k) × up _ c (n);
s107b, Fourier transform is carried out on the correlation value COR (N, K), amplitude is obtained, COR (N, K) ═ FFT { COR (N, K) } I, and three points with the maximum amplitude and the mark number [ m ] where the three points are located are found1,m2,m3,p1,p2,p3]Max3(COR (N, K)), where m1～m3At the maximum of three values, p1～p3The label number corresponding to the maximum three values;
s107c, calculating the approximate signal-to-noise ratio SNR of the correlation value cor (n, k) of the leader sequence:
Figure BDA0002016497520000091
s107d, delaying the preamble sequence of the data after the time domain fine synchronization by one sampling point, and calculating the approximate signal-to-noise ratio of the delayed preamble sequence;
for example, the data after time domain fine synchronization is delayed by 1 sampling point, and is fourier transformed and amplitude calculated with reference signal up _ c (n) point multiplied by cor '(n, k) ═ x' (n +1, k) × up _ c (n): COR '(N, K) ═ FFT { COR' (N, K) } |, find the three points with the greatest amplitude and their indices: [ m'1,m'2,m'3,p'1,p'2,p'3]Max3(COR ' (N, K)), and the approximate SNR ' after the delay is calculated '
Figure BDA0002016497520000092
Wherein m is1～m3The maximum of three values; p1 ', p2 ' and p3 ' are the labels of the three maximums;
it should be noted that, if there is no CFO fractional part or other non-ideality factor, the SNR' will be higher than the threshold TH2. Demodulation result do thereofp_realThe index p being the maximum of the amplitudes1’。
S107e, obtaining a decimal value of the CFO according to the integer value of the CFO, the SNR and the SNR';
in particular, the method comprises the following steps of,
1) when the value of CFO _ Int is an integer, the SNR is judged:
SNR is greater than threshold TH2If so, it indicates that there is no fractional part of CFO, and the CFO value CFO _ Int ═ round (CFO _ Int), CFO _ Frac ═ 0, round is rounding function;
SNR is less than threshold TH2Then, it indicates that there is a CFO fractional part, and the CFO value CFO _ Int' is round (CFO _ Int);
further, when there is a CFO fractional part, it is necessary to determine whether the polarity is positive or negative: the up-chirp signal in the preamble train undergoes demodulation in step S107b to result in dop_realBecause there are decimal pointsIn part, the correct result may be p1Or p2；
The ideal demodulation result should be dop_ideal＝round(CFO_Int)；
Will dop_realComparing with ideal demodulation result, and taking p1And p2Middle and closest dop_idealOne number of (a) is the result of correct demodulation of the up-chirp signal in the preamble sequence, denoted as p0(ii) a If p is0Is less than p1Then, it means that the polarity of the CFO decimal is positive, and is recorded as: CFO _ Frac 'is 1, if the value of p0 is greater than p 1', it indicates that the polarity of the CFO decimal is negative, and it is noted as: CFO _ Frac ═ 1.
2) When the value of CFO _ Int is decimal, the SNR' is determined:
SNR' is greater than threshold TH2If so, it indicates that there is no fractional part of CFO, CFO _ Frac 'is 0, and the CFO value CFO _ Int' is round (CFO _ Int + 0.5);
SNR' is less than threshold TH2Then, it indicates that there is a CFO fractional part, and the real CFO value CFO _ Int ═ round (CFO _ Int +0.5) at this time;
when the CFO decimal exists, the polarity needs to be judged to be positive or negative: the demodulation result in step S107d is dop_realBecause of the presence of decimal numbers, the correct result may be p1' or p2’；
The ideal demodulation result is dop_ideal＝round(CFO_Int+0.5)
Will dop_realComparing with ideal demodulation result to obtain p'1Or p'2Closest to dop_idealOne number of (a) is the result of correct demodulation of the up-chirp signal in the preamble sequence, denoted as p0(ii) a If p is0Is less than p1', indicates that the polarity of the CFO decimal is positive, and is noted as: CFO _ Frac' is 1 if p0Is greater than p1', then, indicates that the polarity of the CFO decimal is negative, and is noted as: CFO _ Frac ═ 1.
S108, demodulating the effective data, and performing carrier frequency offset compensation on the data to obtain demodulated data;
carrying out correlation calculation on the effective data x '(N, K) to obtain a correlation value COR' (N, K);
COR”(N,K)＝|FFT{cor”(n,k)}|
find the two points with the maximum amplitude and the labels where they are: [ m ] "1,m”2,p”1,p”2]Max2(COR "(N, K)); wherein m is1"and m2"two values of maximum amplitude, p1"and p2"is the label corresponding to the largest two amplitudes.
S109, carrier frequency offset CFO compensation is carried out on the demodulated data, the demodulated data are adjusted according to the carrier frequency offset integral value CFO _ Int and the carrier frequency offset decimal value CFO _ Frac, and finally output demodulated data Do are obtained;
when CFO _ Frac is 0, Do ═ p "1-CFO_Int，
When CFO _ Frac is not 0:
1) CFO _ Frac' ═ 1: do ═ min (p "1,p”2)-CFO_Int
2) CFO _ Frac' ═ -1: do ═ max (p "1,p”2)-CFO_Int
Example two
The present application provides a multilevel CSS signal demodulating apparatus 200, as shown in fig. 3, including:
the analog-to-digital conversion module 210 is configured to perform analog-to-digital conversion on the multilevel CSS modulation signal to obtain a digital baseband signal;
a down-sampling module 220, configured to perform down-sampling processing on the digital baseband signal to obtain a down-sampled signal;
further, the down-sampling module 220 includes:
the first down-sampling unit 220a is configured to perform preliminary down-sampling on the digital baseband signal by using a cascaded integrator-comb filter CIC, and filter high-frequency noise to obtain a preliminary down-sampling signal with the high-frequency noise filtered;
and a second down-sampling unit 220b, configured to down-sample the preliminary down-sampled signal further after passing through a compensation filter Comp, compensate for in-band attenuation of the cascaded integrator-comb filter CIC, and output a down-sampled signal x (n, k).
A detection module 230, configured to perform data detection by using a correlation method, and detect whether data arrives;
the detection module 230 includes:
a determination unit 230a for setting an energy detection threshold TH1；
A calculating unit 230b for calculating a correlation energy value of the current input data frame; calculating a correlation energy value p (n, k) of the down-sampled signal:
Figure BDA0002016497520000111
wherein k is a label of an input data frame, M is a length of a frame, n is a label of a sampling point in a frame, and Up _ c (n) is a reference signal generated by a lookup Table (Loop Up Table) Chirp _ LUT;
a comparing unit 230c for comparing the correlation energy value p (n, k) of the current input data frame with the threshold value TH1For comparison, when p (n, k)>TH1Indicating the arrival of data.
A synchronization processing module 240, configured to perform time domain synchronization on input data when data arrives, and output a synchronized signal;
the synchronization processing module 240 includes:
a first synchronization unit 240a, configured to perform time domain coarse synchronization, and perform time domain shift on received data according to a frame label and a sampling point label;
specifically, when the data arrival is detected in S103, that is, the correlation energy value p (n, k) of the current input data frame is greater than the threshold TH1And recording the frame mark number k 'and the sampling point mark number n' at the moment, and performing time domain shift on the received data, and moving k 'frames and n' sampling point periods to obtain the data after coarse synchronization.
A second synchronization unit 240b, configured to perform time domain fine synchronization on the data after the coarse synchronization,
specifically, a position index [ -, fine _ idx ] ═ max (p (n, k)) corresponding to the maximum value of the correlation energy value is found, time domain shifting is performed on the data according to the index, that is, the data after coarse synchronization is moved by (fine _ idx-1) sampling point periods, wherein fine _ idx is a sampling point label of fine synchronization, and fine synchronization is performed according to the label to obtain a synchronized output signal x' (n, k).
A searching module 250, configured to search for a location of a frame separator SFD in the synchronized signal;
the searching module 250 is further configured to search a position where the frame separator SFD is located by using a correlation method, and calculate a correlation energy value p '(n, k) of the output x' (n, k) after fine synchronization;
further, in the above-mentioned case,
Figure BDA0002016497520000121
k is the label of the input data frame, M is the length of a frame, n is the label of a sampling point in a frame, dn _ c (n) is a reference signal generated by a Chirp _ LUT, x '(n, k) is the output after time domain fine synchronization, and p' (n, k) is a correlation energy value;
the search module 250 is further configured to,
finding the position index corresponding to the maximum value of the correlation energy value p' (n, k):
fmidx ', which is the position of the frame separation symbol, max (p (n, k)').
A calculating module 260, configured to calculate a carrier frequency offset CFO;
the calculating module 260 includes:
the integer calculating unit 260a is configured to determine to obtain an integer value CFO _ Int of the carrier frequency offset CFO according to the output signal after the time domain synchronization and the position of the frame separator SFD;
specifically, according to the position of the time domain synchronization and the frame separator SFD, the integer value of the carrier frequency offset CFO can be calculated:
Figure BDA0002016497520000131
a decimal calculating unit 260b for estimating a decimal value CFO _ Frac of the carrier frequency offset CFO.
The fraction calculating unit 260b is specifically configured to:
data transmitted by a transmitter comprises a Preamble sequence (Preamble), and symbols in the Preamble sequence are up _ c (n);
calculating a correlation value cor (n, k) of the leader sequence x' (n, k); performing dot multiplication on a preamble sequence of data x' (n, k) subjected to time domain fine synchronization and a reference signal up _ c (n) of a Chirp _ LUT: cor (n, k) ═ x' (n, k) × up _ c (n);
obtaining a correlation value cor (n, k) of the leader sequence; fourier transform is carried out on the correlation value COR (N, K), amplitude COR (N, K) is obtained as | FFT { COR (N, K) } |, and three points with the maximum amplitude and the mark number [ m, K ] where the three points are located are found1,m2,m3,p1,p2,p3]Max3(COR (N, K)), where m1～m3At the maximum of three values, p1～p3The label number corresponding to the maximum three values;
calculating an approximate signal-to-noise ratio (SNR) of the preamble sequence correlation value cor (n, k):
Figure BDA0002016497520000132
delaying the preamble sequence of the data after the time domain fine synchronization by another sampling point, and calculating the approximate signal-to-noise ratio of the delayed preamble sequence;
illustratively, delaying the data after time domain fine synchronization by one sampling point and performing point multiplication on the data and the reference signal: fourier transform is performed on cor '(n, k) ═ x' (n +1, k) × up _ c (n), and the amplitude is calculated: COR '(N, K) ═ FFT { COR' (N, K) } |, find the three points with the greatest amplitude and their indices: [ m'1,m'2,m'3,p'1,p'2,p'3]Max3(COR' (N, K)), and SNR:
Figure BDA0002016497520000133
obtaining a decimal value of the CFO according to the integral value of the CFO, SNR and SNR';
in particular, the method comprises the following steps of,
1) when the value of CFO _ Int is an integer, the SNR is judged:
SNR is greater than threshold TH2If so, it indicates that there is no fractional part of CFO, and the CFO value CFO _ Int ═ round (CFO _ Int), CFO _ Frac ═ 0, round is rounding function;
SNR is less than threshold TH2Then, it indicates that there is a CFO fractional part, and the CFO value CFO _ Int' is round (CFO _ Int);
further, when there is a CFO fractional part, it is necessary to determine whether the polarity is positive or negative: the up-chirp signal in the preamble train undergoes demodulation in step S107b to result in dop_realBecause of the fractional part, the correct result may be p1Or p2；
The ideal demodulation result should be dop_idealRound (CFO _ Int); will dop_realAnd the ideal demodulation result dop_idealComparing, and taking p1And p2Middle closest to dp_idealOne number of (a) is the result of correct demodulation of the up-chirp signal in the preamble sequence, denoted as p0(ii) a If p is0Is less than p1Then, it means that the polarity of the CFO decimal is positive, and is recorded as: CFO _ Frac' is 1 if p0Is greater than p'1Then, it means that the polarity of the CFO decimal is negative, and is recorded as: CFO _ Frac ═ 1.
2) When the value of CFO _ Int is decimal, the SNR' is determined:
SNR' is greater than threshold TH2If so, it indicates that there is no fractional part of CFO, CFO _ Frac 'is 0, and the CFO value CFO _ Int' is round (CFO _ Int + 0.5);
SNR' is less than threshold TH2Then, it indicates that there is a CFO fractional part, and the real CFO value CFO _ Int ═ round (CFO _ Int +0.5) at this time;
when the CFO decimal exists, the polarity needs to be judged to be positive or negative: the demodulation result in step S107d is dop_realBecause of the presence of decimal numbers, the correct result may be p1' or p2’；
The ideal demodulation result is dop_idealRound (CFO _ Int +0.5), compare dp_realAnd the ideal demodulation result dop_idealFor comparison, p 'was taken'1Or p'2Closest to dop_idealOne number of (a) is the result of correct demodulation of the up-chirp signal in the preamble sequence, denoted as p0(ii) a If p is0Is less than p1', indicates that the polarity of the CFO decimal is positive, and is noted as: CFO _ Frac' is 1 if p0Is greater than p1', then, indicates that the polarity of the CFO decimal is negative, and is noted as: CFO _ Frac ═ 1.
An adjusting module 270, configured to perform carrier frequency offset CFO compensation on the demodulated data, and adjust the demodulated data according to the carrier frequency offset integer value CFO _ Int and the carrier frequency offset decimal value CFO _ Frac.
The adjusting module is used for demodulating the effective data and carrying out carrier frequency offset compensation on the data to obtain demodulated data;
carrying out correlation calculation on the effective data x '(N, K) to obtain a correlation value COR' (N, K);
COR”(N,K)＝|FFT{cor”(n,k)}|
find the two points with the maximum amplitude and the labels where they are: [ m ] "1,m”2,p”1,p”2]Max2(COR "(N, K)); wherein m is1"and m2"two values of maximum amplitude, p1"and p2"is the label corresponding to the largest two amplitudes.
The adjusting module is used for performing carrier frequency offset CFO compensation on the demodulated data, and adjusting the demodulated data according to the carrier frequency offset integer value CFO _ Int and the carrier frequency offset decimal value CFO _ Frac to obtain finally output demodulated data Do;
when CFO _ Frac is 0, Do ═ p "1-CFO_Int，
When CFO _ Frac is not 0:
1) CFO _ Frac' ═ 1: do ═ min (p "1,p”2)-CFO_Int
2) CFO _ Frac' ═ -1: do ═ max (p "1,p”2)-CFO_Int。
After CIC down-sampling, the time domain and the frequency domain are synchronized by a related method, and demodulation is performed by an FFT method, so that the method and the device have lower power consumption and area on the premise of high performance.
Those of skill in the art will understand that the various exemplary method steps and apparatus elements described in connection with the embodiments disclosed herein can be implemented as electronic hardware, software, or combinations of both. To clearly illustrate this interchangeability of hardware and software, various illustrative steps and elements have been described above generally in terms of their functionality. Whether such functionality is implemented as hardware or software depends upon the particular application and design constraints imposed on the overall system. Skilled artisans may implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the present invention.
The steps of a method described in connection with the embodiments disclosed above may be embodied directly in hardware, in a software module executed by a processor, or in a combination of the two. A software module may reside in RAM memory, flash memory, ROM memory, EPROM memory, EEPROM memory, registers, hard disk, a removable disk, a CD-ROM, or any other form of storage medium known in the art. An exemplary storage medium is coupled to the processor such the processor can read information from, and write information to, the storage medium. In the alternative, the storage medium may be integral to the processor. The processor and the storage medium may reside in an ASIC. The ASIC may reside in a subscriber station. In the alternative, the processor and the storage medium may reside as discrete components in a subscriber station.
The disclosed embodiments are provided to enable those skilled in the art to make or use the invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the scope or spirit of the invention. The above-described embodiments are merely preferred embodiments of the present invention, which should not be construed as limiting the invention, and any modifications, equivalents, improvements, etc. made within the spirit and principle of the present invention should be included in the scope of the present invention.

Claims (12)

1. A method for demodulating multilevel CSS signals, comprising:
performing analog-to-digital conversion on the multi-system CSS modulation signal to obtain a digital baseband signal;
performing down-sampling processing on the digital baseband signal to obtain a down-sampled signal;
carrying out data detection on the down-sampling signal, detecting whether data arrive or not, setting a threshold value of energy detection, calculating a related energy value of a current input data frame, and indicating that the data arrive when the related energy value is greater than the threshold value;
when data comes, performing time domain synchronization on input data, and outputting a synchronized signal;
searching the position of a frame separator SFD in the synchronized signal;
judging to obtain an integer value CFO _ Int of the carrier frequency offset CFO according to the output signal after time domain synchronization and the position of the frame separator SFD;
estimating a decimal value CFO _ Frac of a carrier frequency offset CFO;
demodulating the effective data to obtain demodulated data; and performing carrier frequency offset CFO compensation on the demodulated data, and adjusting the demodulated data according to the carrier frequency offset integer value CFO _ Int and the carrier frequency offset decimal value CFO _ Frac to obtain final demodulated data.
2. The method of claim 1, wherein the down-sampling the digital baseband signal comprises:
the digital baseband signal is subjected to preliminary down-sampling through a cascade integral comb filter, and high-frequency noise is filtered, so that a preliminary down-sampling signal with the high-frequency noise filtered is obtained;
and the preliminary down-sampling signal is further down-sampled after passing through the compensation filter, the in-band attenuation of the cascade integrator comb filter is compensated, and the down-sampling signal is output.
3. The method of claim 1, wherein the time domain synchronizing the data comprises coarse synchronization and fine synchronization;
the time domain coarse synchronization is to perform time domain shift on the received data according to the frame label and the sampling point label;
and the time domain fine synchronization is to further synchronize the data after the coarse synchronization, find a position index corresponding to the maximum value of the correlation energy value and shift the time domain of the data according to the position index corresponding to the maximum value of the correlation energy value.
4. The method of claim 1, wherein the searching for the location of the frame delimiter SFD comprises:
searching the position of the frame separator SFD by adopting a correlation method, and calculating a correlation energy value p '(n, k) of the output x' (n, k) after fine synchronization;
find the position index fine _ idx ' corresponding to the maximum value of the correlation energy value p ' (n, k), fine _ idx ' being the position of the frame separation symbol SFD.
5. The method of claim 1, wherein estimating the fractional value of the carrier frequency offset, CFO, comprises:
calculating a correlation value cor (n, k) of the leader sequence x' (n, k);
fourier transform is carried out on the correlation value cor (n, k), amplitude values are solved, and three points with the maximum amplitude values and labels where the three points are located are found;
calculating an approximate signal-to-noise ratio (SNR) of the correlation value cor (n, k) of the preamble sequence;
delaying the leader sequence x '(n, k) by one sampling point, and calculating the approximate signal-to-noise ratio SNR' of the delayed leader sequence;
and calculating to obtain a decimal value CFO _ Frac of the CFO according to the integer values CFO _ Int, SNR and SNR' of the CFO.
6. A multilevel CSS signal demodulation apparatus, comprising:
the analog-to-digital conversion module is used for performing analog-to-digital conversion on the multi-system CSS modulation signal to obtain a digital baseband signal;
the down-sampling module is used for performing down-sampling processing on the digital baseband signal to obtain a down-sampled signal;
the detection module is used for detecting data by adopting a relevant method and detecting whether the data comes or not;
the synchronous processing module is used for carrying out time domain synchronization on input data when the data arrives and outputting a synchronized signal;
the searching module is used for searching the position of the frame separator SFD in the synchronized signal;
the calculation module is used for calculating carrier frequency offset CFO;
the adjusting module is used for demodulating the effective data; and performing carrier frequency offset CFO compensation on the demodulated data, and adjusting the demodulated data according to the carrier frequency offset integer value CFO _ Int and the carrier frequency offset decimal value CFO _ Frac.
7. The multilevel CSS signal demodulation apparatus of claim 6,
the down-sampling module comprises:
the first down-sampling unit is used for performing preliminary down-sampling on the digital baseband signal through a cascade integration comb filter and filtering high-frequency noise to obtain a preliminary down-sampling signal with the high-frequency noise filtered;
and the second down-sampling unit is used for further down-sampling the preliminary down-sampled signal after passing through the compensation filter, compensating the in-band attenuation of the cascade integrator comb filter and outputting the down-sampled signal.
8. The multilevel CSS signal demodulation apparatus of claim 6,
the detection module comprises:
a determination unit for setting a threshold value for energy detection;
the calculating unit is used for calculating the correlation energy value of the current input data frame;
and the comparison unit is used for comparing the correlation energy value of the current input data frame with a threshold value of energy detection, and when the correlation energy value is greater than the threshold value, the arrival of data is indicated.
9. The multilevel CSS signal demodulation apparatus of claim 6,
the synchronous processing module comprises:
the first synchronization unit is used for carrying out time domain coarse synchronization and carrying out time domain shift on the received data according to the frame label and the sampling point label;
and the second synchronization unit is used for carrying out time domain fine synchronization on the data after the coarse synchronization, searching a position index corresponding to the maximum value of the correlation energy value and carrying out time domain shift on the data according to the position index corresponding to the maximum value of the correlation energy value.
10. The multilevel CSS signal demodulation apparatus of claim 6,
the searching module is further configured to search the position of the frame separator SFD by using a correlation method, and calculate a correlation energy value of the output after the fine synchronization;
and searching a position index corresponding to the maximum value of the correlation energy value, wherein the position index corresponding to the maximum value is the position of the frame separation symbol.
11. The multilevel CSS signal demodulation apparatus of claim 6,
the calculation module comprises:
the integer calculating unit is used for judging to obtain an integer value CFO _ Int of the carrier frequency offset CFO according to the output signal after time domain synchronization and the position of the frame separator SFD;
and the decimal calculation unit is used for estimating a decimal value CFO _ Frac of the carrier frequency offset CFO.
12. The multilevel CSS signal demodulation apparatus of claim 11,
the decimal calculating unit is used for:
fourier transform is carried out on the correlation value cor (n, k), amplitude values are solved, and three points with the maximum amplitude values and labels where the three points are located are found;
calculating an approximate signal-to-noise ratio (SNR) of the correlation value cor (n, k) of the preamble sequence;
delaying the leader sequence by a sampling point, and calculating the approximate signal-to-noise ratio (SNR') of the delayed leader sequence;
and calculating to obtain a decimal value CFO _ Frac of the CFO according to the integer values CFO _ Int, SNR and SNR' of the CFO.
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