CN111525828B - Control Method of Bidirectional Isolated Resonant Power Converter Based on Virtual Synchronous Motor - Google Patents

Abstract

The invention provides a control method of a bidirectional isolation type resonant power converter based on a virtual synchronous motor, which aims to solve the problems of lack of rotational inertia, low voltage stability of a power electronic converter, efficiency reduction caused by large reactive power in the operation process and the like in the charging and discharging process of an electric automobile. The bidirectional power converter is composed of a DC/DC level and a DC/AC level, the DC/AC level three-phase converter can be equivalent to a synchronous motor according to the structural similarity of a three-phase synchronous motor model and the three-phase converter, the whole electric vehicle charging pile is equivalent to a synchronous motor at the grid connection point of the electric vehicle charging pile, and the synchronous motor can adaptively respond to voltage and frequency disturbance of a power grid and provide necessary inertia and damping for the power grid. In order to overcome the defect that the power loss is caused by large reactive current of the traditional DAB converter, the zero voltage conduction and the zero current turn-off of a switching device of the interface converter are realized by adding the resonance module, and the integral operation efficiency of the converter is improved.

Description

Control Method of Bidirectional Isolated Resonant Power Converter Based on Virtual Synchronous Motor

technical field

The invention relates to the field of two-way control of the interaction between the large power grid and the power battery of an electric vehicle, in particular a control method for a two-way isolated resonant power converter based on a virtual synchronous motor, which is suitable for realizing friendly and efficient interaction between an electric vehicle and a power grid and energy conservation two-way flow.

Background technique

The shortage of fossil fuels and concerns about air pollution have accelerated the electrification of vehicles. The interconnection of a large number of electric vehicles with the grid will help stabilize the impact of intermittent renewable energy on the grid, and it will also become one of the effective emergency power alternative solutions, which has been generally recognized and welcomed around the world. The performance of electric vehicle charging/discharging/electrical appliances is a key link to ensure the efficiency, speed, and friendly interaction with the grid of electric vehicles.

Bidirectional interface converters that can regulate bidirectional power flow are an important component of electric vehicle chargers. For the bidirectional interface converter, on the one hand, it is required that the electric vehicle charging/discharging equipment and the distribution network have good interaction characteristics, and have high stability and steady-state accuracy when a transient fault occurs in the distribution network. Some scholars have proposed a two-way droop control method that uses the frequency and voltage on both sides of the AC and DC to control the power flow direction, so that both sides of the AC and DC can bear the load evenly. Grid stability is impacted. In order to increase the stability of the system, the virtual synchronous motor control theory and method of the converter have been proposed in recent years, and the three-phase converter is equivalent to a virtual synchronous motor. Some scholars have studied the implementation scheme of virtual synchronous generator with virtual inertia and the control strategy as an inverter power supply, but the interface converter in this scheme can only be applied to a single power flow direction, and can only be applied to power electronic converters in specific occasions .

On the other hand, the charging and discharging process of electric vehicle charging equipment is required to be as fast and efficient as possible, so as to improve the service life and operation safety of electric vehicle power batteries, and at the same time reduce the energy loss in the charging and discharging process. To increase the efficiency of a bidirectional interface converter, it should meet various requirements such as wide output voltage regulation, low electrical stress, no snubber circuit, low circulating current and good switching conditions, and buck/boost operation. The isolation/bidirectional PWM converter structure is adopted to meet the step-down/boost operation and realize bidirectional power flow, but the high-frequency inverter on the current feeding side will be subjected to strong voltage stress caused by the leakage inductance of the converter, which is a bidirectional isolation type The main obstacle to improving the efficiency of the converter. Therefore, for the bidirectional converter operating at high voltage and high power, the DAB (dual-active-bridge) structure suitable for various voltages and high power can be used to significantly improve the transmission power of the interface converter. However, the traditional DAB converter has a large reactive current, which creates electrical stress on its switching elements and increases power loss, reducing the overall efficiency of the interface converter. Therefore, there are still many defects in the existing converter-related control technology in the charging and discharging of electric vehicles. The bidirectional power converter needs a new control method that can not only improve the operating efficiency but also interact with the grid in a friendly manner.

Contents of the invention

In order to solve the problems of lack of moment of inertia in the charging and discharging process of electric vehicles, low voltage stability of power electronic converters, and efficiency reduction caused by large reactive power during operation, the present invention provides a two-way isolation type based on virtual synchronous motors. Resonant power converter control method.

The present invention is realized through the following technical scheme: a control method for a bidirectional isolated resonant power converter based on a virtual synchronous motor, the AC busbar of the power grid is connected to the AC interface converter through the line impedance Z _ac , the filter resistance R _ac and the LC filter AC side; the DC side of the AC interface converter is connected to the DC/DC converter through the DC capacitor C _dc ; the DC/DC converter is finally connected to the power battery through the voltage stabilizing capacitor C _f and the filter inductance L _f ; the control method According to the structural similarity between the three-phase synchronous motor model and the AC interface converter, the AC interface converter is virtualized as a synchronous motor. The control method includes three parts: active power control, virtual excitation control and voltage and current double closed-loop control. Methods as below:

(1) Active power control: set the number of pole pairs of the virtual synchronous motor to 1, and its torque equation can be expressed as:

Among them, J represents the moment of inertia of the synchronous motor, the unit is kg·m ² , ω _N represents the AC rated angular velocity of the grid, the unit is rad/s; _Pe and P _m are the electromagnetic and mechanical power of the synchronous motor respectively; δ is the power angle of the generator , the unit is rad; ω is the virtual rotor angular frequency of the synchronous motor, the unit is rad/s; k _ω is the AC frequency modulation droop coefficient; the active power control part is mainly used to realize the active power closed-loop and generate mechanical torque; the active power is controlled by the AC side The voltage and current are calculated and expressed as:

P＝u _a i _a +u _b i _b +u _c i _c

where u _a , u _b , _uc are the terminal voltages of the synchronous motor, _ia , i _b , _ic are the terminal currents of the synchronous motor;

(2) Virtual excitation control: In the virtual excitation control part, simulate the excitation control of the generator, control and adjust the AC voltage and reactive power, and make it emit reactive power by adjusting the effective value E of the virtual potential of the virtual synchronous motor model; The effective value E of the virtual potential of the virtual synchronous motor consists of three parts:

One is the part ΔE _Q that reflects the reactive power adjustment, the other is the part ΔE _U that adjusts the terminal voltage of the reactor, which can be equivalent to the automatic excitation regulator of the synchronous motor, and the third is the effective value of the no-load potential of the synchronous motor E ₀ ; then the effective value of the virtual potential of the motor is:

E＝E ₀ +ΔE _Q +ΔE _U

The vector value of the virtual potential of the motor is expressed as:

(3) Double closed-loop control of voltage and current: Based on the KVL law, the electromagnetic equation of the synchronous motor can be expressed as:

为交流侧电压；L和R分别为同步电机的定子电感和电阻，其值分别取交流接口的LC滤波器的滤波电感L_ac和滤波电阻R_ac的值，

为交流接口电流，

为交流母线侧电流；通过电压-电流双闭环控制得到信号e，并将其作为调制波输入SPWM调制，产生交流接口变换器的控制信号，控制交流接口变换器各IGBT管的导通和关断。in

is the AC side voltage; L and R are the stator inductance and resistance of the synchronous motor respectively, and their values are the values of the filter inductance L _ac and the filter resistance R _ac of the LC filter of the AC interface respectively,

is the AC interface current,

It is the current on the AC bus side; the signal e is obtained through the voltage-current double closed-loop control, and it is input as a modulation wave into SPWM modulation to generate the control signal of the AC interface converter, and control the conduction and shutdown of each IGBT tube of the AC interface converter .

The bidirectional power converter is composed of DC/DC stage and DC/AC stage. According to the structural similarity between the three-phase synchronous motor model and the three-phase converter, the DC/AC stage three-phase converter can be Equivalent to a synchronous motor, the entire electric vehicle charging pile is equivalent to a synchronous motor at its grid-connected point. This motor can adaptively respond to the voltage and frequency disturbance of the grid, and provide the necessary inertia and damping for the grid.

Further, the primary side and the secondary side of the DC side of the DC/DC converter are connected by a CLC resonant module, the CLC resonant module includes a capacitor C _{v on the primary side for voltage multiplication operation, and a capacitor C v} on the secondary side A resonant structure composed of a resonant capacitor C _r and a resonant inductance L _r for resonant PWM operation; the 8 switches of the DC/DC converter are used for charging or discharging operations, including M ₁ , M ₂ , M ₃ , M ₄ four IGBT tubes and M ₅ , M ₆ , M ₇ , M ₈ four IGBT tubes on the secondary side.

The DC/DC stage of the interface converter adopts the structure of DAB converter to meet the requirements of high power and wide output voltage range. In order to solve the defect of power loss caused by the large reactive current of the traditional DAB converter, a CLC resonant module is added to realize the zero-voltage turn-on and zero-current turn-off of the switching device of the interface converter, and improve the overall operating efficiency of the converter.

The DC control unit of the bidirectional power converter has 8 switches for charging or discharging operation, unlike the traditional resonant structure, the DC/DC converter is only controlled by PWM, which can avoid the efficiency reduction caused by excessive switching frequency, or Audible noise or no-load regulation problems caused by too low a switching frequency. The bidirectional charger proposed in this paper maintains the structural advantages similar to the DAB converter, and at the same time increases the DC secondary side voltage of the interface converter by adopting a voltage doubler rectification structure, that is, keeping _M3 in the on-state during the discharge operation to twice the original, enabling the converter to achieve bidirectional power flow.

Compared with prior art, the beneficial effect that the present invention has is as follows:

(1) At the same time, the stability requirements of the grid voltage and the battery voltage of the electric vehicle during the charging and discharging process of the electric vehicle are considered, and the control of the AC and DC bus voltage can be realized at the same time during the control;

(2) It can enable the converter to achieve stable control of power, perform stable charging and discharging operations on electric vehicles, and make the system as a whole have higher stability and inertia when there are large fluctuations in the power grid;

(3) At the same time, the DC side has the characteristics of wide output voltage range and high power transmission capability, and through the design of the resonant circuit, zero-voltage turn-on and zero-current turn-off can be well realized, which can effectively improve the conversion efficiency of the converter and meet the requirements of The demand for efficient charging and discharging of electric vehicles;

(4) In addition, the change of the charge and discharge state can be effectively controlled through the change of a single switch, which can be easily operated in practical engineering applications and improve safety.

Description of drawings

Figure 1 is a topology diagram of a bidirectional power converter.

Figure 2 is a control block diagram of a bidirectional power converter based on virtual synchronous motor control on the AC side

Fig. 3 is a control block diagram of the DC side bidirectional power converter.

Fig. 4 is a mode diagram of a DC side bidirectional power converter under charging operation.

Fig. 5 is a model diagram of a DC side bidirectional power converter under discharge operation.

Fig. 6 is a waveform change diagram of the DC side bidirectional power converter under charging and discharging operations.

Detailed ways

Specific embodiments of the present invention will be described below in conjunction with the accompanying drawings.

This embodiment is mainly used as a bidirectional power converter for charging and discharging motors of electric vehicles. The structural topology of the bidirectional interface converter is shown in FIG. 1 . The AC busbar of the power grid is connected to the AC side of the interface converter through the line impedance Z _ac and the LC filter (L _ac is the filter inductance, C _ac is the filter capacitor, and R _ac is the filter resistor); the DC side of the interface converter is connected through the DC capacitor C _dc DC/DC converter, the primary side and the secondary side of the DC side of the interface converter are connected by a CLC resonant module, and then through the voltage stabilizing capacitor C _f and the filter inductance L _f , and finally connected to the power battery electric vehicle battery.

The AC side adopts a power control method based on a virtual synchronous motor. Its control block diagram is shown in Figure 2. The AC control unit includes active power control, virtual excitation control and voltage and current double closed-loop control. The control methods of each part are as follows:

(1) Active power control: set the number of pole pairs of the virtual synchronous motor to 1, and its torque equation can be expressed as:

Among them, J represents the moment of inertia of the synchronous motor, the unit is kg·m ² , ω _N represents the AC rated angular velocity of the grid, the unit is rad/s; T _e , T _m and T _d are the electromagnetic torque, mechanical torque and damping of the synchronous motor, respectively Torque; D is the damping coefficient, which is equal to the AC primary frequency modulation droop coefficient k _ω when the effect of the damping winding is not considered; δ is the power angle of the generator, the unit is rad; ω is the virtual rotor angular frequency of the synchronous motor, the unit is rad/s . Due to the existence of the moment of inertia J, the charging/discharging machine exhibits the ability of mechanical inertia when the grid voltage fluctuates. When the AC droop coefficient is selected, the increase of the moment of inertia J can increase the inertia of the AC frequency, but if J is too large, it will reduce the stability of the system, so it should not be too large. In addition, the increase of the damping coefficient D can increase the inertia of the whole system, and the same too large D will affect the stability of the system. The electromagnetic torque of the motor can be obtained from the virtual synchronous motor potential _eabc and the output current _iabc , then the relationship between the electromagnetic torque and the electromagnetic power P _e output by the virtual synchronous generator can be expressed as:

T _e ＝P _e /ω＝(e _a i _a +e _b i _b +e _c i _c )/ω

Under the condition of constant power load with power P, the rated mechanical torque T ₀ of the synchronous motor is inversely correlated with the frequency ω _N of the power supply network (that is, the AC rated angular velocity of the power grid), and is affected by the frequency interference and physical damping of the power supply network After that, there will be a change, the frequency and the speed are positively correlated, and thus bring about the increase or decrease of the mechanical damping with the same amplitude, which is the response to the change of the grid frequency. In view of this, we can adjust the active power command in the grid-side converter interface by adjusting the mechanical torque T _m of the virtual synchronous motor, _which is the difference between the fixed torque T ₀ and the frequency deviation feedback command △T. Therefore, the mechanical torque can be expressed as:

T _m = T ₀ +ΔT = (PP _ref )/ω _N

Among them, _Pref is the selected reference power, and the active power control part is mainly used to realize active power closed-loop and generate mechanical torque. The active power is calculated from the AC side voltage and current, which can be expressed as:

P＝u _a i _a +u _b i _b +u _c i _c

Integrate the obtained mechanical torque minus the electromagnetic torque and damping torque and divide it by the moment of inertia J to obtain the virtual rotor angular frequency ω of the synchronous motor. Continue to integrate the rotor angular frequency to obtain the virtual The power angle δ of the synchronous generator.

(2) Virtual excitation control: In the virtual excitation control part, the excitation control of the generator is simulated, the AC voltage and reactive power are controlled and adjusted, and the virtual potential effective value E of the virtual synchronous motor model is adjusted to make it emit reactive power. The effective value E of the virtual potential of the virtual synchronous motor consists of three parts.

One is the part ΔE _Q that reflects the reactive power adjustment, the other is the part ΔE _U that adjusts the terminal voltage of the reactor, which can be equivalent to the automatic excitation regulator of the synchronous motor, and the third is the effective value E of the no-load potential of the motor ₀ .

Where k _q is the reactive power-voltage droop coefficient, Q _ref and Q are the instantaneous reactive power reference value and actual value output by the AC interface terminal, respectively. k _v is the voltage adjustment coefficient; U _ref and U are the reference value and real value of the effective value of the terminal line voltage of the grid-connected inverter, respectively. Then the effective value of the virtual potential of the motor is:

E＝E ₀ +ΔE _Q +ΔE _U

The vector value of the virtual potential of the motor is expressed as:

(3) Double closed-loop control of voltage and current: Based on KVL law and KCL law, the electromagnetic equation of synchronous motor can be expressed as:

为交流侧电压；L和R分别为同步电机的定子电感和电阻。需要特别注意的是，这里的定子电感L和电阻R与交流接口的滤波电感L_ac和滤波电阻R_ac相对应，

为交流接口电流，

为交流母线侧电流。有功功率控制模拟同步电机的机械运动方程，调节有功功率，该控制环节输出虚拟电势

的频率和相位；虚拟励磁控制模拟同步电机励磁调节，控制无功功率，输出

的有效值。in

is the AC side voltage; L and R are the stator inductance and resistance of the synchronous motor, respectively. It should be noted that the stator inductance L and resistance R here correspond to the filter inductance L _ac and filter resistance R _ac of the AC interface,

is the AC interface current,

is the AC bus side current. The active power control simulates the mechanical motion equation of the synchronous motor to adjust the active power, and the control link outputs the virtual potential

frequency and phase; virtual excitation control simulates synchronous motor excitation regulation, controls reactive power, output

valid value for .

In order to simplify the control, the mathematical model of the interface converter is converted from the abc three-phase stationary coordinate system to the two-phase rotating d-q coordinate system. The transformation matrix is as follows:

In the formula, θ=ωt+θ ₀ represents the angle between the d-axis and the a-axis, and θ ₀ is the angle when t=0.

After coordinate conversion, the mathematical model of the interface converter in the d-q coordinate system is obtained, thus, the three-phase AC component in the interface converter becomes a two-phase DC component. Then there are:

的d、q轴分量；e_d、e_q分别表示交流电压

的d、q轴分量；i_d和i_q分别表示交流电流

的d、q轴分量；i_ld和i_lq分别表示交流电流

的d、q轴分量。In the formula, u _d and u _q respectively represent AC voltage

The d and q axis components of ; e _d and e _q respectively represent the AC voltage

The d, q axis components of ; i _d and i _q respectively represent the alternating current

The d, q axis components of ; i _ld and i _lq respectively represent the alternating current

d, q axis components.

According to the above formula, as shown in Figure 2(b), the coupling items in the voltage equation are ωLi _q and ωLi _d , and the coupling items in the current equation are ωCu _q and ωCu _d . By introducing a negative coupling item in the control, the coupling effect between the d and q axes is eliminated; the interface converter is used to track the reference signal without static error by using PI control. In the double closed loop shown in Figure 2(b), the controlled variables of the voltage loop and the current loop are voltage and current respectively, and the transfer functions when PI control is used are as follows:

Among them, u _d-ref and u _q-ref are the d and q axis components of the given AC voltage reference value respectively, i _d-ref and i _q-ref are the d and q axis components of the given AC voltage reference value respectively, e _d-ref and e _q-ref are the d and q axis components of a given AC voltage reference value respectively. The last item i _d , i _q , u _d and u _q are all feed-forward items, which are the d and q axis components of the voltage and current measured values, which can accelerate the response of the controller. Both G _u (s) and G _i (s) are PI regulators, and their transfer functions are:

In the formula, K _uP and K _uI are the proportional and integral coefficients of the voltage loop respectively. K _iP , K _iI are the proportional and integral coefficients of the current loop respectively.

Finally, the signal e is obtained through voltage-current double closed-loop control, and it is input as a modulation wave into SPWM modulation to generate a control signal for the interface converter and control the on and off of IGBT tubes VT ₁ ~ VT ₆ .

The DC control unit of the bidirectional power converter has 8 switches for charging or discharging, and the specific operations are shown as M ₁ -M ₈ in FIG. 3 . The resonant structure consisting of resonant capacitor C _r and resonant inductance L _r is located on the secondary side for resonant PWM operation, and the capacitor C _v on the primary side is used for voltage multiplication operation. Unlike the traditional resonant structure, the converter is only controlled by PWM, which can avoid the efficiency reduction caused by too high switching frequency, or the audible noise or no-load regulation problem caused by too low switching frequency. The bidirectional charger proposed in this application maintains the structural advantages similar to the DAB converter, and at the same time adopts the voltage doubler rectification structure, that is, keeps _M3 in the conducting state during the discharge operation, so that the DC secondary side voltage of the interface converter doubled, enabling the converter to achieve bidirectional power flow. Among them, V _ref represents the reference voltage of the secondary side of the DC interface, and I _ref represents the reference value of the current of the secondary side.

In the charging and discharging phases, the mode analysis diagrams of Fig. 4 and Fig. 5 are obtained respectively.

Figure 4 (a, b, and c in Figure 4 represent modes 1, 2, and 3, respectively) is a mode diagram of the DC-side bidirectional power converter under charging operation. In charging operation, the power flow is controlled by M ₁ ~ M ₄ , and M ₅ ~M ₈ diodes are used for full bridge rectification. Since the value of C _v is tens of microfarads, it has a large enough value in this mode and can be regarded as a DC coupling capacitor without affecting the charging operation. The mode diagram and its equivalent circuit during charging are shown in Fig. 4. Since the drain-source capacitance of the switching device is usually relatively small, the drain-source capacitance C _ds of the switch can be ignored. At the same time, assuming that the resonant capacitor voltage v _cr does not exceed the battery voltage V _batt , M ₃ is initially in a conduction state. During the whole charging process, the switching tube and soft switching waveforms are shown in Figure 6(a).

Mode 1 (t ₀ ≤t<t ₁ ): When M ₁ is turned on at t ₀ , the primary current _ip flows through M ₁ , M ₃ and the primary side of the converter. The secondary side current i _s increases from zero and flows through C _r , L _r , M ₅ , M ₇ and the secondary side of the converter. v _cr and i _s can be deduced from the equivalent circuit of mode 1 in Fig. 4(a):

in:

V _crf is the peak voltage when v _cr charges and runs, and the primary current _ip refers to the sum of the primary side current _is and the magnetizing current _im . When _M1 is turned off, the primary current is used to charge and discharge the _Cds of _M1 and _M2 . If the C _ds of _M2 is fully discharged before the end of mode 1, zero-voltage switching (ZVS) of _M2 can be achieved. And the ZVS condition of _M2 is easy to satisfy, because ZVS uses the peak value of i _p . This operation is indicated by the light colored line in Fig. 4(a).

Mode 2 (t ₁ ≤t<t ₂ ): when M ₂ is turned on, mode 2 starts. The primary current _ip flows through M ₃ , M ₂ and the primary side of the converter. The secondary current i _s maintains the same current path as the previous state until i _s decreases to zero. From the equivalent circuit of mode 2 in Figure 4(b), v _cr and i _s can be expressed as:

v _cr (t)＝-V _batt -(V _cr (t ₁ )+V _batt )cosω _r (tt ₁ )+Z _r i _Lr (t ₁ )sinω _r (tt ₁ )

Assuming that the dead time between the upper and lower gate signals is short enough to be ignored, i _Lr (t ₁ ) can be obtained from the above formula. as follows:

Where D _f is the duty cycle of M ₁ (or M ₄ ) during charging. The duration of T _2Mf can be deduced from the above formula

Mode 3 (t ₂ ≤t<t ₃ ): after mode 2, only the magnetizing current i _m circulates through M ₃ and M ₂ . In mode 3, the resonant capacitor voltage v _cr remains at v _cr (t ₂ ), the peak value of the resonant capacitor defined by V _crf . V _crf can be obtained:

V _crf should not exceed V _batt to prevent abnormal conduction of body diodes of M ₅ and M ₇ and ensure normal operation. Only the C _ds charging and discharging of _M3 and _M4 adopt peak magnetizing current. If the C _ds of _M4 is fully discharged before the end of mode 3, the ZVS of _M4 can be achieved. Unlike the ZVS condition of _M2 , the ZVS of _M4 is not easy, because ZVS only uses the peak value of the magnetizing current and is affected by the load condition. The operation of the next half-cycle consisting of modes 4-6 is the same as that of the previous half-cycle.

Fig. 5 is a model diagram of a DC side bidirectional power converter under discharge operation. The power flow is controlled by M ₅ ~ M ₈ during the discharge process. In order to improve the voltage gain, a rectifier with diodes _M1 and _M2 is used as a voltage multiplier rectifier, so that _M3 is in a conducting state. Capacitor C _v is used as a power supply capacitor to provide power. Assume that the impedance of the magnetizing inductance of the secondary side of the converter is ω _s L _m /n ² ω, (L _m is the magnetizing inductance of the transformer in the resonant module in Figure 1) and this impedance is related to the impedance of the resonant tank ω _s L _m +1/( ω _s C _r ) is large enough. Here, the unit of the switching frequency ω _s (referring to M ₁ -M ₈ ) is rad/second. Then in the whole charging process, the switching tube and the soft switching waveform are shown in Fig. 6(b).

Mode 1 (t ₀ ≤t<t ₁ ): When M8 is turned on at t ₀ , the secondary current passes through L _r , C _r , M ₈ , M ₆ and the secondary side of the converter. The primary current _ip increases from zero and passes through M ₂ , M ₃ , C _v and the primary side of the converter. From Fig. 5(a), v _cr and the primary current and the primary current ni _p on the secondary side can be approximately derived as:

V _crr is the peak voltage of v _cr during the discharge process, and its expression is shown in the last formula. The secondary current is equal to the sum of ni _p and ni _m . When M ₈ is closed, i _s is used to make M ₇ reach ZVS state.

Mode 2 (t ₁ ≤t<t ₂ ): When M ₇ is turned on, Mode 2 starts. The power flow is shown in Fig. 5(b), and _ip starts to decrease. From the equivalent circuit of mode 2, v _cr and ni _p can be obtained as:

Using similar assumptions to Mode 2 in charging operation, ni _p (t ₁ ) and v _cr (t ₁ ) can be derived. Their derivation is as follows:

In the formula, D _r is the duty cycle of M ₅ (or M ₈ ) in the discharge operation. Mode 2 continues until i _p decreases to zero, the duration of mode 2 in discharge mode T _2Mr is

Mode 3 (t ₂ ≤t<t ₃ ): after mode 2, only the magnetizing current of the secondary side ni _m passes through M ₇ and M ₆ . At the same time, v _cr (t ₂ ) can be deduced as follows:

V _crr is the peak voltage of v _crr during discharge. When _M6 is turned off, the peak magnetizing current on the secondary side charges and discharges the drain-source capacitance C _ds of _M5 and _M6 . The first half cycle consisting of modes 1 to 3 is the charging cycle of the charging capacitor C _v passing through the M ₂ and M ₃ channels, and the next half cycle consisting of modes 4 to 6 is the charging capacitor C _v and M ₁ , M ₃ charge cycle. Except for the rectification operation part, the two half-cycle operations of modes 1-3 and modes 4-6 are basically the same.

Claims (3)

1. The control method of the two-way isolated resonant power converter based on the virtual synchronous motor, the AC busbar of the power grid is connected to the AC side of the AC interface converter through the line impedance Z _ac , the filter resistor R _ac and the LC filter; the DC of the AC interface converter The side is connected to the DC/DC converter through the DC capacitor C _dc ; the DC/DC converter is finally connected to the power battery through the voltage stabilizing capacitor C _f and the filter inductance L _f ; it is characterized in that the control method is based on the three-phase synchronous motor Based on the similarity between the model and the AC interface converter in structure, the AC interface converter is virtualized as a synchronous motor. The control method includes three parts: active power control, virtual excitation control and voltage and current double closed-loop control. The control methods of each part are as follows: (1) Active power control: set the number of pole pairs of the virtual synchronous motor to 1, and its torque equation can be expressed as:

Among them, J represents the moment of inertia of the synchronous motor, the unit is kg·m ² , ω _N represents the AC rated angular velocity of the grid, the unit is rad/s; _Pe and P _m are the electromagnetic and mechanical power of the synchronous motor respectively; δ is the power angle of the generator , the unit is rad; ω is the virtual rotor angular frequency of the synchronous motor, the unit is rad/s; k _ω is the AC frequency modulation droop coefficient; the active power control part is mainly used to realize the active power closed-loop and generate mechanical torque; the active power is controlled by the AC side The voltage and current are calculated and expressed as: P＝u _a i _a +u _b i _b +u _c i _c where u _a , u _b , _uc are the terminal voltages of the synchronous motor, _ia , i _b , _ic are the terminal currents of the synchronous motor; (2) Virtual excitation control: In the virtual excitation control part, simulate the excitation control of the generator, control and adjust the AC voltage and reactive power, and make it emit reactive power by adjusting the effective value E of the virtual potential of the virtual synchronous motor model; The effective value E of the virtual potential of the virtual synchronous motor consists of three parts: One is the part ΔE _Q that reflects the reactive power adjustment, the other is the part ΔE _U that adjusts the terminal voltage of the reactor, which can be equivalent to the automatic excitation regulator of the synchronous motor, and the third is the effective value of the no-load potential of the synchronous motor E ₀ ; then the effective value of the virtual potential of the motor is: E＝E ₀ +ΔE _Q +ΔE _U The vector value of the virtual potential of the motor is expressed as:

(3) Double closed-loop control of voltage and current: Based on the KVL law, the electromagnetic equation of the synchronous motor can be expressed as:

in

is the AC side voltage; L and R are the stator inductance and resistance of the synchronous motor respectively, and their values are the values of the filter inductance L _ac and the filter resistance R _ac of the LC filter of the AC interface respectively,

is the AC interface current,

is the current on the AC bus side, C is the value of C _ac in the filter capacitor of the LC filter; the signal e is obtained through voltage-current double closed-loop control, and it is input as a modulation wave into SPWM modulation to generate the control signal of the AC interface converter and control The turn-on and turn-off of each IGBT tube of the AC interface converter. 2. The bidirectional isolation type resonant power converter control method based on virtual synchronous motor as claimed in claim 1, wherein the primary side and the secondary side of the DC side of the DC/DC converter are connected by a CLC resonant module, so The CLC resonant module includes a capacitor C _v on the primary side for voltage multiplication operation, and a resonant structure composed of a resonant capacitor C _r and a resonant inductance L _r on the secondary side for resonant PWM operation; the DC/DC The converter has 8 switches for charging or discharging operations, including four IGBT tubes M ₁ , M ₂ , M ₃ , and M ₄ on the primary side and four IGBT tubes on the secondary side M ₅ , M ₆ , M ₇ , and M ₈ Tube. 3. The control method of bidirectional isolated resonant power converter based on virtual synchronous motor as claimed in claim 2, characterized in that, (1) in the charging operation, the power flow is controlled by M ₁ ~ M ₄ , M ₅ ~ M ₈ diodes are used for full-bridge rectification; assuming that the resonant capacitor voltage v _cr does not exceed the battery voltage V _batt , M ₃ is in the conduction state at the beginning; a complete charging process is divided into 6 modes according to the time sequence: Mode 1 (t ₀ ≤t<t ₁ ): when M ₁ is turned on at t ₀ , the primary current i _p flows through M ₁ , M ₃ and the primary side of the converter; the secondary side current i _s increases from zero, And flow through C _r , L _r , M ₅ , M ₇ and the secondary side of the converter; v _cr and i _s can be deduced from the equivalent circuit of mode 1:

in:

V _crf is the peak voltage when v _cr is charging and running, V _dc is the voltage across the DC capacitor C _dc ; the primary current i _p is the sum of the primary side current i _s and the magnetizing current _im ; n is the DC/DC converter The turns ratio of the primary and secondary sides; when M1 is turned off, the primary current is used to charge and discharge the drain-source capacitance C _ds of _M1 and _M2 ; if the drain-source capacitance C _ds of M2 is fully discharged before the end of mode 1 , then the zero voltage switching of M ₂ can be realized; Mode 2 (t ₁ ≤t<t ₂ ): when M ₂ is turned on, mode 2 starts; the primary current i _p flows through M ₃ , M ₂ and the primary side of the converter; the secondary current i _s keeps the same current as mode 1 path until i _s decreases to zero; expressed by the equivalent circuit of mode 2, v _cr and i _s as: v _cr (t)＝-V _batt -(V _cr (t ₁ )+V _batt )cosω _r (tt ₁ )+Z _r i _Lr (t ₁ )sinω _r (tt ₁ )

From the above formula, i _Lr (t ₁ ) and v _cr (t ₁ ) can be obtained as follows:

In the formula, D _f is the duty cycle of M ₁ (or M ₄ ) during charging, and T _s is the switching period of the driving signal. From the above formula, the duration T _2Mf of mode 2 can be deduced:

Mode 3 (t ₂ ≤t<t ₃ ): after mode 2, only the magnetizing current _im circulates through M ₃ and M ₂ ; in mode 3, the resonant capacitor voltage v _cr remains at v _cr (t ₂ ), as The peak value of the resonant capacitance defined by V _crf ; V _crf can be obtained as:

V _crf should not exceed V _batt to prevent the abnormal conduction of the body diodes of M ₅ and M ₇ and ensure normal operation; only the C _ds charge and discharge of M ₃ and M ₄ use the peak magnetizing current; if the drain-source capacitance C of M _{4 ds} is fully discharged before the end of mode 3, then the zero-voltage switching of _M4 can be realized; the operation of the next half cycle composed of modes 4 to 6 is the same as that of the previous half cycle; (2) During the discharge process, the power flow is controlled by M ₅ ~ M ₈ ; in order to increase the voltage gain, the rectifier of diode M ₁ and M ₂ is used as voltage multiplier rectification, so that M ₃ is in the conduction state; capacitor C _v is used as the power supply The power supply capacitor; a complete discharge process is divided into 6 modes in chronological order: Mode 1 (t ₀ ≤t<t ₁ ): when M8 is turned on at t ₀ , the secondary current passes through L _r , C _r , M ₈ , M ₆ and the secondary side of the DC/DC converter; the primary current i _p Starting from zero, and passing through M ₂ , M ₃ , C _v and the primary side of the DC/DC converter; v _cr , the primary current _ip and the primary current ni _p of the secondary side can be approximately derived as:

V _crr is the peak voltage of v _crr during the discharge process; the secondary current is equal to the sum of ni _p and magnetizing current ni _m ; when M ₈ is closed, _{is is} used to make M ₇ reach the zero-voltage switching state; Mode 2 (t ₁ ≤t<t ₂ ): when M ₇ is turned on, mode 2 starts and i _p begins to decrease; from the equivalent circuit of mode 2, v _cr and ni _p can be obtained as:

Using similar assumptions as in charge operation mode 2, ni _p (t ₁ ) and v _cr (t ₁ ) can be derived here:

In the formula, D _r is the duty cycle of _M5 or _M8 in the discharge operation, mode 2 continues until i _p decreases to zero, and the duration T _2Mr of mode 2 in the discharge mode is

Mode 3 (t ₂ ≤t<t ₃ ): After mode 2, only the magnetizing current ni _m on the secondary side passes through M ₇ and M ₆ , and v _cr (t ₂ ) can be derived as follows:

v _crr is the peak voltage of v _cr during discharge; when M ₆ is closed, the peak magnetizing current on the secondary side charges and discharges the C _ds of M ₅ and M ₆ ; the first one consists of modes 1~3 The half cycle is the charging cycle of the charging capacitor C _v through the _M2 and _M3 channels, and the next half cycle consisting of modes 4 to 6 is the charging cycle of the charging capacitor C _v and _M1 and _M3 ; except for the rectification operation part , the two half-cycle operations of modes 1-3 and modes 4-6 during the discharge process are the same.

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