CN109995248A - A sampling method to improve output stability of flyback resonant switching power supply - Google Patents

Abstract

A sampling method for improving the output stability of a flyback resonant switching power supply. The sampling module collects the auxiliary winding voltage Vsense of the primary side and compares it with the feedback voltage Vref in the analog form of this cycle. The time T ₁ of Vref<Vsense, Vref>Vsense time T ₂ , Tr=T ₁ -T ₂ , the size of Tr is the clock cycle that is different between T ₁ and T ₂ , according to the size of Tr, calculate the digital error signal err of the current cycle, and adjust the digital feedback voltage VREF After the analog value Vref is obtained by the digital-to-analog converter, it is compared with the voltage signal Vsense of the auxiliary winding to form a closed-loop negative feedback to adjust the value of the feedback voltage Vref, err and VREF are input to the control module, and the control module generates a duty cycle control The signal duty adjusts the on-time and off-time of the primary switch tube to stabilize the output of the switching power supply.

Description

A sampling method to improve output stability of flyback resonant switching power supply

technical field

The invention relates to a flyback resonance type switching power supply, in particular to a sampling method for improving the output stability of the flyback resonance type switching power supply.

Background technique

In the PSR flyback converter system, the sampling module is a bridge connecting the peripheral topology circuit and the internal control circuit, and its importance is irreplaceable. The accuracy of the sampling method is directly related to the constant current and constant voltage accuracy of the entire flyback converter and the stability of the system, and the real-time performance of the sampling method is directly related to the transient characteristics of the entire flyback converter. In a flyback power supply, the primary and secondary are in an isolated state. If you want to detect the output, there are two methods: primary side feedback and secondary side feedback. The principle of the primary side feedback is to detect the information of the load change by accurately sampling the voltage change of the auxiliary winding (N _aux ). Compared with the traditional secondary-side feedback optocoupler plus 431 structure, the biggest advantage of primary-side feedback is that these two chips and a group of components working with them are omitted, which saves space on the system board. , reducing the cost and improving the reliability of the system.

In the primary side feedback, it is necessary to detect the load change information by accurately sampling the voltage change of the auxiliary winding (N _aux ), but due to the existence of other components in the secondary winding and the auxiliary winding, not all auxiliary windings are suitable for detecting the output voltage. , so we can choose to detect representative points in it.

In the topology of the primary-side feedback flyback resonant converter, due to the irregular output voltage waveform, traditional sampling algorithms, such as the inflection point acquisition method, are not applicable. In order to sample the value of the measured output voltage, a new sampling method must be used.

SUMMARY OF THE INVENTION

In order to overcome the limitations and deficiencies of the prior art, the present invention proposes a sampling method for improving the output stability of a flyback resonant switching power supply. Following the change of the output voltage V _o , it is input to the control module to stabilize the output of the switching power supply. In the present invention, the sampling accuracy increases with the increase of the load efficiency, and the effect is the best when the load is fully loaded. The present invention can achieve relatively high measurement accuracy while the components used are small in volume.

In order to achieve the above object, the technical solution adopted in the present invention is: a sampling method for improving the output stability of a flyback resonant switching power supply, based on a MHz-level primary-side feedback flyback resonant converter topology, characterized in that: under full load conditions The change of the primary auxiliary winding voltage following the output voltage V _o is collected by the sampling module and input to the control module. The control module generates a duty cycle control signal duty to adjust the on time and off time of the primary side switch tube to stabilize the switching power supply. output;

The voltage Vs on the secondary winding of the primary-side feedback flyback resonant converter is reflected to the voltage Va of the primary auxiliary winding in the ratio of V _s :V _{a =} N _s :N _aus , where N _s and _Naus are the secondary windings respectively. The number of turns of the side winding and the auxiliary winding, and the proportional relationship between the voltage Vsense across the voltage dividing resistor R ₂ and the voltage Va on the primary auxiliary winding is V _sense :V _{a =} R ₂ :(R ₁ +R ₂ ), R _1. R ₂ is the voltage divider resistor on the primary auxiliary winding. The sampling module collects the voltage waveform Vsense of the auxiliary winding through the voltage divider resistor R _2. Since Vsense is an analog voltage, it shows an approximate sinusoidal ripple, while Now the general control module deals with digital signals, so we need a constant feedback voltage to reflect the size of Vsense to the greatest extent. Vref and VREF are the analog and digital forms of the feedback voltage, respectively, and err is the value of each cycle. The error signal is a digital quantity. VREF is converted into Vref through a digital-to-analog converter. Different forms are used according to the needs. In order to make the value of the feedback voltage closer to Vsense, according to the characteristics of Vsense sinusoidal fluctuations, when collecting the Vsense ripple midpoint The voltage at the point is used as the initial value of the feedback voltage Vref in the analog form. The feedback voltage Vref of the analog quantity is directly compared with the voltage signal Vsense of the voltage dividing resistor R ₂ of the auxiliary winding on the primary side, and the feedback voltage Vref of the analog quantity is obtained through digital-to-analog conversion. The digital feedback voltage VREF, and then the digital feedback voltage VREF is added and subtracted in the calculation module to obtain a feedback voltage that reflects the size of Vsense to the greatest extent. The feedback voltage VREF and the digital error signal err are used to calculate the output voltage V _o and the duty cycle control signal duty of the primary switch.

The sampling module includes a comparator, a DAC and a calculation module, and the calculation module is written by code and can realize the functions of a counter and an adder. Compare the voltage signal Vsense of the voltage dividing resistor R ₂ of the primary auxiliary winding with the feedback voltage Vref in the analog form obtained in the previous cycle through the comparator, and output the comparison result to the calculation module. The voltage signal Vsense of ₂ shows an approximate sinusoidal ripple, and the feedback voltage Vref in the analog form obtained in each cycle is a fixed value. Each cycle can be divided into two parts according to the point where they are equal. Immediately, Vref will continue to be less than Vsense until it is equal, and after the other part is equal, Vref will continue to be greater than _Vsense until it is equal. After the feedback voltage Vref in the analog form obtained in one cycle is equal, the time T ₁ of Vref<Vsense, Vref> the time T ₂ of Vsense, that is, the size of T ₁ is the time of Vref<Vsense. collected by the counter The clock period, the size of T ₂ is the clock period during the period when Vref>Vsense. collected by the counter, the time difference between the two is Tr=T ₁ -T ₂ , and the size of Tr is the clock period that differs between T ₁ and T ₂ . According to the size of Tr, the digital error signal err of the current cycle is calculated, and the size of the digital feedback voltage VREF is adjusted. The digital VREF obtained at this time is input to the digital-to-analog converter, and after the analog Vref is obtained, it is combined with the auxiliary winding. The voltage signal Vsense is compared to form a closed-loop negative feedback to adjust the value of the feedback voltage Vref.

The sampling process of the sampling module is as follows:

Compared with the voltage signal Vsense of the auxiliary winding, Vref has the following three states: Vref<Vsense, Vref=Vsense, Vref>Vsense. The time T ₁ of Vref<Vsense, the size of T ₁ is Vref<Vsense collected by the counter. The clock period of this period of time, the time T ₂ when Vref>Vsense, the size of T ₂ is the clock period of this period of time when Vref>Vsense. collected by the counter. The time difference between the two is Tr=T ₁ -T ₂ , and the size of Tr is the clock period that is different between T ₁ and T ₂ . The size of the error signal err can be judged according to the size of Tr, and their adjustment relationship is as follows;

When Tr<-6, err=-3;

When -6<=Tr<=-4, err=-2;

When -3<=Tr<=-1, err=-1;

When Tr=0, err=0;

When 1<=Tr<=3, err=1;

When 4<=Tr<=6, err=2;

When Tr>6, err=3

According to the value of the error signal err, on the basis of the digital feedback voltage VREF in this cycle, adjust the size of the digital feedback voltage VREF in the next cycle. The next cycle VREF is equal to the current cycle VREF plus the error signal err, that is, in the next cycle VREF=VREF+err.

The advantages and remarkable effects of the present invention:

1. It can sample and process the irregular waveform of the primary-side feedback flyback resonant converter;

2. It can make adjustments according to the change of voltage in real time. The components used are low in cost and occupy a small space, which can ensure the scale of the device.

3. The feedback voltage obtained has two forms, digital and analog, and the voltage of different modes can be selected according to the actual situation.

Description of drawings

Fig. 1 is the topology model of the primary-side feedback flyback resonant converter applicable to the present invention;

Figure 2 is the current waveform on the freewheeling diode of the secondary winding when the converter is stably operating in the current continuous working mode under heavy load conditions;

Figure 3 is the waveform of the voltage Vsense across the voltage dividing resistor R ₂ of the primary auxiliary winding;

Fig. 4 is sampling module flow chart in the present invention;

Fig. 5 is the adjustment process of the sampling module in the present invention.

Detailed ways

Referring to FIG. 1 , the topology used in the present invention is a MHz-level primary-side feedback flyback resonant converter topology. The invention is a sampling algorithm for improving the output stability of switching power supply, which is characterized in that: based on the topology of the MHz-level primary-side feedback flyback resonant converter, the sampling follows the change of V _o under heavy load conditions, and is input to the control module to stabilize The output of the switching power supply.

The voltage Vs on the secondary winding of the primary-side feedback flyback resonant converter is reflected to the voltage Va of the primary auxiliary winding in the ratio of V _s :V _{a =} N _s :N _aus , where N _s and _Naus are the secondary windings respectively. The number of turns of the side winding and the auxiliary winding, and the proportional relationship between the voltage Vsense across the voltage dividing resistor R ₂ and the voltage Va on the primary auxiliary winding is V _sense :V _{a =} R ₂ :(R ₁ +R ₂ ), R _1. R ₂ is the voltage divider resistor on the primary auxiliary winding. The sampling module collects the voltage waveform Vsense of the auxiliary winding through the voltage divider resistor R _2. Since Vsense is an analog voltage, it shows an approximate sinusoidal ripple, while Now the general control module deals with digital signals, so we need a feedback voltage of constant size, which can reflect the size of Vsense to the greatest extent. Vref and VREF are the analog and digital forms of the feedback voltage, respectively, and err is each cycle. The error signal is a digital quantity. VREF can be converted into Vref through a digital-to-analog converter, and different forms can be used as required. In order to make the value of the feedback voltage closer to Vsense, the voltage at the midpoint of the Vsense ripple can be collected as the initial value of the feedback voltage Vref in analog form according to the characteristics of the sinusoidal fluctuation of Vsense. The analog feedback voltage Vref can be directly compared with the voltage signal Vsense of the primary auxiliary winding voltage divider resistor R ₂ , and the analog feedback voltage Vref can be converted to a digital feedback voltage VREF through digital-to-analog conversion. The feedback voltage VREF is added and subtracted in the calculation module to obtain a feedback voltage that can reflect the magnitude of Vsense to the greatest extent. Therefore, the digital feedback voltage VREF and the digital error signal err required by the control module are obtained through the processing of the calculation module, so as to calculate the output voltage V _o and the duty cycle control signal duty of the primary switch.

The control modules in Figures 1 and 4 can adopt the prior art, and include a multi-mode judgment module, a digital PID module, a control voltage module, a PWM drive module, a comparator and a digital-to-analog converter. The sampling module collects information on the resistance voltage division on the primary auxiliary winding; the digital PID module calculates the control voltage compensation V _PI at this time according to the error signal err output by the sampling module, and outputs it to the multi-mode judgment module and the control voltage module And the PWM drive module; the control voltage module outputs the peak voltage Vpeak required by the closed-loop control according to the control voltage compensation V _PI and the multi-mode judgment module output current state quantity state, because Vpeak is a digital quantity, it cannot be directly connected with the analog quantity. The voltage Vp of the sampling resistor on the primary side is compared, so after Vpeak passes through the digital-to-analog converter DAC, it is converted into an analog quantity and Vp is compared with the comparator Comp, and the comparison result Comp_Ip is output to the PWM drive module; the multi-mode judgment module is based on the control voltage. The compensation value V _PI and the digital feedback voltage VREF output by the sampling module are used to judge the mode, and output the current state value state; the PWM drive module is based on the control voltage compensation value V _PI output by the digital PID module, and the comparison result Comp_Ip output by the comparator Comp The steady-state Vpeak provided by the control voltage module and the current state quantity state output by the multi-mode judgment module are obtained and output the duty cycle signal of the switch tube and the synchronous rectifier tube; and the PWM waveform that regulates the main switch tube of the switching power supply is obtained through the drive module. , to achieve constant voltage output.

According to whether the current Is of the secondary winding in the primary-side feedback flyback converter drops to 0 in one cycle, the working modes of the converter can be divided into continuous current mode (CCM), discontinuous current mode (DCM) and critical working mode (CRM), when working in CCM mode, after the switch is turned off, I _s will be The slope gradually decreases to a certain value, and the fall time is represented by _Tr . During _Tr , the voltage on the secondary winding is V _s =V _o +V _d , where _Vo is the output voltage, and V _d is the voltage drop of the freewheeling diode on the secondary winding. The diode is turned on, which can be equivalent to a resistor connected in series with an ideal power supply, that is, V _d =V _f +I _d *R _d , where V _f is the conduction voltage drop of the freewheeling diode on the secondary winding, and I _d is the secondary winding current on the freewheeling diode.

The waveform of the current I _d of the freewheeling diode on the secondary winding is shown in Figure 2. In the primary feedback flyback resonant converter topology, the current flowing through the inductor decreases linearly when the voltage on the transformer is constant and the leakage inductance L of the primary winding of the transformer The resonance effect of _res and the parasitic capacitance C _oss of the switch tube, the waveform of I _d is the slope of The linear decreasing superposition of a frequency is oscillation.

According to the above analysis, it can be seen that:

The waveform of Vsense is shown in Figure 3. Since the conduction voltage drop V _f of the freewheeling diode on the secondary winding is a fixed value and V _f <<V _o , the conduction voltage of the diode can be ignored. R _d represents the on-resistance of the diode. In low-power primary-side feedback flyback resonant converters, in order to ensure efficiency, the freewheeling diode generally chooses a diode with low on-resistance, that is, R _d is very small, and I _d * R _d The slope will be very small and the voltage difference between adjacent midpoints will approach zero. From this, it can be inferred that the magnitude of V _d can be ignored, or compensation is performed by the compensation module, and the compensation part is not described here. Due to the influence of the output capacitor, the output voltage V _o has a ripple, and the voltage difference between the peak and the trough is large. Due to the approximate linear relationship between Vsense and V _o , the waveform of Vsense also shows an approximate sinusoidal ripple. Therefore, in the sampling process, the point at the midpoint of the ripple can be collected. According to the characteristics of sinusoidal fluctuation, the value of the feedback voltage Vref obtained at this time can better reflect the average value of the output voltage V _o .

The sampling circuit is shown in Figure 4. The sampling circuit includes a comparator, a DAC, and a calculation module. The calculation module is written in code and can function as a counter and an adder. Compare the magnitude of the voltage signal Vsense of the voltage dividing resistor R ₂ of the primary auxiliary winding with the magnitude of the feedback voltage Vref in the analog form obtained in the previous cycle through the comparator, and output the comparison result to the calculation module. Since the voltage signal Vsense of the voltage dividing resistor R ₂ of the auxiliary winding on the primary side exhibits an approximately sinusoidal ripple, and the feedback voltage Vref in the analog form obtained in each cycle is a fixed value, each cycle can be determined according to their equal points. Divided into two parts, one part is equal and then Vref continues to be smaller than Vsense until it is equal, and the other part is equal and then Vref continues to be greater than Vsense until it is equal. In the calculation module, the counter is used to collect the voltage signal Vsense of the auxiliary winding voltage divider resistor R2 _on the primary side in this cycle and the analog feedback voltage Vref obtained in the previous cycle. After the two voltages are equal, the time T1 of Vref< _Vsense , The time T ₂ of Vref>Vsense, that is, the size of T ₁ is the clock period of the period of Vref<Vsense. collected by the counter, and the size of T ₂ is the clock period of the period of Vref>Vsense. collected by the counter. The time difference between the two is Tr=T ₁ -T ₂ , and the size of Tr is the clock period that is different between T ₁ and T ₂ . According to the size of Tr, the digital error signal err of the current cycle is calculated, and the size of the digital feedback voltage VREF is adjusted. The digital VREF obtained at this time is input to the digital-to-analog converter, and after the analog Vref is obtained, it is combined with the auxiliary winding. The voltage signal Vsense is compared to form a closed-loop negative feedback to adjust the value of the feedback voltage Vref.

The sampling process is shown in Figure 5:

Compared with the voltage signal Vsense of the auxiliary winding, Vref will have the following three states:

Vref<Vsense, Vref=Vsense, Vref>Vsense., Vref<Vsense time T ₁ , the size of T ₁ is the clock cycle of this period of Vref<Vsense. collected by the counter, Vref>Vsense time T ₂ , T The size of ₂ is the clock cycle of this period of time when Vref>Vsense. collected by the counter. The time difference between the two is Tr=T ₁ -T ₂ , and the size of Tr is the clock period that is different between T ₁ and T ₂ . The size of the error signal err can be judged according to the size of Tr, and their adjustment relationship is as follows;

When Tr<-6, err=-3;

When -6<=Tr<=-4, err=-2;

When -3<=Tr<=-1, err=-1;

When Tr=0, err=0;

When 1<=Tr<=3, err=1;

When 4<=Tr<=6, err=2;

When Tr>6, err=3

According to the value of the error signal err, on the basis of the digital feedback voltage VREF in this cycle, adjust the size of the digital feedback voltage VREF in the next cycle. The next cycle VREF is equal to the current cycle VREF plus the error signal err, that is, in the next cycle VREF=VREF+err.

Referring to Figure 5(a), if T ₁ -T ₂ >0, err>0, the value of VREF in this cycle is small, and in the next cycle, VREF should increase the voltage value of err units, that is, VREF=VREF+err ;

Referring to Figure 5(b), if T ₁ -T ₂ =0, err = 0, the value of VREF in this cycle can well feedback the size of Vsense, and in the next cycle, VREF remains unchanged, that is, VREF=VREF+ 0=VREF+err;

Referring to Figure 5(c), if T ₁ -T ₂ <0, err<0, the value of VREF in this cycle is larger, and in the next cycle, VREF should reduce the absolute value of err by the voltage value of units, that is, VREF =VREF-|err|=VREF+err.

The above content is a further detailed description of the present invention in conjunction with the specific preferred embodiments. It cannot be considered that the specific implementation of the present invention is limited to these descriptions. The invention described here can have many changes, and such changes cannot artificially deviate from the present invention. spirit and scope. Therefore, all changes obvious to those skilled in the art are included within the scope of the claims.

Claims (3)

1. a kind of sampling method that improves the output stability of flyback resonant class switching power supply, based on MHz-level primary feedback flyback resonant converter topology, it is characterized in that: under full load condition, collect primary side auxiliary winding voltage following output by sampling module The change of the voltage V _o is input to the control module, and the control module generates the duty ratio control signal duty to adjust the on-time and off-time of the primary side switch tube to stabilize the output of the switching power supply; The voltage Vs on the secondary winding of the primary-side feedback flyback resonant converter is reflected to the voltage Va of the primary auxiliary winding in the ratio of V _s :V _{a =} N _s :N _aus , where N _s and _Naus are the secondary windings respectively. The number of turns of the side winding and the auxiliary winding, and the proportional relationship between the voltage Vsense across the voltage dividing resistor R ₂ and the voltage Va on the primary auxiliary winding is V _sense :V _{a =} R ₂ :(R ₁ +R ₂ ), R _1. R ₂ is the voltage divider resistor on the primary auxiliary winding. The sampling module collects the voltage waveform Vsense of the auxiliary winding through the voltage divider resistor R _2. Since Vsense is an analog voltage, it shows an approximate sinusoidal ripple, while Now the general control module deals with digital signals, so we need a constant feedback voltage to reflect the size of Vsense to the greatest extent. Vref and VREF are the analog and digital forms of the feedback voltage, respectively, and err is the value of each cycle. The error signal is a digital quantity. VREF is converted into Vref through a digital-to-analog converter. Different forms are used according to the needs. In order to make the value of the feedback voltage closer to Vsense, according to the characteristics of Vsense sinusoidal fluctuations, when collecting the Vsense ripple midpoint The voltage at the point is used as the initial value of the feedback voltage Vref in the analog form. The feedback voltage Vref of the analog quantity is directly compared with the voltage signal Vsense of the voltage dividing resistor R ₂ of the auxiliary winding on the primary side, and the feedback voltage Vref of the analog quantity is obtained through digital-to-analog conversion. The digital feedback voltage VREF, and then add and subtract the digital feedback voltage VREF in the calculation module to obtain the feedback voltage that reflects the size of Vsense to the greatest extent, and obtain the digital feedback required by the control module through the processing of the calculation module. The voltage VREF and the digital error signal err are used to calculate the output voltage V _o and the duty cycle control signal duty of the primary switch. 2. the sampling method that improves the output stability of the flyback resonant switching power supply according to claim 1, is characterized in that: the sampling module comprises a comparator, a DAC and a calculation module, and the calculation module is written by code and can realize the counter and the role of the adder. Compare the voltage signal Vsense of the voltage dividing resistor R ₂ of the primary auxiliary winding with the feedback voltage Vref in the analog form obtained in the previous cycle through the comparator, and output the comparison result to the calculation module. The voltage signal Vsense of ₂ shows an approximate sinusoidal ripple, and the feedback voltage Vref in the analog form obtained in each cycle is a fixed value. Each cycle can be divided into two parts according to the point where they are equal. Immediately, Vref will continue to be less than Vsense until it is equal, and after the other part is equal, Vref will continue to be greater than _Vsense until it is equal. After the feedback voltage Vref in the analog form obtained in one cycle is equal, the time T ₁ of Vref<Vsense, Vref> the time T ₂ of Vsense, that is, the size of T ₁ is the time of Vref<Vsense. collected by the counter The clock period, the size of T ₂ is the clock period during the period when Vref>Vsense. collected by the counter, the time difference between the two is Tr=T ₁ -T ₂ , and the size of Tr is the clock period that differs between T ₁ and T ₂ . According to the size of Tr, the digital error signal err of the current cycle is calculated, and the size of the digital feedback voltage VREF is adjusted. The digital VREF obtained at this time is input to the digital-to-analog converter, and after the analog Vref is obtained, it is combined with the auxiliary winding. The voltage signal Vsense is compared to form a closed-loop negative feedback to adjust the value of the feedback voltage Vref. 3. the sampling method that improves the output stability of flyback resonant switching power supply according to claim 2, is characterized in that: the sampling process of sampling module is as follows: Compared with the voltage signal Vsense of the auxiliary winding, Vref has the following three states: Vref<Vsense, Vref=Vsense, Vref>Vsense. The time T ₁ of Vref<Vsense, the size of T ₁ is Vref<Vsense collected by the counter. The clock period of this period of time, the time T ₂ when Vref>Vsense, the size of T ₂ is the clock period of this period of time when Vref>Vsense. collected by the counter. The time difference between the two is Tr=T ₁ -T ₂ , and the size of Tr is the clock period that is different between T ₁ and T ₂ . The size of the error signal err can be judged according to the size of Tr, and their adjustment relationship is as follows; When Tr<-6, err=-3; When -6<=Tr<=-4, err=-2; When -3<=Tr<=-1, err=-1; When Tr=0, err=0; When 1<=Tr<=3, err=1; When 4<=Tr<=6, err=2; When Tr>6, err=3 According to the value of the error signal err, on the basis of the digital feedback voltage VREF in this cycle, adjust the size of the digital feedback voltage VREF in the next cycle. The next cycle VREF is equal to the current cycle VREF plus the error signal err, that is, in the next cycle VREF=VREF+err.