CN109302109B - Flux weakening control method and control device for permanent magnet synchronous motor - Google Patents

Abstract

The invention discloses a field weakening control method and a control device for a permanent magnet synchronous motor. The method comprises: using the cross field weakening coefficient k of the dq axis to determine the current error values Δi _d and Δi _q of the d and q axes of the permanent magnet synchronous motor Make corrections to obtain _Δ'id and Δ'i _q ; calculate the voltage vector obtained by Δ'id and Δ'i _q after passing through the _d and q-axis current proportional-integral regulators respectively; output the corresponding drive signal according to the calculated voltage vector , to realize the control of the permanent magnet synchronous motor. The present invention is insensitive to motor parameters in the process of controlling motor operation, with small calculation amount and no need for filters, so that it is less affected by changes in motor parameters, the system response delay is small, and the performance is better.

Description

Field weakening control method and control device for permanent magnet synchronous motor

technical field

The invention relates to the technical field of permanent magnet synchronous motor control, in particular to a field weakening control method and control device of a permanent magnet synchronous motor.

Background technique

In the field of permanent magnet synchronous motor speed control, there are many applications that require high-speed operation of the motor. However, since the motor back electromotive force is proportional to the stator flux linkage and the motor speed, when the speed exceeds a certain value, it may be caused by the back electromotive force. The so-called voltage saturation occurs when the (EMF) is greater than the maximum stator voltage that the inverter can provide. The solution is to increase the DC bus voltage through a boost circuit; another method is to use field weakening control. The basic idea of field weakening control is to reduce the size of the EMF by giving the demagnetized d-axis current. Make the system have enough voltage to ensure that the motor runs in the high-speed area. Field weakening is a simple and effective method. From the given aspect of the d-axis current, scholars have proposed some methods to realize the high-speed field weakening operation of the motor.

For example, a look-up table method needs to calculate a look-up table in advance. During the actual operation of the system, the d-axis current information is obtained by looking up the table, but the running state of the motor changes in real time. Although this method of looking up the table is relatively simple But the error is bigger.

Another method is to solve the required d-axis current based on the analytical method of the permanent magnet synchronous motor model. Although this method is simple and fast, it is sensitive to the motor parameters. It is necessary to calculate the d-axis field weakening current at the current moment according to the voltage limit ellipse and the current limit circle. This analytical method requires the use of parameters such as the permanent magnet flux linkage and d-axis inductance of the motor. The calculation results are affected by the motor during the operation of the motor. The effect of parameter changes is greater.

In addition, there are literatures to obtain the d-axis field weakening current by means of error feedback such as voltage error feedback, current error feedback, etc., but these methods based on error feedback all require the design of filters, and the addition of filters will cause the system response to change. The delay affects the performance of the system. Therefore, it is necessary to develop a field weakening control method of permanent magnet synchronous motor with better performance, which is insensitive to motor parameters, requires little calculation and does not require filter design.

SUMMARY OF THE INVENTION

In view of this, the purpose of the present invention is to propose a field weakening control method and control device for a permanent magnet synchronous motor, which is insensitive to motor parameters, has a small amount of calculation, and does not require a filter during the operation of the control motor, thus being affected by changes in motor parameters. The impact is small, the system response delay is small, and the performance is better.

Based on the above purpose, the present invention provides a field weakening control method for a permanent magnet synchronous motor, comprising:

Correct the current error values Δid and Δi _q of the _d and q axes of the permanent magnet synchronous motor by using the cross field weakening coefficient k of the dq axis to obtain _Δ′id and Δ′i _q ;

Calculate the voltage vector obtained after Δ′i _d and Δ′i _q pass through the d and q-axis current proportional-integral regulators respectively;

The corresponding driving signal is output according to the calculated voltage vector, so as to realize the control of the permanent magnet synchronous motor.

Further, before the current error values Δi _d and Δi _q of the d and q axes of the permanent magnet synchronous motor are corrected by using the dq axis cross field weakening coefficient k, the method further includes:

和q轴电流初始值

以及当前的d轴电流采样值i_d和q轴电流采样值i_q分别计算d、q轴的电流误差值Δi_d和Δi_q。According to the d-axis current reference value of PMSM

and q-axis current initial value

and the current d-axis current sampling value id and q-axis current sampling value i _q to calculate the current error values _Δid and Δi _q of the _d and q axes, respectively.

Wherein, the coefficient k is a coefficient proportional to the rotational speed of the permanent magnet synchronous motor.

Wherein, the coefficient k=ω _r ·L;

Wherein, ω _r is the rotational speed of the permanent magnet synchronous motor, and L is the average inductance of the permanent magnet synchronous motor.

Or, the coefficient k is calculated according to the following expression 3:

为设置的乘积系数，ω_rbase为所述永磁同步电机的额定转速，ω_r为所述永磁同步电机的实际转速。in,

is the set product coefficient, ω _rbase is the rated speed of the permanent magnet synchronous motor, and ω _r is the actual speed of the permanent magnet synchronous motor.

The present invention also provides a field weakening control device for the permanent magnet synchronous motor, comprising:

The coefficient k is calculated according to the following expression 3:

为设置的乘积系数，ω_rbase为所述永磁同步电机的额定转速，ω_r为所述永磁同步电机的实际转速。in,

is the set product coefficient, ω _rbase is the rated speed of the permanent magnet synchronous motor, and ω _r is the actual speed of the permanent magnet synchronous motor.

In the technical solution of the embodiment of the present invention, the current error values Δid and Δi _q of the _d and q axes of the permanent magnet synchronous motor are corrected by using the cross field weakening coefficient k of the _d and q axes to obtain Δ'id and Δ'i _q ; Calculate the voltage vector obtained by Δ'id and Δ'i _q after passing through the _d and q-axis current proportional-integral regulators respectively; output the corresponding drive signal according to the calculated voltage vector to realize the control of the permanent magnet synchronous motor. In this motor control process, since there is no permanent magnet flux linkage, d-axis inductance and other parameters involved in the motor, it is not sensitive to changes in motor parameters, and is less affected by changes in motor parameters. Filters are not needed, and the amount of calculation is small, thus achieving a relatively high performance. excellent performance.

Description of drawings

1 is an internal structure diagram of a permanent magnet synchronous motor speed control system provided by an embodiment of the present invention;

2 is a flowchart of a method for field weakening control of a permanent magnet synchronous motor provided by an embodiment of the present invention;

3 is a block diagram of the internal structure of a field weakening control device for a permanent magnet synchronous motor provided by an embodiment of the present invention;

4 is a graph showing a simulation result of a permanent magnet synchronous motor with a rated frequency of 100 Hz provided by an embodiment of the present invention with a half-load sampling rate of 10 kHz, and the motor is directly increased from 5 Hz to 150 Hz;

5 is a screenshot of an oscilloscope data waveform of a simulation experiment provided by an embodiment of the present invention;

6 is a screenshot of the oscilloscope data waveform of the simulation experiment after the load is increased according to an embodiment of the present invention;

i_q和Δi_d的示波器数据波形截图；Fig. 7 is provided by the embodiment of the present invention through the DA output under the load of 7.5Nm,

Screenshots of oscilloscope data waveforms of i _q and _Δid ;

FIG. 8 is a torque and rotational speed curve diagram of a motor provided by an embodiment of the present invention.

Detailed ways

In order to make the objectives, technical solutions and advantages of the present invention clearer, the present invention will be further described in detail below with reference to specific embodiments and accompanying drawings.

The following describes in detail the embodiments of the present invention, examples of which are illustrated in the accompanying drawings, wherein the same or similar reference numerals refer to the same or similar elements or elements having the same or similar functions throughout. The embodiments described below with reference to the accompanying drawings are exemplary and are only used to explain the present invention, but not to be construed as a limitation of the present invention.

It will be understood by those skilled in the art that the singular forms "a", "an", "the" and "the" as used herein can include the plural forms as well, unless expressly stated otherwise. It will further be understood that when we refer to an element as being "connected" or "coupled" to another element, it can be directly connected or coupled to the other element or intervening elements may also be present. Furthermore, "connected" or "coupled" as used herein may include wirelessly connected or wirelessly coupled. As used herein, the term "and/or" includes all or any element and all combination of one or more of the associated listed items.

It should be noted that all expressions using "first" and "second" in the embodiments of the present invention are for the purpose of distinguishing two entities with the same name but not the same or non-identical parameters. It can be seen that "first" and "second" It is only for the convenience of expression and should not be construed as a limitation to the embodiments of the present invention, and subsequent embodiments will not describe them one by one.

The inventor of the present invention considers that the current error values Δi _d and Δi _q of the d and q axes of the permanent magnet synchronous motor are corrected by using the dq-axis cross field weakening coefficient k to obtain _Δ'id and Δ'i _q ; calculate The voltage vector obtained by Δ′i _d and Δ′i _q after passing through the d and q-axis current proportional-integral regulators, respectively, does not require similar traditional voltage amplitude feedback or current error feedback through the filter to obtain the d-axis voltage vector. The reference value of the field weakening current; instead, the cross field weakening coefficient k is used to realize the mutual adjustment and compensation of the dq-axis currents according to the actual operating conditions and the current working conditions. The specific performance is that when the motor runs in the high-speed region, the error feedback of the q-axis current is subtracted from the actual d-axis error feedback current to provide more field weakening current; at the same time, on the basis of the actual q-axis error feedback current On top of this, the error feedback of the d-axis current is added to ensure that enough output current is provided to the motor. The whole process is the mutual adjustment of the actual current of the dq-axis, without the participation of the filter. The specific implementation process is to correct the current error values Δi _d and Δi _q of the d and q axes of the permanent magnet synchronous motor by using the cross field weakening coefficient k of the dq axis to obtain _Δ′id and Δ′i _q ; calculate _Δ′id and Δ′i _q respectively through the d and q-axis current proportional-integral regulators to obtain the voltage vector; and then, according to the calculated voltage vector output corresponding driving signal, realize the control of the permanent magnet synchronous motor. In this motor control process, since there is no permanent magnet flux linkage, d-axis inductance and other parameters involved in the motor, it is not sensitive to changes in motor parameters, and is less affected by changes in motor parameters. Filters are not needed, and the amount of calculation is small, thus achieving a relatively high performance. excellent performance.

The technical solutions of the embodiments of the present invention will be described in detail below with reference to the accompanying drawings.

The internal structure of a permanent magnet synchronous motor speed control system provided by an embodiment of the present invention, as shown in FIG. 1 , includes: a three-phase voltage source, a permanent magnet synchronous motor, a DC side capacitor, a three-phase diode rectifier bridge, a voltage and current Sampling circuit, DSP controller and inverter and its drive circuit.

Among them, the voltage and current sampling circuit can use the voltage Hall sensor and the current Hall sensor to collect the DC side voltage and the a and b phase currents of the permanent magnet synchronous motor respectively. The DSP controller completes the operation of the method proposed by the present invention, and outputs the six-channel switching pulse signals after passing through the driving circuit to obtain the final driving signals of the six switching tubes of the inverter.

A specific process of a method for field weakening control of a permanent magnet synchronous motor provided by an embodiment of the present invention, as shown in FIG. 2 , includes the following steps:

Step S201: Calculate the current error values Δi _d and Δi _q of the d and q axes.

和q轴电流初始值

以及当前的d轴电流采样值i_d和q轴电流采样值i_q分别计算d、q轴的电流误差值Δi_d和Δi_q，即

In this step, according to the reference value of the d-axis current of the permanent magnet synchronous motor

and q-axis current initial value

and the current d-axis current sampling value id and q-axis current sampling value i _q to calculate the current error values Δi _d and Δi _q of the _d and q axes respectively, namely

可以采用现有方法得到。例如，根据转矩指令

得到

其中n_p为永磁同步电机的电机极对数，ψ_f为永磁体磁链，例如，n_p＝4。而转矩指令

可以通过速度环的PI(比例积分)调节器得到，如表达式一所示：Among them, the initial value of the q-axis current

can be obtained using existing methods. For example, according to the torque command

get

where n _p is the number of pole pairs of the permanent magnet synchronous motor, and ψ _f is the permanent magnet flux linkage, for example, n _p =4. while the torque command

It can be obtained by the PI (proportional-integral) regulator of the speed loop, as shown in Expression 1:

In expression 1, ω _r ^* is the reference speed of the permanent magnet synchronous motor, ω _r is the actual speed of the permanent magnet synchronous motor, s is the integral operator; K _p and K _i are the ratios in the PI regulator of the speed loop, respectively gain and integral gain.

一般为零。Among them, the initial given d-axis current reference value

Usually zero.

Step S202: Correct the current error values Δi _d and Δi _q of the d and q axes of the permanent magnet synchronous motor by using the dq-axis cross field weakening coefficient k to obtain _Δ'id and Δ'i _q .

In this step, the current error values Δi _d and Δi _q of the d and q axes of the permanent magnet synchronous motor are corrected by the cross field weakening coefficient k of the dq axis, and the corrected current error values Δ′id of the _d and q axes are obtained. and Δ′i _q .

Among them, the field weakening coefficient k is designed according to the current loop transfer function; the current loop transfer function is shown in Expression 2:

分别为系统计算电流环传递函数时的参考电流和参考电压矢量，j为复数运算算子，k_g＝K_p/L＝K_i/R，K_p和K_i分别为系统电流环比例系数和积分系数，s为积分算子。In Expression 2, R and L are the stator resistance and average inductance of the permanent magnet synchronous motor, respectively,

are the reference current and reference voltage vector when the system calculates the current loop transfer function, j is the complex operator, k _g =K _p /L=K _i /R, K _p and K _i are the system current loop proportional coefficient and Integral coefficient, s is the integral operator.

k is a coefficient proportional to the angular velocity of the permanent magnet synchronous motor, that is, a coefficient that increases proportionally with the speed. When k=ω _r ·L, the current loop transfer function can be obtained to be consistent with the complex vector PI, so k can be designed is: k=ω _r ·L, where ω _r is the actual speed of the permanent magnet synchronous motor.

More preferably, considering the operating state of the actual system, the coefficient k should be within a certain range, that is, k(ωr)min=0, then k can be calculated according to the following expression 3:

是一个与实际系统有关的乘积系数，可由本领域技术人员根据经验设置，比如，可以设置

ω_rbase为永磁同步电机的额定速度。in,

is a product coefficient related to the actual system, which can be set by those skilled in the art according to experience, for example, can be set

ω _rbase is the rated speed of the permanent magnet synchronous motor.

Step S203: Calculate the voltage vector obtained after Δ'id and Δ'i _q pass through the _d and q-axis current PI (proportional-integral) regulators, respectively.

In this step, according to the voltage equation on the two-phase rotating coordinate system in the mathematical model of the permanent magnet synchronous motor, as shown in Expressions 4 and 5, it can be calculated that Δ′i _d and Δ′i _q pass through the d and q axes respectively. The resulting voltage vectors _ud and u _q after the current PI regulator:

In expressions 4 and 5, L _q , L _d , i _q , and id represent the q-axis inductance, d-axis inductance, q-axis current, and _d -axis current of the permanent magnet synchronous motor, respectively, s is the integral operator, and ω _r is The actual speed of the permanent magnet synchronous motor;

In addition, K _p and K _i in Expression 4 are the proportional coefficient and integral coefficient of the d-axis current PI regulator, respectively; K _p and _Ki in Expression 5 are the proportional coefficient and integral coefficient of the q-axis current PI regulator, respectively coefficient.

Step S204 : output a corresponding driving signal according to the calculated voltage vector, so as to realize the control of the permanent magnet synchronous motor.

Specifically, the voltage vectors u _d and u _q obtained by calculation are firstly transformed into two-phase rotation to two-phase stationary coordinates to obtain the transformed voltage vectors u _α and u _β ; further, the voltage vectors u _α and u After _β is subjected to space vector pulse width modulation (SVPWM), a drive signal for driving the switch tube of the inverter is obtained, so as to realize the control and drive of the permanent magnet synchronous motor.

Based on the above method, an embodiment of the present invention provides a permanent magnet synchronous motor field weakening control device that can be set in a DSP controller. The internal structural block diagram, as shown in FIG. 3 , includes: a current error correction unit 301, a voltage vector A calculation unit 302 and a drive signal output unit 303 .

和q轴电流初始值

以及当前的d轴电流采样值i_d和q轴电流采样值i_q分别计算d、q轴的电流误差值Δi_d和Δ′i_q，进而利用dq轴交叉弱磁系数k对永磁同步电机的d、q轴的电流误差值Δi_d和Δi_q进行修正，得到Δ′i_d和Δ′i_q。Wherein, the current error correction unit 301 is used to correct the current error values Δi _d and Δi _q of the d and q axes of the permanent magnet synchronous motor by using the dq-axis cross field weakening coefficient k to obtain _Δ′id and Δ′i _q ; Specifically, the current error correction unit 301 may be based on the d-axis current reference value of the permanent magnet synchronous motor

and q-axis current initial value

and the current d-axis current sampled value id and q-axis current sampled value i _q to calculate the current error values _Δid and Δ′i _q of the _d and q axes, respectively, and then use the dq-axis cross field weakening coefficient k for the permanent magnet synchronous motor The current error values Δid and Δi _q of the _d and q axes are corrected to obtain _Δ′id and Δ′i _q .

The voltage vector calculation unit 302 is used to calculate the voltage vector obtained after Δ′id and Δ′i _q pass through the _d and q-axis current proportional-integral regulators respectively; specifically, the voltage vector calculation unit 302 can The voltage equation on the two-phase rotating coordinate system in the model is used to calculate the voltage vector obtained by passing Δ′id and Δ′i _q through the _d and q-axis current proportional-integral regulators respectively.

The drive signal output unit 303 is used to output the corresponding drive signal according to the calculated voltage vector, so as to realize the control of the permanent _magnet synchronous _motor ; After the coordinate transformation from two-phase rotation to two-phase stationary, the transformed voltage vectors u _α and u _β are obtained; the voltage vectors u _α and u _β are subjected to space vector pulse width modulation to obtain the voltage vectors used to drive the permanent magnet synchronous motor. The drive signal of the switch tube of the inverter realizes the control and drive of the permanent magnet synchronous motor.

For the specific implementation method of the functions of the above units, reference may be made to the respective step methods in the flow as shown in FIG. 2 , which will not be repeated here.

In the technical solution of the embodiment of the present invention, the current error values Δid and Δi _q of the _d and q axes of the permanent magnet synchronous motor are corrected by using the cross field weakening coefficient k of the _d and q axes to obtain Δ'id and Δ'i _q ; Calculate the voltage vector obtained by Δ'id and Δ'i _q after passing through the _d and q-axis current proportional-integral regulators respectively; output the corresponding drive signal according to the calculated voltage vector to realize the control of the permanent magnet synchronous motor. In this motor control process, since there is no permanent magnet flux linkage, d-axis inductance and other parameters involved in the motor, it is not sensitive to changes in motor parameters, and is less affected by changes in motor parameters. Filters are not needed, and the amount of calculation is small, thus achieving a relatively high performance. excellent performance.

The effectiveness of the method proposed in the present invention can be verified by simulation and experiments.

Figure 4 shows the simulation results of a permanent magnet synchronous motor with a rated frequency of 100Hz and a half-load sampling rate of 10kHz, the motor is directly increased from 5Hz to 150Hz, that is, the result of direct acceleration from 0.1 times the speed to 1.5 times the rated speed. The maximum current of the system is set to 10A, but the current is relatively large due to deep field weakening, so the simulation and experimental speed are only increased to 1.5 times the rated value. The load is 5Nm (half load), channel 1 is the command value and actual value of the motor speed, channel 2 is the electromagnetic torque, channel 3 is the flux linkage, and channel 4 is the a-phase current. As can be seen from Figure 4, when the motor speed is directly increased from 0.1 rated speed to 1.5 times the rated speed, the motor can still run in the constant torque zone at the beginning, and when the speed exceeds 0.1 times the rated speed, the motor runs in constant torque In the power zone, the final torque is stabilized at the load torque of 5Nm.

Figure 5 shows the experimental results corresponding to the simulation in Figure 4. The oscilloscope channel 1 corresponds to the motor speed, channel 2 corresponds to the torque, channel 3 corresponds to the permanent magnet flux linkage, and channel 4 corresponds to the motor phase a current. Fig. 6 is the experimental waveform corresponding to the experiment done in Fig. 5 only when the load is increased to 7.5Nm. The channel correspondence is consistent with Fig. 5, which proves that the method of the present invention can achieve good performance under different loads of the motor.

i_q和Δi_d的波形，对应的通道1为速度波形，通道2为i_q，通道3为

通道4为Δi_d。In order to more intuitively observe the change of the dq-axis current during the acceleration process of weak field, Figure 7 shows the output through the digital-to-analog conversion DA (Digital to Analog Convert) under the load of 7.5Nm.

The waveforms of i _q and _Δid , the corresponding channel 1 is the velocity waveform, channel 2 is i _q , and channel 3 is

Channel 4 is _Δid .

Figure 8 is the torque and rotational speed curve diagram of the motor. It can be seen that the motor is in the constant torque region when the base speed is below, and the motor is in the constant power region above the base speed.

As will be appreciated by those skilled in the art, the present invention includes apparatuses for performing one or more of the operations described in this application. These devices may be specially designed and manufactured for the required purposes, or they may include those known in general purpose computers. These devices have computer programs stored in them that are selectively activated or reconfigured. Such a computer program may be stored in a device (eg, computer) readable medium including, but not limited to, any type of medium suitable for storing electronic instructions and coupled to a bus, respectively Types of disks (including floppy disks, hard disks, optical disks, CD-ROMs, and magneto-optical disks), ROM (Read-Only Memory, read-only memory), RAM (Random Access Memory, random access memory), EPROM (Erasable Programmable Read-Only Memory, Erasable Programmable Read-Only Memory), EEPROM (Electrically Erasable Programmable Read-Only Memory, Electrically Erasable Programmable Read-Only Memory), flash memory, magnetic card or optical card. That is, a readable medium includes any medium that stores or transmits information in a form that can be read by a device (eg, a computer).

Those skilled in the art will understand that computer program instructions can be used to implement each block of these structural diagrams and/or block diagrams and/or flow diagrams, and combinations of blocks in these structural diagrams and/or block diagrams and/or flow diagrams . Those skilled in the art can understand that these computer program instructions can be provided to a general-purpose computer, a professional computer or a processor of other programmable data processing methods to implement, so that the present invention can be executed by a processor of a computer or other programmable data processing method. The block or blocks specified in the block or blocks of the block diagrams and/or block diagrams and/or flow diagrams of the invention are disclosed.

Those skilled in the art can understand that the various operations, methods, steps, measures, and solutions discussed in the present invention may be alternated, modified, combined or deleted. Further, other steps, measures, and solutions in the various operations, methods, and processes that have been discussed in the present invention may also be alternated, modified, rearranged, decomposed, combined, or deleted. Further, steps, measures and solutions in the prior art with various operations, methods, and processes disclosed in the present invention may also be alternated, modified, rearranged, decomposed, combined or deleted.

Those of ordinary skill in the art should understand that the discussion of any of the above embodiments is only exemplary, and is not intended to imply that the scope of the present disclosure (including the claims) is limited to these examples; under the spirit of the present invention, the above embodiments or There may also be combinations between technical features in different embodiments, steps may be carried out in any order, and there are many other variations of the different aspects of the invention as described above, which are not provided in detail for the sake of brevity. Therefore, any omission, modification, equivalent replacement, improvement, etc. made within the spirit and principle of the present invention shall be included within the protection scope of the present invention.

Claims (6)

1. A flux weakening control method for a permanent magnet synchronous motor is characterized by comprising the following steps:

according to d-axis current reference value of permanent magnet synchronous motor

And initial value of q-axis current

And the current d-axis current sample value i_dAnd q-axis current sample value i_qRespectively calculating current error values delta i of d and q axes_dAnd Δ i_q；

Current error value delta i of d and q axes of permanent magnet synchronous motor by using dq axis cross weak magnetic coefficient k_dAnd Δ i_qCorrecting to obtain corrected d and q axis current error value delta' i_dAnd Δ' i_qThe method specifically comprises the following steps: at Δ i_dIs subtracted by k times the corrected q-axis current error value delta' i_qTo obtain Delta' i_dAt Δ i_qIs added with k times of corrected d-axis current error value delta' i_dTo obtain Delta' i_q；

Calculating Delta' i_dAnd Δ' i_qVoltage vectors are obtained after the voltage vectors pass through a d-axis current proportional-integral regulator and a q-axis current proportional-integral regulator respectively;

outputting a corresponding driving signal according to the calculated voltage vector to realize the control of the permanent magnet synchronous motor;

wherein the coefficient k is a coefficient proportional to the angular velocity of the permanent magnet synchronous motor.

2. The method of claim 1, wherein the calculating Δ' i_dAnd Δ' i_qThe voltage vectors obtained after passing through the d-axis current proportional-integral regulator and the q-axis current proportional-integral regulator respectively are as follows:

calculating the delta' i according to a voltage equation on a two-phase rotating coordinate system in a mathematical model of the permanent magnet synchronous motor_dAnd Δ' i_qAnd voltage vectors obtained after the voltage vectors pass through the d-axis current proportional-integral regulator and the q-axis current proportional-integral regulator respectively.

3. The method according to claim 1, wherein outputting the corresponding driving signal according to the calculated voltage vector to realize control of the permanent magnet synchronous motor specifically comprises:

the calculated voltage vector u_d、u_qAfter the coordinate transformation from two-phase rotation to two-phase static, the transformed voltage vector u is obtained_αAnd u_β；

Will voltage vector u_αAnd u_βAnd obtaining a driving signal of a switching tube of an inverter for driving the permanent magnet synchronous motor after space vector pulse width modulation, and realizing control and driving of the permanent magnet synchronous motor.

4. Method according to claim 1, characterized in that said coefficient k ═ ω_r·L；

Wherein, ω is_rAnd L is the actual speed of the permanent magnet synchronous motor, and L is the average inductance of the permanent magnet synchronous motor.

5. A permanent magnet synchronous motor flux weakening control device is characterized by comprising:

a current error correction unit for correcting current error value delta i of d and q axes of the permanent magnet synchronous motor by using dq axis cross weak magnetic coefficient k_dAnd Δ i_qCorrecting to obtain corrected d and q axis current error value delta' i_dAnd Δ' i_qThe method specifically comprises the following steps: at Δ i_dOn the basis ofK times subtracted corrected q-axis current error value Δ' i_qTo obtain Delta' i_dAt Δ i_qIs added with k times of corrected d-axis current error value delta' i_dTo obtain Delta' i_q(ii) a Wherein, Δ i_dIs based on d-axis current reference value of permanent magnet synchronous motor

And the current d-axis current sample value i_dCalculated,. DELTA.i_qIs based on the initial value of q-axis current of the permanent magnet synchronous motor

And the current q-axis current sample value i_qCalculating to obtain;

a voltage vector calculation unit for calculating Δ' i_dAnd Δ' i_qVoltage vectors are obtained after the voltage vectors pass through a d-axis current proportional-integral regulator and a q-axis current proportional-integral regulator respectively;

the driving signal output unit is used for outputting a corresponding driving signal according to the calculated voltage vector to realize the control of the permanent magnet synchronous motor;

wherein the coefficient k is a coefficient proportional to the angular velocity of the permanent magnet synchronous motor.

6. The apparatus of claim 5,

the drive signal output unit is specifically configured to output the calculated voltage vector u_d、u_qAfter the coordinate transformation from two-phase rotation to two-phase static, the transformed voltage vector u is obtained_αAnd u_β(ii) a Will voltage vector u_αAnd u_βAnd obtaining a driving signal of a switching tube of an inverter for driving the permanent magnet synchronous motor after space vector pulse width modulation, and realizing control and driving of the permanent magnet synchronous motor.

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