CN109194228A - A kind of starting of salient pole type synchronous motor position-sensor-free and method for control speed - Google Patents
A kind of starting of salient pole type synchronous motor position-sensor-free and method for control speed Download PDFInfo
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- CN109194228A CN109194228A CN201811199304.XA CN201811199304A CN109194228A CN 109194228 A CN109194228 A CN 109194228A CN 201811199304 A CN201811199304 A CN 201811199304A CN 109194228 A CN109194228 A CN 109194228A
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
- H02P21/18—Estimation of position or speed
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
- H02P21/34—Arrangements for starting
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P25/00—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details
- H02P25/02—Arrangements or methods for the control of AC motors characterised by the kind of AC motor or by structural details characterised by the kind of motor
- H02P25/022—Synchronous motors
- H02P25/024—Synchronous motors controlled by supply frequency
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- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P27/00—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage
- H02P27/04—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage
- H02P27/06—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters
- H02P27/08—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters with pulse width modulation
- H02P27/085—Arrangements or methods for the control of AC motors characterised by the kind of supply voltage using variable-frequency supply voltage, e.g. inverter or converter supply voltage using DC to AC converters or inverters with pulse width modulation wherein the PWM mode is adapted on the running conditions of the motor, e.g. the switching frequency
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- Power Engineering (AREA)
- Control Of Ac Motors In General (AREA)
- Control Of Motors That Do Not Use Commutators (AREA)
Abstract
The present invention provides a kind of starting of salient pole type synchronous motor position-sensor-free and method for control speed, it is characterized in that, whole system is mainly estimated by stationary state rotor-position, the funtion parts such as the control of speed rotating vector and pattern switching timing and initializing signal generation form, and speed rotating vector control section controls two stages by speed open loop and speed closed loop rotating vector again and forms, to guarantee that motor during startup centainly will not be with inverting and spinner velocity starts from scratch and smoothly accelerates to specified motion state for the moment.There are the application scenarios of strict demand when the invention is particularly suitable for those electric motor startings to steering and velocity variations, as motor driven heavy load requires motor steering to rotate according to the direction of regulation always to when very high-revolving application, and require speed to start from scratch continuously and smoothly to increase to the application scenarios of specified revolving speed.
Description
Technical Field
The present invention relates to a position sensorless start-up and speed control method for a salient pole synchronous motor, and more particularly to a start-up control and continuous speed vector control method that do not involve reverse rotation.
Background
Compared with induction motors and direct current DC motors, permanent magnet synchronous motors have the advantages of high efficiency, high speed and dynamic characteristics, so that the permanent magnet synchronous motors are already used in industrial and domestic electromechanical equipment such as electric vehicles, tramways, elevators, air conditioners, pumps and the like. The control techniques for synchronous motors used in these applications are currently mainly two control techniques, according to the requirements of control performance and cost: (1) for some simple application scenes such as the application of a fan, a pump and a compressor, the traditional V/F control method is often adopted because the hardware is simple to build, the cost is low, and the speed regulation performance meeting the application requirement can be obtained in many times; (2) for the situations with special control requirements, such as sensorless control, where high speed and high starting torque are required, and torque control is required, the rotating vector control method is often adopted.
The basic idea of the rotating vector control technology is to decompose stator current into excitation current and torque current and achieve the purpose of accurately controlling output torque by independently controlling the two current components. During this current switching process, the absolute position of the rotor (i.e., the angle of orientation of the magnetic field) must be accurately obtained. Meanwhile, position information of the rotor is also required when the two-dimensional rotating coordinate system is transformed into the three-phase stationary coordinate system.
In order to determine the rotor position, in synchronous motor control with a position sensor, sensors such as an encoder and a hall device are generally used to accomplish this task, but there are problems in that the cost of the system is increased, the wiring is complicated, the volume is increased, and the reliability of the system is lowered. Currently, as the device size becomes smaller and the number of high-speed application scenarios increases, a position-sensor-less control technique without mechanical constraints is gradually adopted. One problem that arises is that the position information of the rotor cannot be obtained directly from the sensor. The solution commonly used today is to estimate the rotor position information by some other signal that can be detected, such as a method of estimating the rotor angle in real time by induced electromotive force information and a method of estimating the rotor angle in real time by a response signal after injecting a high frequency electric signal. However, in consideration of the processing speed and the amount of data to be processed by the processor, the rotor angle is generally estimated by a method of inducing an electromotive force and is used in a rotary vector control system. In the method of estimating the angle of the rotor using the induced electromotive force, since there is no detectable current signal generated due to the induced electromotive force when the motor is in a stationary state, absolute position information of the rotor in the stationary state cannot be obtained. Therefore, many proposed starting methods for the sensorless motors generally apply a synchronization signal for calibration and then accelerate, and then use the induced electromotive force method to complete the estimation of the rotor position for feedback control. One problem with this approach is that the direction of rotation at start-up is uncertain when the synchronisation signal is applied without knowledge of the initial position of the rotor. This problem is not allowed for applications with strict starting direction requirements. If a very large load is fixedly connected to the output shaft of the motor, the speed of the rotor is required to be continuously accelerated from a static state in a single direction (energy loss or system damage caused by short reversal and the like are generated). Therefore, the present invention provides a method for unidirectional start and continuous speed control without a position sensor.
Disclosure of Invention
The technical problem to be solved by the invention is as follows: when the salient pole type permanent magnet synchronous motor is started and controlled in speed without a position sensor, the motor can be ensured to be continuously controlled in speed from a static state according to a preset direction all the time, and the possibility that the motor is reversed for a while with a specified rotating direction in the starting process of the existing starting method is eliminated.
The invention solves the technical problem and adopts the technical scheme that the salient pole type permanent magnet synchronous motor sensorless rotation vector control method mainly comprises 5 parts. The 5 functional parts are a driver and a passive object, a static state rotor position estimating part, a speed open loop vector control part, a speed closed loop vector control part, and a motion mode switching timing and initialization signal generating part, respectively.
(1) Driver and controlled object
The three-phase salient pole permanent magnet synchronous motor is a controlled object of the invention. The invention adopts the three-phase alternating current sinusoidal signal to drive the motor, and has the advantage that different control strategies can be conveniently executed on the motor through the adjustment of the amplitude and the phase of the current vector. As shown in fig. 1, the controlled motor 100 is connected to a three-phase inverter bridge circuit 102 as a driver. The three-phase inverter bridge circuit is a three-phase bridge circuit comprising switching elements of anti-parallel diodes. The unit 102 can regulate the voltage applied to the motor ports by controlling the terminals 104, 106,108,110,112,114 to be on and off. In the speed open-loop and speed closed-loop control, the unit 102 mainly generates an alternating voltage signal by Pulse Width Modulation (PWM) performed by a controller on a direct current voltage connected with the unit and provides the alternating voltage signal for the controlled motor.
For the sake of convenience, the several coordinate systems referred to herein will be described in connection with the unit 100 of fig. 1, first, the uvw coordinate system 116 is a stationary coordinate system represented by an original three-phase ac signal, the αβ coordinate system 118 is a stationary coordinate system represented by a two-dimensional representation in which the physical quantities represented are ac signals, the dq coordinate system 122 is a coordinate system rotating synchronously with the rotor on the basis of the αβ coordinate system in which the physical quantities represented are dc components, and the rotation vector control is generally performed in this coordinate system, and in the position-sensorless synchronous motor, the δ γ coordinate system 120 is introduced as a coordinate system fixedly connected to the estimated rotor position because the rotor angle value cannot be directly obtained.
(2) Rotor position estimating section in stationary state
In order to ensure that the motor does not generate negative torque during starting and acceleration, the rotor position information when the motor is stationary needs to be estimated. The invention adopts the technical scheme that a high-frequency voltage pulse signal is directly provided for the motor through a switch of the control unit 102, and then the rotor position is estimated through a current response value and the physical characteristics of the salient pole type motor.
In order to ensure that the motor rotor is always kept in a static state in the initial position estimation stage of the motor, a power supply module with smaller voltage is used for initial rotor position estimation, and when the working operation stage is entered, the power supply module is switched to a normal working power supply module to supply power. Estimating rotor position theta in motor stationary staterHas the following formula
As mentioned above, the invention adopts the current component i in αβ coordinate system measured after injecting high frequency voltage pulse signalsα,isβTo estimate the initial position of the rotor.
Substituting equation 1 into a loop mathematical model of the salient pole permanent magnet synchronous motor expressed in an αβ coordinate system can obtain a state equation of an input voltage vector and an output response current vector, which is as follows
Wherein, the coefficient matrix A 'and B' in equation 2 are as follows
Wherein, A and B in the equations 3 and 4 are
Wherein L isd，LqThe inductance of the winding is expressed in the dq coordinate system.
Therefore, according to equations 2,3,4 and 5, it can be known that when a certain input voltage signal is given, the initial current of the motor is zero, and the current response is 2 times of the rotor electric rotation angle, namely 2 thetarThe sine and cosine function of (a) varies. The position of the motor rotor can be estimated from the angle information included in the response current and the magnetic saturation physical characteristics of the motor.
(3) Velocity open loop vector control section
When the motor moves at low speed, the rotor angle error estimated according to the back electromotive force is large, so the invention drives and accelerates the motor by a speed open loop vector method from the standstill. When the speed is accelerated to a certain rotating speed, the speed is switched to the speed closed-loop vector control so as to improve the influence of the rotating speed on the interference from the outside.
In order to ensure that the rotor is not reversed in the starting stage, the invention designs a speed open-loop controller to generate starting and initial accelerating driving currentThis current flow ensures α from the initial angular position of the rotor0Drive torque is initiated in a specified direction. In order to cope with the change of the external load, the current closed-loop rotating vector control is carried out by combining the components of the three-phase current measured in real time in the dq coordinate system.
The implementation of the speed open loop controller here consists of two main parts. The first part is to preliminarily estimate a deviation Δ θ between the presumed axis coordinate system δ γ and the dq coordinate system representing the actual position of the rotating shaft at low speedc. The second part is a generating driverThe current drives the deviation towards 0. Since oscillations in current and speed occur when switching from the speed open-loop mode to the closed-loop mode when there is a phase angle deviation between the estimated and actual axes, the phase angle between the estimated and actual axes is made as close to 0 as possible during the open-loop control phase in order to reduce signal instability due to the existing rotational angle difference.
Here the spindle error Δ θcCalculated using equation 6 below
Wherein R issIs a stator winding of a permanent magnet synchronous motor;a voltage command value (after non-interference control) represented in the dq coordinate system outputted for the current controller; l isd,LqThe inductance value of the stator coil in the dq coordinate system; omegaeThe current electrical angular speed of the motor rotor is used, and the actual rotating speed of the rotor cannot be measured at the current electrical angular speed, so that the rotating shaft error is estimated by adopting an input rotating speed instruction; i.e. isd,isqThe components represented in the dq coordinate system for the current measured in real time.
The rotation shaft error delta theta is compared with a given 0 value, and then the motor is driven to operate by integrating the driving current output by the controller, so that the delta theta gradually approaches 0 before entering a speed closed-loop control phase. The output current command of the speed open-loop controller is recorded asIt has the following calculation formula
Wherein, KIThe angle initial value of the integral controller is set as the estimated rotor initial angle; i ismaxIs the current limit amplitude.
(4) Velocity closed loop vector control section
After the motor speed reaches a certain value, the speed control mode is switched to speed closed-loop rotating vector control. Compared with the speed open-loop vector control, the control mode adds a speed control closed loop and a moment control closed loop. Depending on the requirements for control performance, the torque control loop may sometimes be omitted.
In this control phase, the estimation of the rotor speed and the real-time rotational angle position is estimated from the generalized induced back electromotive force information defined below.
The loop equation expressed in the dq coordinate system can be expressed as follows after deformation
In transforming equation 10 to fixed coordinate system αβ, it may be expressed as follows
The second term on the right side of the above formula is defined as the generalized induced electromotive force expressed in a fixed coordinate system, and is expressed as the following formula
As can be seen from equation 12, the generalized induced electromotive force defined herein is composed of the motor parameters and the rotor position and rotation speed, so that the rotor position and rotation speed information can be extracted from the generalized induced electromotive force.
(5) Motion mode and signal switching timing generation
For the convenience of the description of the present invention, the motion state in which the motor is accelerated from the stopped state to the specified operation speed without reversal is divided into 3 mode states, i.e., a rotor initial angle estimation mode (denoted as "mode 1"), a motor start and acceleration mode under speed open loop vector control (denoted as "mode 2"), and a motor operation mode under speed closed loop vector control (denoted as "mode 3").
The part mainly realizes the functions of three aspects:
①, the control sequence for switching from mode 1 to mode 2 generates the electrical angle information and speed information required for vector control in mode 2 because the system has no position sensor, the speed at this stage is directly taken as the input speed command, the angle is integrated using the speed command, and the initial value of the angle is set to the rotor position angle estimated in mode 1.
②, and generates the electric rotation angle information and the rotation speed information required for the vector control in mode 3, and selects and outputs the rotor electric rotation angle and the rotation speed estimated from the electromotive force in mode 3.
③ in order to avoid signal discontinuity or current, speed, etc. signal oscillation during mode switching, the present invention needs to initialize some physical parameters of the target motion mode to the corresponding physical quantity at the end of the previous motion mode.
Drawings
FIG. 1 is a three-phase inverter and synchronous machine object for driving a motor according to the present invention and three coordinate systems referred to in the description of the present invention;
FIG. 2 is a block diagram of the PMSM start timing control and overall system control in an embodiment of the present invention;
FIG. 3 is a block diagram of a speed open loop current command generation employed in an embodiment of the present invention;
FIG. 4 shows the inverter switch on/off timing sequence when the initial rotor angle is estimated;
fig. 5 shows the comparison between the estimated values of the rotor electrical angle and the speed at the open-loop and closed-loop speed stages and the actual values of the motor, where 0.9s is the switching point from the open-loop speed stage to the closed-loop speed stage, and the trajectory shown by the solid line in the figure shows that during the actual starting and accelerating process, the motor does not have the rotor reversion and the speed is continuously increased from 0.
Detailed Description
Hereinafter, embodiments for carrying out the present invention will be described in detail with reference to the accompanying drawings. Fig. 2 shows an example of a drive control device for a three-phase permanent magnet synchronous motor according to the present invention. Here, a description is given of the functions of the respective portions referred to in fig. 2 in the order of motor start-up.
The unit 100 shown in fig. 1 and 2 is the controlled object of the system, i.e. a salient pole three-phase permanent magnet synchronous motor without a position sensor. Directly connected to the controlled object is a three-phase inverter bridge circuit (unit 102 shown in fig. 1 and 2) which mainly generates an alternating voltage signal to the controlled motor by the pulse width modulation PWM of the controller from the direct voltage connected to the three-phase inverter bridge circuit. The motor is driven by the three-phase alternating current sinusoidal signal, and different control methods can be conveniently realized by adjusting the amplitude and the phase of the current vector.
(1) At rest stateLower rotor position α0Estimation of (mode 1)
According to the present invention, it is necessary to estimate the initial position of the rotor before the motor is started. The implementation of this function requires the power supply 1 unit 200, the oscillator unit 202, the current detection unit 204, the coordinate transformation units 206, 208 and the units 100,102 of fig. 2 to be implemented together.
Unit 200 is a power supply module dedicated to initial angle estimation. In the present embodiment, in order to ensure that the motor rotor remains stationary during the angle estimation phase, a small dc voltage of 10V is used.
Element 202 generates a pulse width TsThe timing logic unit of the high-frequency pulse voltage signal. This sequence is used to control the switching of the 6 switches in unit 102 to apply a pulsed voltage signal to the motor stator. The present embodiment utilizes a high speed controller to control the inverter switches to provide the pulse width TsInitial angle estimation was performed for a 3ms pulsed voltage signal. Fig. 3 shows the switching on and off sequence of the inverter bridge circuit when a pulse voltage signal is supplied to the U-phase alone. When u is+When it is logic 0, u+When the switch 104 is turned off and is logic 1, u+Switch 104 is closed and the control logic for the other inverter switches 106,108,110,112,114 is the same as the operation of switch 104.
The cell 204 is a current detection cell. In the embodiment, the current sensor is adopted to acquire the first phase current i of the permanent magnet synchronous motorsuSecond phase current isvThen a third phase current IswIs obtained by the following formula
isw＝-isu-isvEquation 13
Unit 206 is a coordinate transformation unit that implements two transformations, respectively:
① three-phase transformation to two-phase stationary αβ coordinate transformation, the current component i converted to αβ coordinate transformation is needed in the initial angle estimation stagesα,isβThe calculation formula is
② three-phase to two-phase rotational dq coordinate transformation for converting the obtained three-phase alternating current to a two-phase direct current component i represented in a dq coordinate system rotating synchronously with the rotorsd,isq. The calculation for performing this transformation using Sin functions only is as follows
Where K is a transform coefficient, when performing energy-invariant coordinate absolute transformationWhen performing a relative transformation with constant amplitude before and after the transformation
The unit 208 is a unit that specifically performs angle estimation. The unit is represented bysα,isβThe current component is used as an input, and finally, a unique rotor initial angle value α is output0. When applying pulse width T to U phasesAfter voltage signal of, i.e. voltage mode
Solve the following after substituting equation 2
Wherein, item IsαAnd Δ IαβAre respectively shown as the following formula
RsIs a stator winding of a permanent magnet synchronous motor; l isd,LqFor a given d, q-axis inductance value
The initial angle α can be obtained from equation 170Is thetarOr thetar+ pi, and the unique position angle is determined from the two, the physical property of field saturation of the salient pole permanent magnet synchronous machine can be used for selective determination.
After "mode 1" is completed, the switching unit 210 is switched to enter the speed open-loop vector control mode (i.e., "mode 2") under the control of the initial value setting signal unit 212 in the sport mode and signal switching timing generation functional block.
(2) Open loop vector control of velocity (mode 2)
When the motor is at rest and moves at low speed, the back electromotive force is small, so the invention drives the motor by speed open loop from rest to accelerate to a certain speed, and switches to closed loop speed vector control0Generating a drive current that produces a positive torqueAnd (3) combining the components of the three-phase current measured in real time in the dq coordinate system to perform current closed-loop rotation vector control so as to respond to the change of the external load.
The speed open-loop driving motor is mainly composed of 9 functional units, namely a controlled object unit 100, a three-phase inverter bridge circuit unit 102, a phase current detection unit 204, a coordinate transformation unit 206, a speed open-loop driving current generation unit 208, a current controller unit 230, a 2-phase/3-phase coordinate transformation unit 232, a PWM signal transformation unit 234 and a DC power supply 2 unit 236. The controlled object unit 100, the three-phase inverter bridge circuit unit 102, the phase current detection unit 204, and the coordinate transformation unit 206 are the same as described above. The speed open loop driving current generating unit 208, the current controller unit 230, the 2-phase/3-phase coordinate transforming unit 232, the PWM signal converting unit 234, and the DC power supply 2 unit 236 will be described below.
The speed open loop drive current generation unit 208 is mainly composed of two functional blocks. One function is to correct the deviation of the rotation angle Δ θ between the control axis δ γ and the actual rotor axis dq at low speedcPresume; the other is a current command signal for making the deviation angle approach to zeroA controller is generated. When there is a phase angle deviation between the control axis δ γ and the actual rotor axis dq, current and speed oscillations occur when switching from the speed open-loop mode to the closed-loop mode. In order to reduce signal instabilities due to the existing difference in rotation angle, the deviation angle between the control shaft and the actual rotational shaft is brought towards zero as much as possible in the open-loop control phase. Therefore, the open-loop drive current generation unit includes a spindle error estimation unit and a speed open-loop controller unit. In the phase of speed open loop, the position error delta theta between the control shaft and the actual rotor shaft is calculated by adopting equation 9cIt can be seen from the equation that it is suitable for estimating the error of the rotating shaft when the motor is at rest or at low speed.
Wherein,outputting a voltage command value for the current controller, and providing the voltage command value for d and q axes of a motor stator after non-interference processing; i.e. isd,isqFor real-time phase power obtained according to equation 15A current d, q-axis current component; angular velocity ω of 220 unit output in mode 2e_fbkFor electric angular speed command, i.e. input speed ω of mechanical shaftm_refThe output after passing through the angular velocity conversion unit 216 is as follows
ωe_fbk＝Npωm_refEquation 21
The speed open loop controller unit outputs a value delta theta through a rotating shaft error estimation unitcComparing with a given 0 value, and then driving the motor to operate by integrating the driving current output by the controllercAnd gradually approaches 0 before entering the speed closed-loop control phase. The speed open loop current controller function is realized by the following 3 formulas
Wherein, KIFor the integral proportional constant of the controller, K is taken in this exampleI＝1；ImaxIs a current limit amplitude
The current closed loop controller unit 230 takes as input the difference between the d-axis, q-axis current command and the current component detected in real time in the dq coordinate system, and in this embodiment employs a PI controller to generate the voltage command required in the dq coordinate system. And in order to facilitate the setting of controller parameters, a non-interference control method is adopted to decouple the mutual influence of the currents between the d-axis and the q-axis. Then, in this embodiment, an internal model method is used to design the proportionality coefficient K of the PI controllerP_sd,KP_sqAnd integral coefficient KI_sd,KI_sqAre respectively shown as the following formula
Wherein, Tsd_cl，Tsq_clRepresenting the d-axis and q-axis closed-loop current time constants, respectively. These two parameters can be adjusted within a certain range according to the requirement of the response speed of the system.
The 2-phase/3-phase coordinate transformation unit 232 is used for outputting two-phase voltage commands expressed in d-axis and q-axis output by the current PI controller(after non-interference control) to voltage command value expressed in three-phase uvwThe equation for implementing the conversion with the uniform sin function is shown below.
Wherein K' is a transformation coefficient when performing energy-invariant coordinate absolute transformationWhen relative transformation is performed with the amplitude unchanged before and after transformation, K' is 1.
The PWM signal conversion unit 234 is a circuit that generates a PWM wave from a command value of an applied voltage.
The DC power supply 2 unit 236, unlike the DC power supply 1 unit 200 employed in the initial angle estimation, is an operating power supply employed when switching to mode 2 and mode 3.
(3) Velocity closed loop vector control (mode 3)
Under the control of the motion mode and signal switching timing generation part switch unit 228, the switching timing unit 220, the control system accesses the closed loop speed control. Element 228 switches the d-axis and q-axis current commands of the open loop phase to the current commands generated by torque controller element 240. Unit 220 determines the pole rotation angle and the rotational speed source used in the control system. In a closed-loop speed control system, the two data are the output of the rotor angle and speed estimation unit 222.
As shown in fig. 2, the speed closed-loop control system implementation mainly consists of the following functional units. In addition to the controlled object unit 100, the three-phase inverter bridge circuit unit 102, the phase current detection unit 204, the coordinate conversion unit 206, the current controller unit 230, the 2-phase/3-phase coordinate conversion unit 232, the PWM signal conversion unit 234, and the DC power supply 2 unit 236 described in the speed open-loop vector control section, the speed controller unit 238, the torque controller unit 240, the torque estimation unit 242, the rotor speed and rotation speed estimation unit 222, and the conversion unit 218 that converts the electrical rotation angular speed into the mechanical rotation speed are included.
The speed controller unit 238 commands ω with the speedm_refAnd real-time estimation of the rotational speed omegam_fbkThe PI controller is adopted in the embodiment to generate a torque command, and a proportional coefficient K of the PI controller is setP_wAnd integral coefficient KI_wAs follows
The torque controller unit 240 receives the difference between the torque command output from the speed controller and the torque estimated in real time, and outputs d-axis and q-axis current commands. The unit also employs a PI controller and sets the proportionality coefficient K of the PI controllerP_trqAnd integral coefficient KI_trqAs follows
In the above formula, NpIs the number of magnetic pole pairs; Ψ is the magnetic field strength; t isisq_clA time constant set for the q-axis current closed loop; t isclThe response time constant is closed loop for the set torque.
The torque estimation unit 242 uses the filtered real-time measured current value isdlpf,isplpfThe feedback torque is estimated by the following torque calculation formula,
Tclc＝1.5Np{Ψ+(Ld-Lq)isdlpf}isqlpfequation 29
The rotor speed and rotation speed estimating unit 222 estimates the rotor speed and rotation speed by an extended induced electromotive force method. Equations 10,11,12 and the following equation are executed to output the rotor angle, and the rotor angle is differentiated to derive the angular velocity.
The present invention has not been described in detail as is known to those skilled in the art.
Claims (7)
1. A position sensorless control and drive technique for salient pole synchronous motor is composed of the functions of estimating the position of rotor in static state, controlling the speed and rotation vector, switching mode, generating initial signal, etc.
2. The position sensorless synchronous motor control and drive of claim 1 wherein there is no possibility of motor reversal during start-up.
3. The control and drive device of a position sensorless synchronous motor according to claim 1, characterized in that the speed control is composed of speed open loop and closed loop rotation vectors, and the speed open loop control generates a current command that causes the motor to generate a forward torque from a standstill.
4. The control and drive device of a sensorless synchronous machine of claim 1 wherein there is a current control closed loop in the speed open loop control stage to accommodate external load variations.
5. The open-loop control of speed according to claim 3, wherein the generation of the current command is performed by a controller using the estimated value of the magnetic field direction and the deviation value of the actual magnetic field direction as inputs, to reduce the oscillation of the speed and current signals generated when the speed control mode is switched.
6. The vector control of claim 3, wherein when the open-loop control of speed is switched to the closed-loop vector control, the initial values of the angular velocity and the angular velocity are initialized to the respective corresponding values at the end of the open-loop control of speed in the estimation module of the angular velocity and the rotor angle used in the closed-loop control of speed to ensure the continuity of the signal, thereby avoiding the oscillation and instability of the signal.
7. The closed-loop vector control of speed according to claim 3, wherein the estimation of the rotor angle information used in the vector control is performed using an extended induced electromotive force.
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| CN201811199304.XA Pending CN109194228A (en) | 2018-10-15 | 2018-10-15 | A kind of starting of salient pole type synchronous motor position-sensor-free and method for control speed |
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| WO2024235323A1 (en) * | 2023-05-18 | 2024-11-21 | 华润微集成电路(无锡)有限公司 | Forward and reverse rotation control circuit for sensorless motor, method and motor controller |
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