CN107508517A - A kind of low-speed electronic automobile AC induction motor vector control method and system - Google Patents

Abstract

The present invention provides a low-speed electric vehicle AC asynchronous motor vector control method and system, comprising: step 10, judging the sector where the voltage vector is located; step 20, calculating the action time of the basic voltage vector; step 30, calculating the conduction of each bridge arm time and build the dead zone time; step 40, drive and control the motor. The invention can effectively reduce current distortion, running noise and torque ripple.

Description

A low-speed electric vehicle AC asynchronous motor vector control method and system

technical field

The invention belongs to the field of low-speed electric vehicles, in particular to a vector control method and system for AC asynchronous motors of low-speed electric vehicles.

Background technique

With the continuous increase of the dual pressure of resources and environment, electric vehicles have become the development direction of the future automobile industry. Judging from China's current market capacity and technical level, low-speed electric vehicles with a speed of 40-60km/h have the advantages of good economic performance, energy saving and environmental protection, resource conservation, low cost of use, and convenient charging. The most environmentally friendly and easy-to-promote means of transportation is the strategic choice for my country to realize green transportation.

Compared with traditional vehicles, the power of low-speed electric vehicles is transmitted through flexible cables, and the arrangement of drive motors and transmissions is varied, and devices such as couplings and transmission shafts are omitted, so the structure is relatively simple. The motor drive system is generally composed of drive motor, control system, deceleration and transmission device, wheels, etc. It is one of the most critical parts of the entire low-speed electric vehicle. The motor drive system converts the energy of the power battery into the mechanical energy of the motor by receiving the commands sent by the control system, and transmits the power to the wheels through the transmission system to ensure the normal driving of the vehicle. In terms of performance, it is necessary to ensure that the vehicle can start and stop frequently, accelerate and decelerate, ride comfort, and passability in harsh environments. Therefore, there are higher requirements for the drive system of low-speed electric vehicles.

From the development of low-speed micro electric vehicles to the present, the technical upgrade of the motor drive system has mainly undergone three major technical transformations. For the structure and characteristics of small electric vehicles, the first generation designed a separate excitation motor and controller drive system with the advantages of speed regulation and commutation; as people's requirements for driving experience gradually increase, the permanent magnet DC brushless motor The second-generation drive system also came into being. Compared with the separately excited motor, the brushless DC motor has higher reliability and better driving experience, and the control method is simple, but there is a large torque ripple in the control method. This provides a direction for the improvement of the next-generation drive system; the third-generation drive system is closer to the requirements of electric vehicles. For the use characteristics of small electric vehicles, the AC electric control system based on vector control is more suitable. The stability is stronger, and the control algorithm and control strategy are more advanced. AC motor controllers are mainly divided into AC asynchronous and permanent magnet synchronous according to the control target motor. Due to the high price of rare earth permanent magnet materials, the price of permanent magnet synchronous motors and permanent magnet DC brushless motors used in the automotive field is much higher than that of AC asynchronous motors with the same power. Micro electric vehicles that are sensitive to price factors do not have sufficient reasons and time to accept these two types of motors as drive motors. At present, the most suitable drive system for low-speed electric vehicles is AC asynchronous motor (ACIM) and control system. The AC asynchronous motor drive system has been widely used due to its advantages of low cost, good reliability, and good speed regulation performance.

In order to take into account the reliability and comfort of vehicle operation, AC asynchronous motors using vector control are currently widely used in the field of low-speed electric vehicles. In the AC asynchronous motor vector control system, since the closed-loop control of the speed can improve the dynamic performance of the system, the speed of the motor is often measured by a photoelectric encoder or a tachogenerator. With the in-depth study of vector control technology, some sensors are expensive, require high installation accuracy, and their signals are also susceptible to electromagnetic interference. In drive systems with speed sensors, the speed feedback becomes unreliable. This not only increases the cost of the drive system, but also limits its application in harsh environments. At present, speed sensorless operation has become one of the important research contents in the field of AC drives, and has been widely used in vector control. Almost all large foreign inverter manufacturers have their own high-performance sensorless vector control inverters. product. At the same time, the research on flux observation and speed estimation in speed sensorless control can also be applied to other AC drive fields such as locomotive traction, frequency conversion air conditioner, etc. Therefore, the research on the speed sensorless vector control method of AC asynchronous motor has theoretical significance and important Value. The speed sensorless control strategy uses the stator voltage and current signals that are easily obtained, and through the analysis of the motor model in the static coordinate system, obtains the control algorithm of the speed and feeds it back to the control system, which not only realizes the high speed of the AC motor Performance control also reduces system hardware complexity and cost.

However, the following technical problems exist in the prior art: the actual application of the voltage space vector pulse width modulation (SVPWM) technology in the vector control will cause current distortion due to the influence of the dead time.

Contents of the invention

The purpose of the present invention is to overcome the above-mentioned deficiencies in the prior art, and provide a low-speed electric vehicle AC asynchronous motor vector control method and system, which can effectively reduce current distortion, running noise and torque ripple.

A kind of low-speed electric vehicle AC asynchronous motor vector control method proposed by the present invention is characterized in that said method comprises the steps:

Step 10, judge the sector where the voltage vector is located, specifically, according to the Clark transformation, obtain the vector control model in the two-phase stationary coordinate system, determine the sector where the voltage vector is located by analyzing U _α and U _β , and obtain three judgments according to formula 27 Vector U ₁ , U ₂ , U ₃ , judge the sector where the current composite vector is located according to the positive or negative of the three vectors;

Step 20, calculate the action time of the basic voltage vector, specifically after determining the sector where the current composite vector is located, calculate the conduction time, and define the intermediate variables X, Y, Z, the expression of which is formula 29:

In order to avoid errors in the calculation, when T<T ₁ +T ₂ , it is necessary to limit the on-time of the switch so that the maximum output voltage is within the inscribed circle of the vector hexagon to ensure safe operation. The constraints The condition is Formula 30:

Step 30, calculate the conduction time of each bridge arm and build the dead zone time, specifically according to the seven-segment SVPWM vector control algorithm and the basic vector conduction time, calculate the switching time points T _cm1 , T _cm2 , T cm2 corresponding to each sector _cm3 , define variables T _a , T _b , T _c as:

After calculation, the switching time points corresponding to each sector are shown in the following table:

Step 40, drive and control the motor, specifically, compare the triangular wave signal generator with each switching point to obtain three-way PWM waveform signals, and transmit the three-way PWM waves to the dead-time module, and the dead-time module uses transmission delay The module delays the low-level initial signal, subtracts the absolute value of the delayed signal from the high-level signal to obtain the dead time, and uses the dead time to multiply the initial high-level signal and the delayed signal respectively, so as to embed it into the timing signal generated by SVPWM , and finally the signal added to the dead time is used as the input control signal of the inverter to make the inverter generate the required voltage waveform to realize the drive control of the motor.

Preferably, in step 10: if U1>0, then A=1, otherwise A=0; if U2>0, then B=1, otherwise B=0; if U3>0, then C=1, otherwise C = 0, there are 8 combinations of A, B, and C, but in actual operation, U1, U2, and U3 will not be 0 or 1 at the same time, and there are actually 6 combinations that correspond to 6 sectors. The relation is Equation 28.

Sector＝C×4+B×2+A (Formula 28)

The present invention proposes a low-speed electric vehicle AC asynchronous motor vector control system, which is characterized in that the system includes:

The sector judgment device where the voltage vector is located is used to obtain the vector control model in the two-phase stationary coordinate system according to the Clark transformation, and determine the sector where the voltage vector is located by analyzing U _α and U _β , and obtain three judgment vectors U according to formula 27 ₁ , U ₂ , U ₃ , judge the sector where the current composite vector is located according to the positive and negative of the three vectors;

The basic voltage vector action time calculation device is used to calculate the conduction time after determining the sector where the current composite vector is located, and define the intermediate variables X, Y, Z, the expression of which is Formula 29:

In order to avoid errors in the calculation, when T<T ₁ +T ₂ , it is necessary to limit the on-time of the switch so that the maximum output voltage is within the inscribed circle of the vector hexagon to ensure safe operation. The constraints The condition is Formula 30:

The conduction time calculation and dead time construction device of each bridge arm is used to calculate the switching time points T _cm1 , T _cm2 , T _cm3 corresponding to each sector according to the seven-segment SVPWM vector control algorithm and the basic vector conduction time, Define variables T _a , T _b , T _c as:

After calculation, the switching time points corresponding to each sector are shown in the following table:

Step 40, the motor drive control device is used to compare the triangular wave signal generator with each switching point to obtain three-way PWM waveform signals, and transmit the three-way PWM waves to the dead-time module, and the dead-time module uses a transmission delay module Delay the low-level initial signal, subtract the absolute value of the delayed signal from the high-level signal to obtain the dead time, use the dead time to multiply the initial high-level signal and the delayed signal, and then embed it into the SVPWM generated timing signal, Finally, the signal added to the dead time is used as the input control signal of the inverter to make the inverter generate the required voltage waveform to realize the drive control of the motor.

Preferably, if U1>0, then A=1, otherwise A=0; if U2>0, then B=1, otherwise B=0; if U3>0, then C=1, otherwise C=0, A, There are 8 combinations of B and C, but in actual operation, U1, U2, and U3 will not be 0 or 1 at the same time. There are actually 6 combinations that correspond to 6 sectors. The relationship between the combination and the sector is formula 28 .

Sector＝C×4+B×2+A (Formula 28)

Description of drawings

In order to illustrate the technical solutions of the embodiments of the present invention more clearly, the accompanying drawings used in the embodiments will be briefly introduced below. Obviously, the accompanying drawings in the following description are only some embodiments of the present invention. Those of ordinary skill in the art can also obtain other drawings based on these drawings without any creative effort.

Fig. 1 is a sectional view of a squirrel-cage three-phase AC induction motor;

Fig. 2 is a vector control block diagram of an AC asynchronous motor;

Fig. 3 is a Clark transformation diagram;

Fig. 4 is a Park transform relationship diagram;

Fig. 5 is a model diagram of an AC asynchronous motor;

Fig. 6 is a three-phase power inverter structure;

Fig. 7 is a voltage vector diagram;

Fig. 8 is a schematic diagram of voltage vector synthesis;

Fig. 9 is a schematic diagram of voltage vector synthesis;

Fig. 10 is the error voltage vector diagram;

Fig. 11 is error voltage vector composite diagram;

Figure 12 is an ideal and actual switch state diagram;

Figure 13 is a trigger diagram of the PWM signal switch before and after compensation;

Fig. 14 is a block diagram of a sector judgment simulation module;

Fig. 15 is intermediate variable X, Y, Z calculation model;

Fig. 16 is a block diagram of the basic voltage vector action time simulation module;

Fig. 17 is a block diagram of dead zone time generation module;

Fig. 18 is a block diagram of Tcm1, Tcm2, Tcm3 computing modules;

Figure 19 is a block diagram of the 6-way PWM generation module.

detailed description

The present invention will be described in further detail below in conjunction with the accompanying drawings.

Three-phase AC asynchronous motor is a rotating machine controlled by three-phase AC. In order to realize adjustable speed, the voltage source generally uses an inverter composed of power switching components to generate an approximate sine wave with adjustable amplitude and frequency. AC voltage. A cross-sectional view of a three-phase AC induction motor with three pairs of poles is shown in Figure 1. There are a, b, and c three-phase windings embedded in the stator slot. In order to generate an approximately sinusoidal magnetic potential in the air gap, the windings are distributed. When the stator winding flows through the three-phase symmetrical sinusoidal alternating current with a phase angle difference of 120°, a magnetic field vector rotating at the frequency of the stator current will be formed in the air gap. According to the different rotor structure of the three-phase AC induction motor, it can be divided into squirrel-cage type and wound type, among which the squirrel-cage type is the most common. There is a conductor (made of copper or aluminum), which is longer than the iron core, and two end rings are used to short-circuit the conductor at both ends of the iron core to form a short-circuit winding. If the iron core is removed, the remaining winding is shaped like a squirrel cage, so it is called squirrel cage winding.

The so-called vector control is to transform the motor vector relationship expressed on the static coordinate system to the rotating coordinate axis system oriented by the air gap magnetic field, stator magnetic field or rotor magnetic field, so as to achieve the purpose of real-time control of the motor torque. Because the vector control method of the rotor magnetic field orientation is simple and easy, the decoupling is convenient, and the control precision is good, the present invention is based on the rotor magnetic field orientation. The three-phase stator currents I _A , I _B , and I _C of the AC motor are converted from the three-phase stationary coordinate system to the two-phase stationary coordinate system to obtain I _α , I _β . Then transform from the two-phase static coordinate system to the two-phase rotating coordinate system, and make the rotating shaft along the direction of the rotor flux linkage, and obtain the excitation current component I _d of the AC motor, and the torque current component I _q are respectively equivalent to the DC motor Excitation current and torque current. In this way, by controlling _Id and _Iq , the AC motor can be controlled according to the control method of the DC motor.

The basic block diagram of vector control is shown in Figure 2, and the entire control process is realized through software. Clark transformation is the transformation from three-phase stationary to two-phase stationary coordinate system, and Park transformation is the transformation from two-phase stationary coordinate system to two-phase rotating coordinate system. When performing coordinate transformation, the current rotor position θ must be known, which represents the clamping distance between axes and axes. angle, given by the rotor flux observer. The key to the realization lies in the construction of the rotor flux observer, that is, the determination of the rotor flux position angle, which involves the current decoupling problem of the AC asynchronous motor. Therefore, it is necessary to study the model of AC asynchronous motor, coordinate transformation and current decoupling on this basis.

The basis of vector control is the establishment of the mathematical model of the three-phase asynchronous motor, and the current, magnetic flux and speed of the three-phase AC motor all affect each other. In addition, the stator and rotor of the three-phase motor are equivalent to three windings respectively, and each winding has its own electromagnetic inertia when generating magnetic flux, plus the electromechanical inertia of the motion system, the hysteresis factor of the frequency conversion device, etc., these factors are It is determined that the asynchronous motor is a high-order, nonlinear, strongly coupled multivariable system.

If there is no coordinate axis transformation, the vector control will not be realized, because the asynchronous motor is strongly coupled in the three-phase stationary coordinate system, and there are many and complex variables to be controlled. When realizing the vector control, it is simplified by decoupling the asynchronous motor. The variables to be controlled are first transformed from the three-phase stationary coordinate system variables to the two-phase stationary coordinate system variables through the Clarke transformation, and then transformed from the two-phase stationary reference coordinate system variables to the two-phase rotating reference coordinate system variables through the Park transformation. The speed vector and flux linkage vector of the motor are completely decoupled, and they can be adjusted separately. And the current before and after the transformation produces an equivalent rotating magnetic field.

The Clark transformation refers to the transformation from a stationary three-phase coordinate system to a stationary two-phase coordinate system. Taking constant power current conversion as an example, first consider the conversion of three-phase stationary windings A, B, C to two-phase stationary windings α, β, referred to as 3S/2S conversion. Its transformation relationship is shown in Figure 3. The conversion formula is:

If the three-phase winding is Y-shaped and there is no neutral wire, then there is i _A +i _B +i _C ＝0, which can be obtained after being inserted into the above formula:

Similarly, Clark transform can also be used in voltage vector and flux linkage vector.

Park transformation refers to the transformation of a two-phase stationary coordinate system into a two-phase rotating coordinate system. Taking the current transformation as an example here, first consider the transformation from the two-phase stationary windings α, β to the two-phase stationary windings d, q, referred to as 2S/2R transformation. Its transformation relationship is shown in Figure 4. It can be seen from Figure 4 that there is the following relationship between i _α , i _β and _id , i _q :

Converted to a matrix form:

Further converted to an inverse matrix:

Similarly, Park transformation can also be used in voltage vector and flux linkage vector.

Mathematical model is the basis of asynchronous motor research, it is mainly composed of voltage equation, flux equation, torque equation and motion equation. When implementing vector control for AC asynchronous motors, according to the different coordinate systems, it is necessary to analyze the motor models in the three-phase stationary coordinate system, the two-phase stationary coordinate system and the two-phase rotating coordinate system respectively.

When studying multivariable mathematical models of asynchronous motors, the following assumptions are often made: Neglecting space harmonics, assuming that the three-phase windings of the motor have a symmetrical spatial difference in electric angle, the generated magnetomotive force is distributed along the air gap circle according to the sinusoidal law. Neglecting magnetic saturation, the winding has constant self and mutual inductance. The resistance of each winding is constant independent of frequency and temperature. The motor rotor is equivalent to a wound rotor, which is converted to the stator side, and the converted three-phase winding turns are equal. In this way, the physical model of the three-phase asynchronous motor is shown in Figure 5. The three-phase winding axes A, B, and C in Fig. 5 are fixed in space, and the rotor winding axes a, b, and c rotate with the rotor with the A axis as the reference coordinate axis, and the electrical angle between the rotor axis a and the stator axis A is Spatial angular displacement variable. It is stipulated that the positive direction of each winding voltage, current, and flux linkage conforms to the convention of the motor and the right-hand screw rule. The mathematical model of the three-phase asynchronous motor in the three-phase stationary coordinate system can be described by the voltage, flux linkage, torque and motion equation of the system.

(1) Voltage equation:

The voltage balance equation of the three-phase stator winding is:

The voltage equation of the corresponding three-phase rotor winding converted to the stator side is:

In Equation 5 and Equation 6: u _A , u _B , u _C , u _a , u _b , u _c are the instantaneous values of stator and rotor phase voltages, i _A , i _B , i _C , i _a , i _b , i _c ψ _A , ψ _B , ψ _C , ψ _a , ψ _b , ψ _c are the full flux linkages of the three-phase stator and rotor, R ₁ , R ₂ are the stator and rotor winding resistances.

(2) Flux linkage equation:

The flux linkage of each phase winding of AC asynchronous motor includes its own self-inductance flux linkage and mutual induction flux linkage of other windings to him, and its flux linkage equation can be expressed as the following form:

In formula 7: ψ _A , ψ _B , ψ _C are the flux linkages of the three-phase stator windings, ψ _a , ψ _b , ψ _c are the flux linkages of the three-phase rotor windings; L _A , L _B , L _C are the three-phase stator windings The self-inductance of the windings, L _a , L _b , L _c are the self-inductances of the three-phase rotor windings, i _A , i _B , i _C are the currents of the three-phase stator windings, _{ia , i b , ic} are the three-phase rotor windings The current of the winding, L _Ab , L _AB .... is the mutual inductance of each phase winding.

There are two types of magnetic flux interlinked with the motor windings: one is the leakage flux that only interlinks with a certain phase winding and does not pass through the air gap, and the other is the interphase mutual induction flux that passes through the air gap. is the main one. The mutual inductance flux is divided into two categories: (1) The position between the three phases of the stator and the three phases of the rotor is fixed, so the mutual inductance is a constant value; (2) The position between any phase of the stator and any phase of the rotor is changing, and the mutual inductance is the q function of the angular displacement.

(3) Electromagnetic torque equation:

Assuming that the linear magnetic circuit and the magnetomotive force are distributed sinusoidally in space, according to the principle of conservation of electromechanical energy, the expression of the electromagnetic torque Te can be obtained as follows:

The above formula shows that T _e is a function of the stator current, rotor current and rotor angle, that is, the drag torque is a multivariable, nonlinear and strongly coupled function.

(4) Motion equation:

In general, the resistance torque damping and torsional elastic torque in the electric drive system are ignored, and when the load is a constant torque load, the motion equation of the motor is:

In Formula 9: TL is the load torque; J is the moment of inertia of the unit.

The model of the three-phase asynchronous motor in the two-phase stationary coordinate system is also the model in the α-β coordinate system. Transform the voltage, current, and flux linkage in the three-phase static coordinate (A-B-C) into the two-phase static coordinate system α-β by Clark, and then the mathematical model of the three-phase asynchronous motor in the two-phase static coordinate system can be obtained.

(1) Voltage equation:

For a motor with a squirrel-cage rotor, u _α2 = u _β2

(2) Flux linkage equation:

(3) Torque equation:

T _e ＝n _p L _m (i _β1 i _α2 -i _α1 i _β2 ) (Formula 12)

The model of the three-phase asynchronous motor in the two-phase rotating coordinate system is also the model in the d-q coordinate system. The model of the three-phase asynchronous motor in the two-phase rotating coordinate system can be obtained by transforming the electric quantity in the two-phase stationary coordinate (α-β) to the two-phase rotating coordinate system (d-q) through Park.

(1) Voltage equation:

(2) Flux linkage equation:

(3) Torque equation:

T _e =n _p L _m (i _d1 i _q2 -i _q1 i _d2 ) (Formula 15)

Taking the AC asynchronous motor used in low-speed electric vehicles as the research object above, the relationship between flux linkage, voltage, current and other variables is analyzed based on the motor theory. Combined with the principle of coordinate transformation, the mathematical models of asynchronous motors in different coordinate systems are deduced. Under the rotor field orientation, the vector control equation of the system is established, and the decoupling of the system is realized, which paves the way for the following research on the motor control strategy.

The principle of SVPWM technology in vector control will be deeply analyzed below, and the implementation steps and methods of SVPWM control algorithm will be given. And analyze the influence of dead zone time in the practical application of SVPWM, and propose a dead zone compensation method based on time.

Analysis and Realization of SVPWM Control

Space Vector Pulse Width Modulation (SVPWM), actually corresponds to a special combination of switch trigger sequence and pulse width of the power device of the three-phase voltage source inverter in the permanent magnet synchronous motor or AC induction motor. The trigger sequence and combination will generate three-phase sinusoidal currents with a phase difference of 120° electrical angle and less distortion in the stator coils. Both practice and theory can prove that compared with the direct sinusoidal pulse width modulation (SPWM) technology, SVPWM will generate less harmonics in the output voltage or the current in the motor coil, which improves the DC power supply of the power inverter. Power utilization efficiency.

The following is the structure of a typical three-phase voltage source inverter, as shown in Figure 6. In Figure 6, V _a , V _b , and V _c are the voltage outputs of the inverter, and Q ₁ to Q ₆ are 6 power transistors, which are respectively controlled by VT ₁ , VT ₂ , VT ₃ , VT ₄ , VT ₅ , and VT ₆ These 6 control signals are controlled. When the power transistor in the upper half of the inverter bridge is turned on, that is, when VT ₁ , VT ₂ or VT ₃ is 1, the corresponding power transistor in the lower half is turned off (VT ₄ , VT ₅ or VT ₆ is 0) so The states of a, b, and c being 0 or 1 will determine the waveforms of the three-phase output voltages of Va, Vb, and Vc.

The relationship between the line voltage vector, phase voltage vector and switch variable vector output by the inverter bridge can be expressed by Equation 16 and Equation 17:

In the above formula: Vdc is the DC supply voltage of the voltage source inverter.

It is not difficult to see that because the switch variable vector [a, b, c] has 8 different combination values, that is, the switching states of the three power transistors in the upper half of the inverter bridge have 8 different combinations, so the output There are 8 corresponding combinations of phase voltage and line voltage. According to different switch and phase voltage states, the space voltage vector under the α-β axis can be obtained through Clark transformation, and the conversion formula can be expressed by the following formula 18 and formula 19. In Table 1, V _AN , V _BN , and V _CN represent the phase voltages of the three outputs, and V _α and V _β represent the space voltage vectors under the α-β axis system.

V _α =V _AN (Equation 18)

Table 1 The phase voltage of the switching state of the power tube and the coordinate components of the α-β axis

The eight switch states in Table 1 correspond to eight basic voltage vectors. These eight voltage vectors divide their space into six areas with equal voltage amplitudes and 60° phase differences, as shown in Figure 7. It can be seen from Figure 7 that any voltage vector can be obtained by combining two adjacent basic voltage vectors in its area. After determining the sector where the output voltage is located, calculate the action time of the two adjacent basic voltage vectors T ₁ and T ₂ are used to realize the output of the synthesized vector, and when the PWM period is subdivided sufficiently small, the track of the voltage vector will be approximately a circle. However, the trajectories of the voltage vector and the flux vector are coincident, so the flux trajectory is also an approximate circle. The SVPWM algorithm realizes the vector control of the motor by continuously adjusting the PWM output and duty cycle in different sectors.

Figure 8 is a schematic diagram of voltage vector synthesis. Taking the first sector as an example, the synthesized voltage vector U _s is composed of U ₀ and U ₆₀ , T ₁ and T ₂ represent the action time of U ₀ and U ₆₀ respectively, and T represents the carrier wave cycle time. According to the principle of vector synthesis, the formula (20) can be obtained:

Using the formula (Formula 20) and according to the trigonometric sine law, it can be analyzed (Formula 21):

Then T1 and T2 are calculated as:

After determining the time of T1 and T2, calculate the action time of the inserted zero vector T0 according to formula (Formula 23).

T ₀ =TT ₁ -T ₂ (Equation 23)

Through the above method, calculate T ₁ and T ₂ corresponding to the frequency required by the system, so as to achieve the purpose of frequency conversion. The principle of adding a zero vector is: to minimize the switching times of the power switching device. In order to make the flux linkage move smoothly, the zero vector is divided into several parts on average, and multiple points are inserted into the flux linkage trajectory. The total action time T ₀ remains unchanged. Thereby reducing the torque ripple of the motor. According to the above analysis, T ₀ , T ₁ , and T ₂ are evenly divided to realize this function through the PWM module of the MCU. The waveform of the PWM output is shown in Figure 9. The PWM output method in Figure 9 is generally called seven-segment SVPWM. This method effectively divides the 0 vector evenly, making the output of the synthesized vector more stable and smooth, and the synthesized flux linkage is also closer to a circular flux linkage, which can achieve Better SVPMW control effect.

Research and Design of SVPMW Dead Zone Compensation

The seven-segment SVPWM control needs to continuously switch the state of the power tube to achieve. In actual use, due to the existence of the inherent storage time of the power device, the turn-on time is shorter than the turn-off time, and it is easy for two power tubes with the same phase bridge arm to be turned on complementary In order to avoid the short-circuit fault of this situation, the turn-on signal is usually sent after a delay of a dead time. The turn-on, turn-off time and dead zone settings of the power tube cause the actual output voltage waveform of the power tube to produce nonlinear distortion compared with the ideal given voltage waveform, causing dead zone effects such as motor current waveform distortion and torque ripple, especially Affect the low-speed performance of the motor. In order to reduce the impact caused by the nonlinear distortion of the motor current waveform, a dead zone compensation method based on time compensation is designed to compensate for the impact of the inherent delay of the power tube device on the entire system.

The structure of the three-phase motor inverter is shown in Figure 6. As far as the output voltage of the single-phase inverter is concerned, the dead zone effect produces a series of distorted pulses, and the pulse polarity is related to the current polarity. Since the three-phase currents are 120° electrical different from each other, six error voltage vectors are formed in the motor, as shown in Figure 10.

When equal-amplitude conversion is used, the amplitude of the error voltage vector is 4VdcTe/3T, where Vdc is the DC bus voltage, T is the carrier period, and the error time Te is:

T _e =T _d +T _on -T _off (Formula 24)

Among them, Td is the dead time; Ton is the time required for the power tube to be turned on; Toff is the time required for the power tube to be turned off.

The direction of the vector of the error voltage is determined by the polarity of the three-phase current. It is defined that the current flowing into the motor winding is positive. In Figure 10, the corresponding relationship between the vector and the current polarity is: the number 1 means that the phase current is positive, and 0 means that the phase current Negative, according to the combination sequence of abc phase, corresponding to the voltage vector in Figure 10.

Taking the sector where the current ia, ib, ic are positive and negative as an example, the formula for calculating the magnitude and phase of the synthesized voltage vector is derived. The angular range of the current vector in this polarity sector is [330°, 360°] and [0, 30°]. θ represents the angle between the given voltage vector and the α axis, and the voltage vector synthesis is shown in Figure 3-6. Figure 11(a) shows that θ is within the angle range of [330°, 360°], and Figure 11(b) shows that θ is within the angle range of [0°, 30°]. It can be seen that when θ = 0, the Usyn amplitude is the smallest (Formula 24). In other current polarity mountainous areas, the same method can be used to analyze that the influence of dead zone distortion is repeatable.

|U _syn |＝|U _s |-|ΔU ₄ | (Formula 25)

Taking the a-phase bridge arm as an example, Figure 3-7 is the waveform diagram of the theoretical starting signal state and the actual switching state within one carrier cycle. A+L, A-L are ideal power tube switching signals, and A+R, A-R are actual The switch signal diagram of the power tube, where the shaded part is the state where the two tubes are not conducting, at this time the output voltage of the bridge arm is determined by the freewheeling diode. When ia>0, the diode of the lower bridge arm is turned on, which is equivalent to the conduction of the switch tube of the lower bridge arm. During the actual switch, the shaded part of A+R is at low level, and A-R is at high level. After analysis, the actual The turn-on time is shortened by Te from the ideal turn-on time, while the actual turn-on time of the lower tube is extended by Te. All that needs to be done is to extend the turn-on time of the upper tube by Te and correspondingly shorten the turn-on time of the lower tube by Te, so that the actual switch time and the ideal give The fixed time is consistent to ensure that the output voltage value is equal to the ideal value. In the same way, it can be analyzed that when ia<0, the ideal opening time of the upper tube is shortened to Te. Time compensation is realized by adjusting the on-time of phase a PWM in one modulation cycle, and the compensation method of phase b and c is the same as that of phase a.

The seven-segment SVPWM method is realized by vector synthesis, and a more simplified algorithm can be obtained due to the insertion of two zero vectors U0 and U7. Analyze positive and negative current polarity with abc. As shown in FIG. 12 , according to the above-mentioned time compensation method, the Te time is extended when A+ is high, and the Te time is shortened by B+ and C+. Comprehensive three-phase results: the action time of vector U4 increases by 2Te compared with the ideal given time, the action time of vector U6 does not change, and the action time of zero vector decreases by 2Te. Keeping the switching tube signal of the bc phase unchanged, only extending the high level of A+ by 2Te can also achieve the same compensation effect, which effectively simplifies the algorithm.

t _a represents the triggering time of the switch tube signal of the upper bridge arm of phase a, and t' _a indicates the triggering time of the signal of the switch tube of the upper bridge arm of phase a after compensation, which can be obtained through the above analysis (Equation 26).

t' _a =t _a -T _e (Formula 26)

When the abc current polarity is positive or negative, the time compensation of other voltage vector synthesis sectors is analyzed, and the same result is obtained, and then the other current polarity sectors are analyzed, and the final compensation method is shown in Table 2.

Table 2 The phase voltage of the switching state of the power tube and the coordinate components of the α-β axis

According to the current polarity in Table 2, time compensation is performed on different phase lines in different current polarity intervals to realize the time-based SVPWM dead zone compensation algorithm.

In order to verify the effectiveness of the SVPWM control algorithm, a system model is established on the basis of the MATLAB/Simulink platform, and the simulation analysis is carried out. This model is also the basic algorithm of the sensorless vector control algorithm. The steps to realize the construction of the SVPWM algorithm simulation model are as follows:

Construction of the sector judgment module where the voltage vector is located

According to the above-mentioned Clark transformation, the vector control model in the two-phase stationary coordinate system can be obtained, and the sector where the voltage vector is located can be determined by analyzing U _α and U _β . Combined with the above analysis, three judgment vectors U ₁ , U ₂ , U ₃ (Formula 26) are obtained, and the sector where the current composite vector is located can be judged according to the positive or negative of the three vectors.

Redefine, if U1>0, then A=1, otherwise A=0. If U2>0, then B=1, otherwise B=0. If U3>0, then C=1, otherwise C=0. It can be seen that there are 8 combinations of A, B, and C, but in actual operation, U1, U2, and U3 will not be 0 or 1 at the same time. Therefore, the actual combination has 6 total combinations which correspond to exactly 6 sectors. The relationship between combination and sector (Formula 28) determines:

Sector＝C×4+B×2+A (Formula 28)

According to (Formula 27), the sector where the current vector is located can be determined, and the established sector judgment simulation module is shown in Figure 14:

Basic voltage vector action time calculation module construction

After determining the sector where the current composite vector is located, the on-time needs to be calculated. The switch on-time in different sectors has been calculated above. For the convenience of modeling, intermediate variables X, Y, and Z are defined. The expressions are:

In order to avoid errors in the calculation, when T<T ₁ +T ₂ , it is necessary to limit the on-time of the switch so that the maximum output voltage is within the inscribed circle of the vector hexagon to ensure safe operation. Its constraints are:

Combining the above calculation results, establish a simulation module as shown in Figure 15 that uses X, Y, and Z as intermediate variables, and calculates the conduction times T ₁ and T ₂ of adjacent basic vectors according to the intermediate variables, as shown in Figure 16 .

Calculation of conduction time of each bridge arm and construction of dead time module

According to the above analysis of the seven-segment SVPWM vector control algorithm and the basic vector conduction time, in order to make the algorithm run smoothly, it is necessary to design the switching time points T _cm1 , T _cm2 , and T _cm3 of the switching tubes of each bridge arm, and define the variable T _a , T _b , T _c are:

After calculation, the switching time points corresponding to each sector are shown in Table 3:

Table 3 Switching schedule of each sector

Three channels of PWM waveform signals are obtained by comparing the triangular wave signal generator with each switching point, as shown in Figure 19. These three channels of PWM waves will be transmitted to the dead time module. The structure of the dead time module is shown in Figure 17. It uses the transmission delay module to delay the low-level initial signal. The absolute value of the high-level signal minus the delay signal is the dead time. The dead time is multiplied with the initial high-level signal and the delay signal respectively, so as to be embedded into the timing signal generated by SVPWM. Finally, the signal added to the dead time is used as the input control signal of the inverter to make the inverter generate the required voltage waveform, and finally the drive control of the motor can be realized.

The present invention combines the power characteristics of low-speed electric vehicle power supply and asynchronous motor, establishes a DC-AC inverter control system with a high-efficiency, easy-to-control SVPWM control algorithm, and establishes a simulation model for the control algorithm, including The establishment of the sector judgment module, the switch conduction time module, the switch switching point module and the control signal generation module converts the motor target control quantity into the corresponding input voltage value to realize the control of the motor to the target quantity. On the basis of realizing SVPWM control, a dead zone compensation method based on time compensation is designed, which effectively improves the stability of SVPWM control and reduces current distortion and torque ripple.

In the low-speed electric vehicle drive system, the current rotor position is usually determined by the feedback information of the encoder on the motor, but due to the limitation of the installation position of the encoder, the working environment is relatively harsh, and there is a risk of failure, and once a failure occurs, the lighter The performance of the system will be reduced, and the serious one will directly cause the car to completely idle, and can only wait for rescue. Therefore, it is very necessary to study the fault-tolerant control of AC asynchronous motor drive system. At present, there are many redundant protection measures for motor drive systems, and they are being widely used in situations where continuous operation of the motor is required. However, some of these methods have the problem of short-term electromagnetic torque shock, and if the system continues to operate after a fault, its performance will be permanently degraded. The most common ones are household appliances, such as air conditioners and fans. But for motor drive control systems used in low-speed electric vehicles, degenerated control performance is better than no control action. Therefore, a redundant control scheme for speed sensor failure is necessary. In order to realize fault redundant control, a sensorless vector control scheme is required. The present invention adopts a sensorless vector method based on an adaptive sliding mode observer, and determines the sliding mode control mechanism by estimating the current and flux deviation, and makes the control The state of the system is finally stable on the designed sliding mode hyperplane. Sliding mode control has good dynamic response, and is also outstanding in robustness and simplicity.

Speed Observer Model of AC Induction Motor

The sliding mode observer is a closed-loop flux estimation method based on an ideal model, which realizes the estimation of the rotor flux and rotational speed through the detected stator current and stator voltage. In the stationary coordinate system, the state equation of the induction motor is:

In the formula: λ _dr , λ _qr are d, q axis rotor magnetic flux components; i _ds , i _qs are d, q axis rotor magnetic flux components stator current components; ω _r is rotor speed; R _s , R _r are constant Rotor resistance L _s , L _r , L _m are the stator inductance, rotor inductance and stator-rotor mutual inductance; T _r is the motor rotor time constant; σ is the total leakage inductance coefficient.

In the speed sensorless control, it is necessary to observe the instantaneous magnetic flux value and the rotor speed. The speed observer of the present invention belongs to the closed-loop speed observer, and its specific design is as follows:

In the formula: is the estimated rotor flux component; is the estimated stator current component; Estimate the speed for the rotor.

Define the difference between estimated and true components as Equation 35:

According to this, Equation 36 can be obtained:

According to the induction motor state equation, state estimation equation and difference equation, the switching surface of the sliding mode variable structure control can be determined by formula 37.

Stability Analysis of Sliding Mode Observer

Current Stability Analysis

The Lyapunov function that defines constant positive is:

Its differential equation is:

According to (Equation 36):

It can be seen from the characteristics of Li's equation that if the conditions I'<0 and I≥0 are satisfied, the current regulation is stable. Therefore, if the formula 40 is less than 0, the simplified formula can be obtained (formula 41):

If order According to the sliding mode variable structure control theory, the sufficient condition for the stability of the model is:

According to formula 42, if the K value is large enough to meet this sufficient condition, the estimated current component will converge to the true values i _ds and i _qs , and the difference between them will approach zero.

Analysis of Rotor Flux Stability

when In the state model, the curve is the condition for current stability. in continuous control In , according to the sliding mode variable structure theory, the stability condition should be written as According to Equation 36, it can be obtained that:

Will Putting it into Equation 36, we can get:

The roots of the system described by Equation 43 are:

According to the stability analysis of control theory, the system is stable. Therefore, in the feedback condition Under this condition, the velocity observer will be by motion, so formula 44 is stable.

is a constant.

At steady state, Equation 36 can be rewritten as:

because is any value, so the stable condition satisfying formula 45 is c _d =c _q =0, and

Therefore, in order to ensure the correctness of the velocity observer, the control variable must meet the condition of

If Equation 46 is used as the feedback quantity of the observer, only one gain parameter needs to be determined, and a fast and robust response of the rotor flux and torque can be achieved. Compared with the traditional sliding mode speed observer, it reduces The complexity of observer parameter design. In actual motor control, in order to prevent the high-frequency components in the estimated speed from adversely affecting the operation of the motor, a low-pass filter is added after the estimated ideal speed, and its output is used as an estimate of the actual speed And used for system control.

In the above analysis, K must be greater than or equal to ω _max , and the impact of K value on system performance will be discussed below.

(1) Increase the K value, The trembling will get bigger.

(2) Increasing the K value will speed up the calculation of the rotor flux and motor torque. In fact, the observer will make the system converge to the correct flux value. If K is very large, the convergence rate is theoretically infinite.

In the traditional sliding mode velocity observer, the K value is a fixed value suitable for the whole frequency range. Considering the above two points of analysis, if in the process of speed regulation, K value can be based on If it is automatically adjusted for changes, the system will have better and more stable performance in the entire speed range, especially the tremor phenomenon in the high frequency band can be well suppressed. Considering the above problems, a PI controller can be added to the system to ensure that the observer can respond according to The change adjustment parameter K. The PI controller is designed as follows:

Combining the advantages of sliding mode variable structure technology and PI controller, the expression of estimated velocity can be obtained as:

At this point, formula 48 can be equivalent to:

In this way, the K _d value will be automatically adjusted as the frequency changes, which can not only effectively improve the overall performance of the system, but also effectively weaken the trembling phenomenon in the high frequency band.

Aiming at the problem that the speed sensor of the low-speed electric vehicle controller is easily damaged, a sensorless control algorithm based on an adaptive sliding mode observer is proposed to realize the fault-tolerant control of the speed sensor. The observer is simple in structure, has good robustness to parameters, and has a wide speed estimation range. The observer is applied to the vector control of indirect rotor field orientation. Both have good current tracking ability and speed estimation ability, and the parameters of the sliding mode speed observer can follow the adaptive change of the estimated speed of the system, which not only simplifies the design of the observer, but also effectively eliminates the Inherent high frequency jitter problem.

It is to be understood that the described embodiments of the present invention may be realized by hardware, software, firmware, middleware, microcode or any combination thereof. For hardware implementation, the processing unit can be implemented in one or more Application Specific Integrated Circuits (ASICs), Digital Signal Processors (DSPs), Digital Signal Processing Devices (DSPDs), Programmable Logic Devices (PLDs), Field Programmable Gate Arrays ( FPGA), processors, controllers, microcontrollers, microprocessors, other electronic units designed to perform the functions described in the present invention, or combinations thereof. When the embodiments are implemented in software, firmware, middleware or microcode, program code or code segments, they may be stored in a machine-readable medium such as a memory component.

It will be apparent to those skilled in the art that the present invention is not limited to the details of the exemplary embodiments described above, but that the invention can be embodied in other specific forms without departing from the spirit or essential characteristics of the invention. Accordingly, the embodiments should be regarded in all points of view as exemplary and not restrictive, the scope of the invention being defined by the appended claims rather than the foregoing description, and it is therefore intended that the scope of the invention be defined by the appended claims rather than by the foregoing description. All changes within the meaning and range of equivalents of the elements are embraced in the present invention. Any reference sign in a claim should not be construed as limiting the claim concerned.

In addition, it should be understood that although this specification is described according to implementation modes, not each implementation mode only contains an independent technical solution, and this description in the specification is only for clarity, and those skilled in the art should take the specification as a whole , the technical solutions in the various embodiments can also be properly combined to form other implementations that can be understood by those skilled in the art.

Claims (4)

1. a kind of low-speed electronic automobile AC induction motor vector control method, it is characterised in that methods described includes following step Suddenly：

Step 10, sector where judging voltage vector, specially converts according to Clark, obtains the arrow under two-phase rest frame Controlling model is measured, by analyzing U_α、U_βSector where determining voltage vector, show that three judge vector U according to formula 27₁、U₂、 U₃, according to the sector where the current resultant vector of positive negative judgement of three vectors；

Step 20, basic voltage vectors action time is calculated, behind sector where specially determining current resultant vector, calculates conducting Time, intermediate variable X, Y, Z are defined, its expression formula is formula 29：

In order to avoid the error in calculating, work as T<T₁+T₂When, it is necessary to limit switch conduction times, the maximum that makes it export Voltage is safe for operation to ensure in the inscribed circle of vector hexagon, and its constraints is formula 30：

Step 30, calculate the ON time of each bridge arm and build dead time, specially calculated according to seven segmentation SVPWM vector controlleds Method and basic vector ON time, calculate switching time point T corresponding to each sector_cm1、T_cm2、T_cm3, defined variable T_a、T_b、T_c For：

After calculating, switching time corresponding to each sector, point was as shown in the table：

Step 40, control is driven to motor, specifically by triangular signal generator compared with each switching point Three tunnel PWM waveforms are obtained, the three roads PWM ripples are passed to dead time module, dead time module transmission delay mould Block postpones low level initial signal, and what high level signal subtracted postpones signal thoroughly deserves dead time, utilizes dead time Multiplication is carried out with initial high level signal, postpones signal respectively, so as to be embedded into SVPWM generation clock signals, finally will Add the signal of dead time makes inverter produce required voltage waveform, realization pair as the input control signal of inverter The drive control of motor.

2. low-speed electronic automobile AC induction motor vector control method according to claim 1, it is characterised in that：In step In rapid 10：If U1>0, then A=1, otherwise A=0；If U2>0, then B=1, otherwise B=0；If U3>0, then C=1, otherwise C=0, A, B, C share 8 kinds of combinations, but U1, U2, U3 will not be 0 simultaneously or be 1 simultaneously in actual motion, actually have 6 kinds of combinations just right 6 sectors are answered, relational expression corresponding to combination and sector is formula 28.

Sector=C × 4+B × 2+A (formula 28).

3. a kind of low-speed electronic automobile AC Motor Vector Control System, it is characterised in that the system includes：

Sector judgment means, for being converted according to Clark, obtain the vector controlled under two-phase rest frame where voltage vector Model, by analyzing U_α、U_βSector where determining voltage vector, show that three judge vector U according to formula 27₁、U₂、U₃, according to Sector where the current resultant vector of positive negative judgement of three vectors；

Basic voltage vectors action time computing device, behind sector where determining current resultant vector, ON time is calculated, Intermediate variable X, Y, Z are defined, its expression formula is formula 29：

In order to avoid the error in calculating, work as T<T₁+T₂When, it is necessary to limit switch conduction times, the maximum that makes it export Voltage is safe for operation to ensure in the inscribed circle of vector hexagon, and its constraints is formula 30：

The ON time of each bridge arm calculates and dead time builds device, for according to seven segmentation SVPWM vector control algorithms and Basic vector ON time, calculate switching time point T corresponding to each sector_cm1、T_cm2、T_cm3, defined variable T_a、T_b、T_cFor：

After calculating, switching time corresponding to each sector, point was as shown in the table：

Step 40, motor drive control device, for being obtained by triangular signal generator compared with each switching point Three tunnel PWM waveforms, the three roads PWM ripples are passed to dead time module, dead time module transmission delay module is prolonged Slow low level initial signal, what high level signal subtracted postpones signal thoroughly deserves dead time, is distinguished using dead time Multiplication is carried out with initial high level signal, postpones signal, so as to be embedded into SVPWM generation clock signals, will finally be added The signal of dead time makes inverter produce required voltage waveform as the input control signal of inverter, realizes to motor Drive control.

4. low-speed electronic automobile AC Motor Vector Control System according to claim 3, it is characterised in that：If U1 >0, then A=1, otherwise A=0；If U2>0, then B=1, otherwise B=0；If U3>0, then C=1, otherwise C=0, A, B, C share 8 kinds Combination, but U1, U2, U3 will not be 0 simultaneously or be 1 simultaneously in actual motion, actually there is 6 kinds of combinations corresponding 6 sectors just, group Relational expression corresponding to conjunction and sector is formula 28.

Sector=C × 4+B × 2+A (formula 28).