CN104022993B - SLM method for lowering peak-to-average power ratio of SFBC MIMO-OFDM system - Google Patents

Abstract

The invention discloses an SLM method for lowering the peak-to-average power ratio of an SFBC MIMO-OFDM system. Information source bits of each antenna pass through a baseband modulation unit and a series-parallel-connection conversion unit, are subjected to different phase rotation, and then are subjected to IFFT modulation to obtain time domain signals, time domain odd-even signals are obtained through the circulation shift property of time domain signals of an FFT, and the time domain odd-even signals are subjected to different lengths of time domain circulation shifts and then subjected to time domain equivalent SFBC encoding to obtain alternative sequence pairs with different PAPRs. On the basis of a traditional SLM algorithm, more alternative sequence sets with different PAPRs can be obtained only through circulation shift and phase rotation of the time domain signals, and the PAPR inhibition performance is improved. Meanwhile, a receiving end recovers the odd-even signals, circulation shift factors and phase rotation factors by comparing the distance between a reverse rotation sequence and a nearest signal constellation point, and sideband secondary information does not need to be transmitted.

Description

A SLM Method for Reducing Peak-to-Average Power Ratio of SFBC MIMO-OFDM System

technical field

The invention belongs to the technical field of wireless communication, in particular to an SLM method for reducing the peak-to-average power ratio of an SFBC MIMO-OFDM system.

Background technique

As we all know, the multiple-input multiple-output orthogonal frequency division multiplexing (MIMO-OFDM for short) system obtains diversity gain by using frequency, time and different antennas, which can effectively resist multipath and noise in wireless communication, and become the future mobile One of the main candidate technologies for multimedia communication.

The MIMO-OFDM signal is a multi-carrier modulation signal, and one of its main disadvantages is that the signal peak power to average power ratio (PAPR for short) is relatively high. At present, a mainstream method for reduced space-frequency block coding (SFBC) MIMO-OFDM system is selective mapping method (abbreviated as SLM method). Without loss of generality, a two transmit antenna SFBC MIMO-OFDM system will be discussed below. The principle of the transmitting end of the traditional SLM method is shown in Figure 1, and the principle of the receiving end is shown in Figure 2. The original OFDM signal of each antenna is multiplied by U rotation phase sequences with a modulus value of 1, and U numbers representing the same information are obtained. Output the signal, and then perform SFBC on these U signals to obtain two signals. These two signals are used as a sequence pair to perform IFFT modulation to obtain corresponding time-domain candidate signals. Calculate the PAPR of each candidate sequence and select the larger One of the sequence pairs is used as the PAPR of this sequence pair, and a minimum value is selected among the PAPRs of all sequence pairs as the PAPR of the entire SFBC MIMO-OFDM system, and this sequence pair is used as a transmission signal. In order to be able to correctly demodulate the received signal at the receiving end, the sideband side information of the used phase factor must be transmitted at the transmitting end. In the SFBC MIMO-OFDM system, one disadvantage of the traditional SLM method is that in order to obtain better PAPR suppression performance, it is usually necessary to select a sequence pair with the smallest PAPR from multiple candidate pairs for transmission, and each candidate signal Both need to be obtained by IFFT modulation, and the computational complexity of the algorithm is relatively high. Another shortcoming of the traditional SLM method is that in order for the system to restore the original signal at the receiving end, the transmitter needs to transmit the sideband side information of the phase factor used, and the transmission of the sideband side information reduces the spectrum utilization efficiency of the system.

Contents of the invention

The purpose of the present invention is to overcome the deficiencies of the prior art, to provide a method under the number of IFFT operations of the traditional SLM algorithm, only need to operate such as cyclic shift and phase rotation of the time domain signal to obtain multiple sequences with different PAPR Set, the receiving end restores the parity signal, cyclic shift factor and phase rotation factor by comparing the distance between the reverse rotation sequence and its nearest signal constellation point, which greatly reduces the computational complexity of the system and reduces the peak-to-average power ratio of the SFBC MIMO-OFDM system The SLM method.

The purpose of the present invention is to realize by following technical scheme: a kind of SLM method that reduces the peak-to-average power ratio of SFBC MIMO-OFDM system, comprises two parts of transmission processing process and receiving processing process, and the specific flow of described transmission processing process is:

S101: The source bits of each antenna pass through the baseband modulation unit and the serial-to-parallel conversion unit to obtain the original frequency domain signal X, and the original frequency domain signal X enters the phase rotation sequence generator for different phase rotations and performs IFFT modulation, that is, the original frequency domain The domain signal X is multiplied by the phase rotation factor P ^v to obtain V corresponding time domain signals x ^v through the IFFT modulation unit, where 1≤v≤V, and V represents the number of phase rotation factors;

S102: For the time-domain signal x ^v , use the cyclic shift property of the time-domain signal of FFT, that is, F[x(n), k]=x(nk) _N <=>F[X(n), k]=X (n) e ^-j2πkn/N , to obtain the parity signal in the time domain corresponding to the parity sequence in the frequency domain, where (·) _N represents a modulo N operation, and k=N/2 is selected so that the parity sequence in the frequency domain produces different phases Variety:

According to the above two formulas, the parity sequence of the frequency domain signal can be distinguished by the cyclic shift of the time domain signal. Therefore, the time domain parity can be obtained by the time domain signal ^xv and its cyclically shifted signal The signals are:

S103: Obtain an odd signal in the time domain and time-domain even signal Time-domain equivalent SFBC coding is performed after performing time-domain cyclic shifts of different lengths respectively, each time-domain cyclic shift and time-domain equivalent SFBC coding are performed to obtain a candidate sequence pair, and the time-domain equivalent The SFBC encoding method is as follows: the conjugate inversion of the time-domain signal using FFT corresponds to the basic property of the conjugate of the frequency-domain signal, that is, FFT[x((Nn)N) ^* ]=[X(n)] ^* , every Odd signal in sub-time domain and time-domain even signal The corresponding coded signal is obtained by carrying out the cyclic shift and then the conjugate inversion. The SFBC coding form of the frequency domain signal is expressed as:

Odd signal in time domain and time-domain even signal The cyclic shift factors of are: with M represents the number of cyclic shifts, and M different time-domain signal sequences are obtained through the time-domain cyclic shift operation:

In the above formula, 1≤m, n≤M; the time-domain parity signal undergoes M time-domain cyclic shifts of different lengths, and then undergoes time-domain equivalent SFBC coding, and a total of M ² different candidate sequence pairs are obtained, Since the cyclic shift of the time domain signal corresponds to the phase rotation of the frequency domain signal, the SFBC coding form of the frequency domain signal is uniformly expressed as:

The time-domain parity signals corresponding to the V time-domain signals are subjected to the time-domain equivalent SFBC encoding after the time-domain cyclic shift operation, and a total of ^two candidate sequence pairs of VM are obtained;

S104: Calculate the PAPR of each candidate sequence pair, select the one with the largest PAPR as the PAPR of the entire sequence pair, then select the minimum PAPR value among all sequence pairs as the PAPR of the system, and select a pair of sequence pairs with the smallest PAPR as the transmission sequence, the transmission sequence enters the cyclic prefix unit to add a cyclic prefix after passing through the parallel-to-serial conversion unit, and then transmits through the antenna after passing through the D/A conversion unit and the radio frequency unit;

The specific flow of the receiving process is as follows:

S201: The received signal of the antenna passes through the radio frequency unit, the A/D conversion unit, the decyclic prefix unit, and the serial-to-parallel conversion unit in turn, and then performs FFT demodulation to obtain the frequency domain signal R=[R _e , R _o ], and the form of the received signal is expressed :

In the formula, H _ie and H _io respectively represent the transmission channel gain, N _i represents the channel noise, where i=1, 2, for the channel gain of adjacent subcarrier signals, H _1e =H _1o =H ₁ , H _2e ＝H _2o ＝H ₂ , so SFBC decoding is performed on the frequency domain signal R=[R _e , R _o ] to obtain the signal [Y _e , Y _o ]:

In the formula, α ² ＝|H ₁ | ² +|H ₂ | ² , the [Y _e , Y _o ] obtained by the solution is the time-domain parity signal at the transmitter The signal after the cyclic shift of different lengths is equivalent to the signal obtained by phase rotation of the frequency domain signal;

S202: Perform blind signal detection on the frequency-domain parity signal obtained by SFBC decoding: perform reverse phase rotation on the solved signal [Y _e , Y _o ], recover the cyclic shift factor and phase rotation factor P ^v , and obtain the reverse Multiply the recovered phase rotation factor P ^v to the rotation sequence to obtain the detection signal, and perform parallel-to-serial conversion and baseband demodulation on the obtained detection signal to obtain the original signal.

The signal blind detection in step S202 includes recovering the cyclic shift factor and recovering the phase rotation factor. The method for recovering the cyclic shift factor is:

Perform reverse phase rotation on the solved signal [Y _e , Y _o ], due to the time-domain parity signal at the transmitter VM cyclic shift factors are used respectively. Therefore, the parity signals Y _e and Y _o respectively require VM complex multiplications to realize reverse phase rotation:

In the formula, 1≤v≤V, 1≤m≤M, 1≤vm≤VM, Through the above formula, the odd and even signals respectively obtain VM reverse rotation sequences. There must be a sequence in these reverse rotation sequences, and all its frequency points have been rotated to the constellation points of the modulation signal. Due to the existence of noise, the frequency points may deviate from The original constellation point, but the sum of the distances between all frequency points and the nearest constellation point is the smallest in probability, so first put the reverse rotation sequence with It is judged as the nearest constellation point Y ^Q (n), and then calculate the sum of the distances from all frequency points to the constellation point Y ^Q (n), the time-domain odd signal and the time-domain even signal respectively get VM distance values, from Select a cyclic shift factor corresponding to the smallest distance among all distances with time-domain parity signal with The cyclic shift factors of , denoted as u′ _e and u′ _o :

In the above formula, Y ^Q (n)∈Q, Q is the signal constellation diagram of the modulation mode selected by the transmitter, and the recovered cyclic shift factors u′ _e and u′ _o correspond to the reverse rotation signal with Respectively as the recovered odd signal and even signal;

The method for recovering the phase rotation factor P ^v is: since the original frequency domain signal X is multiplied by the phase rotation factor P ^v each time, the parity sequence of the time domain signal x ^v is IFFT modulated with The cyclic shift factor to use with different, therefore, the phase rotation factor P ^v used by the transmitting end is judged by the obtained vectors of the cyclic shift factors u′ _e and u′ _o .

The beneficial effect of the present invention is: the original signal undergoes different phase rotations and is subjected to IFFT modulation to obtain a time-domain signal, and then the time-domain signal sequence corresponding to the parity sequence in the frequency domain is obtained by using the properties of IFFT, and the parity sequence is cyclically shifted with different lengths Afterwards, multiple candidate sequence pairs with different PAPRs are obtained through time-domain equivalent SFBC encoding, and finally the sequence pair with the best PAPR performance is selected from all candidate sequences for transmission. Therefore, under the number of IFFT operations of the traditional SLM algorithm, multiple sequence sets with different PAPRs can be obtained only by operations such as cyclic shift and phase rotation of the time domain signal, which greatly reduces the computational complexity of the system; the receiving end passes Comparing the distance between the reverse rotation sequence and its nearest signal constellation point to restore the parity signal, cyclic shift factor and phase rotation factor, the blind detection of the received signal is realized. Compared with the traditional SLM method, the computational complexity is greatly reduced, and it does not Sideband side information needs to be transmitted.

Description of drawings

Fig. 1 is the transmitter block diagram of traditional SLM algorithm;

Fig. 2 is the receiver block diagram of traditional SLM algorithm;

Fig. 3 is the launch processing flow chart of improved SLM algorithm of the present invention;

Fig. 4 is a flow chart of time-domain cyclic shift and time-domain equivalent SFBC coding in the transmission process of the present invention;

Fig. 5 is the equivalent SFBC coding schematic diagram of the present invention;

Fig. 6 is the flow chart of the receiving process of the improved SLM algorithm of the present invention;

FIG. 7 is a flow chart of signal blind detection in the receiving process of the present invention.

detailed description

The technical solution of the present invention will be further described below in conjunction with the accompanying drawings, but the content protected by the present invention is not limited to the following description.

A SLM method for reducing the peak-to-average power ratio of a SFBC MIMO-OFDM system, including two parts, a transmission processing process and a receiving processing process, as shown in Figure 3, the specific flow of the transmission processing process is:

S101: The source bits of each antenna pass through the baseband modulation unit and the serial-to-parallel conversion unit to obtain the original frequency domain signal X, and the original frequency domain signal X enters the phase rotation sequence generator for different phase rotations and performs IFFT modulation, that is, the original frequency domain The domain signal X is multiplied by the phase rotation factor P ^v to obtain V corresponding time domain signals x ^v through the IFFT modulation unit, where 1≤v≤V, and V represents the number of phase rotation factors;

S102: For the time-domain signal x ^v , use the cyclic shift property of the time-domain signal of FFT, that is, F[x(n), k]=x(nk) _N <=>F[X(n), k]=X (n) e ^-j2πkn/N , to obtain the parity signal in the time domain corresponding to the parity sequence in the frequency domain, where (·) _N represents a modulo N operation, and k=N/2 is selected so that the parity sequence in the frequency domain produces different phases Variety:

According to the above two formulas, the parity sequence of the frequency domain signal can be distinguished by the cyclic shift of the time domain signal. Therefore, the time domain parity can be obtained by the time domain signal ^xv and its cyclically shifted signal The signals are:

S103: Obtain an odd signal in the time domain and time-domain even signal Using time-domain cyclic shifts of different lengths and then performing time-domain equivalent SFBC coding, each time-domain cyclic shift and time-domain equivalent SFBC coding is used to obtain a candidate sequence pair, as shown in Figure 4. The time-domain equivalent SFBC coding method described above is shown in Figure 5. The conjugate inversion of the time-domain signal using FFT corresponds to the basic property of the conjugate of the frequency-domain signal, that is, FFT[x((Nn)N) ^* ] ＝[X(n)] ^* , each time domain odd signal and time-domain even signal The corresponding coded signal is obtained by carrying out the cyclic shift and then the conjugate inversion. The SFBC coding form of the frequency domain signal is expressed as:

Odd signal in time domain and time-domain even signal The cyclic shift factors of are: with M represents the number of cyclic shifts, and M different time-domain signal sequences are obtained through the time-domain cyclic shift operation:

In the above formula, 1≤m, n≤M; the time-domain parity signal undergoes M time-domain cyclic shifts of different lengths, and then undergoes time-domain equivalent SFBC coding, and a total of M ² different candidate sequence pairs are obtained, Since the cyclic shift of the time domain signal corresponds to the phase rotation of the frequency domain signal, the SFBC coding form of the frequency domain signal is uniformly expressed as:

The time-domain parity signals corresponding to the V time-domain signals are subjected to the time-domain equivalent SFBC encoding after the time-domain cyclic shift operation, and a total of ^two candidate sequence pairs of VM are obtained;

S104: Calculate the PAPR of each candidate sequence pair, select the one with the largest PAPR as the PAPR of the entire sequence pair, then select the minimum PAPR value among all sequence pairs as the PAPR of the system, and select a pair of sequence pairs with the smallest PAPR as the transmission Sequence, denoted as [x′ ₁ , x′ ₂ ,], the transmission sequence enters the cyclic prefix unit to add a cyclic prefix after passing through the parallel-to-serial conversion unit, and then transmits through the antenna after passing through the D/A conversion unit and the radio frequency unit;

As shown in Figure 6, the specific flow of the receiving process is as follows:

S201: The received signal of the antenna passes through the radio frequency unit, the A/D conversion unit, the decyclic prefix unit, and the serial-to-parallel conversion unit in turn, and then performs FFT demodulation to obtain the frequency domain signal R=[R _e , R _o ], and the form of the received signal is expressed :

In the formula, H _ie and H _io respectively represent the transmission channel gain, N _i represents the channel noise, where i=1, 2, for the channel gain of adjacent subcarrier signals, H _1e =H _1o =H ₁ , H _2e ＝H _2o ＝H ₂ , so SFBC decoding is performed on the frequency domain signal R=[R _e , R _o ] to obtain the signal [Y _e , Y _o ]:

In the formula, α ² ＝|H ₁ | ² +|H ₂ | ² , the [Y _e , Y _o ] obtained by the solution is the time-domain parity signal at the transmitter The signal after the cyclic shift of different lengths is equivalent to the signal obtained by phase rotation of the frequency domain signal;

S202: Perform blind signal detection on the frequency-domain parity signal obtained by SFBC decoding: perform reverse phase rotation on the solved signal [Y _e , Y _o ], recover the cyclic shift factor and phase rotation factor P ^v , and obtain the reverse Multiply the recovered phase rotation factor P ^v to the rotation sequence to obtain the detection signal, and perform parallel-to-serial conversion and baseband demodulation on the obtained detection signal to obtain the original signal.

As shown in Figure 7, the signal blind detection in step S202 includes two parts: recovering the cyclic shift factor and recovering the phase rotation factor, and the method for recovering the cyclic shift factor is:

Perform reverse phase rotation on the solved signal [Y _e , Y _o ], due to the time-domain parity signal at the transmitter VM cyclic shift factors are used respectively. Therefore, the parity signals Y _e and Y _o respectively require VM complex multiplications to realize reverse phase rotation:

In the formula, 1≤v≤V, 1≤m≤M, 1≤vm≤VM, Through the above formula, the odd and even signals respectively obtain VM reverse rotation sequences. There must be a sequence in these reverse rotation sequences, and all its frequency points have been rotated to the constellation points of the modulation signal. Due to the existence of noise, the frequency points may deviate from The original constellation point, but the sum of the distances between all frequency points and the nearest constellation point is the smallest in probability, so first put the reverse rotation sequence with It is determined to be the nearest constellation point Y ^Q (n), and then calculate the sum of the distances from all frequency points to the constellation point Y ^Q (n), the time-domain odd signal and the time-domain even signal respectively get VM distance values, from Select a cyclic shift factor corresponding to the smallest distance among all distances with time-domain parity signal with The cyclic shift factors of , denoted as u′ _e and u′ _o :

In the above formula, Y ^Q (n)∈Q, Q is the signal constellation diagram of the modulation mode selected by the transmitter, and the recovered cyclic shift factors u′ _e and u′ _o correspond to the reverse rotation signal with Respectively as the recovered odd signal and even signal;

The method for recovering the phase rotation factor P ^v is: since the original frequency domain signal X is multiplied by the phase rotation factor P ^v each time, the parity sequence of the time domain signal x ^v is IFFT modulated with The cyclic shift factor to use with are different, therefore, the phase rotation factor P ^v used by the transmitter can be judged by the obtained vectors of the cyclic shift factors u′ _e and u′ _o , for example, if , then it is judged that the phase rotation factor used by X is P ¹ .

When the number of subcarriers of OFDM is N and the oversampling rate is L, the number of complex multiplications and complex additions required for one IFFT operation are LN/2log ₂ LN and LNlog ₂ LN respectively. When the number of candidate sequence pairs generated is VM ² , it can be seen from Figure 1 that the traditional SLM algorithm needs 2VM ² IFFT operations of LN points to modulate the frequency domain sequence after SFBC encoding to obtain VM ² time domain candidates For sequence pairs, the required number of complex multiplications and complex additions is VM ² LN/2log ₂ LN and 2VM ² LNlog ₂ LN, respectively. The algorithm proposed by the present invention also produces ² candidate sequence pairs of VM. The original frequency domain signal only needs V times of LN point IFFT operations to obtain the time domain signal, and the required number of complex multiplications and complex additions are respectively VLN/2log ₂ LN And VLNlog ₂ LN, when performing operations such as cyclic shifting, conjugate inversion, and time-domain phase rotation on the parity signal in the time domain to obtain multiple candidate sequence pairs, the number of complex multiplications and complex additions required are 2VMLN and 2VM ² LN, therefore, the total times of complex multiplication and complex addition of the algorithm of the present invention are VLN(1/2 log ₂ LN+2M) and VLN(log ₂ LN+2M ² ) respectively.

In order to measure the reduction performance of computational complexity, the computational complexity reduction ratio (CCRR) is generally used, which is defined as:

In the formula, CSB-SLM represents the algorithm of the present invention, and C-SLM represents the traditional algorithm. The following table shows that when the number of subcarriers N=256, the oversampling rate L=4, the number of candidate sequence pairs generated by the C-SLM algorithm and the CSB-SLM algorithm is K=VM ² , the required Complex multiplication and complex addition times and CCRR function values.

Table 1 Computational complexity of the transmitter

It can be seen from Table 1 that when the same number of candidate sequences is generated, the CSB-SLM algorithm can greatly reduce the computational complexity of the C-SLM algorithm. When the number of candidate sequences is 12, the complex multiplication and complex multiplication required by CSB-SLM Compared with the C-SLM algorithm, the number of additions reaches 77.5% and 77.5% respectively. It is worth noting that with the increase of the number of candidate sequences, the ability of the algorithm of the present invention to reduce complexity will further increase. Therefore, the CSB-SLM algorithm proposed in this paper has a greater advantage in reducing complexity.

When the number of time-domain candidate sequences generated by the transmitting end is VM ² , the traditional SLM algorithm transmitting end needs to transmit the sideband side information of the phase rotation factor P used, which requires a total of log ₂ VM ² bits. The algorithm proposed by the present invention , if the shift factor U and the phase rotation factor P are also transmitted as sideband side information, log ₂ VM ² bits are also required. For the traditional SLM algorithm, the received signal needs an FFT operation of the LN point to obtain the frequency domain signal, and then according to the received sideband information, the frequency domain signal is directly multiplied by the corresponding phase rotation factor to restore the original signal, therefore, the required complex number The number of multiplications and the number of complex additions are LN/2log ₂ LN+2N and LN log ₂ LN, respectively. In the SLM algorithm proposed by the present invention, the received signal also requires an FFT operation of the LN point to obtain the frequency domain signal and decode it to obtain the frequency domain parity sequence. If U and P are known, the parity sequence is directly based on with The reverse rotation signal is obtained, and then multiplied by the phase rotation factor to obtain the original signal. At this time, the complexity of the receiving end is equivalent to that of the traditional SLM algorithm; if U and P are unknown, 2MV times of N-point complex multiplication are required to realize the reverse rotation sequence and The distance judgment of the corresponding modulated signal constellation point restores the cycle-out shift factor. Therefore, the total number of times of complex multiplication and the number of times of complex number addition required by the SLM algorithm receiving end proposed by the present invention are respectively LN/2log ₂ LN+2MVN(L+ 1) and LNlog ₂ LN+2MVN.

Claims (2)

1. a kind of SLM methods of reduction SFBC MIMO-OFDM system peak-to-average power ratios, including transmitting processing procedure and receiving area Reason process two parts, it is characterised in that：Described transmitting processing procedure idiographic flow is：

S101：The source bits of every antenna obtain original frequency domain signal X after baseband modulation unit and serioparallel exchange unit, Original frequency domain signal X carries out carrying out IFFT modulation after different phase places into phase place sequencer, i.e., original frequency Domain signal X and phase rotation coefficient P^vV corresponding time-domain signal x is obtained by IFFT modulating units after multiplication^v, wherein, 1≤v ≤ V, V represent phase rotation coefficient number；

S102：For time-domain signal x^v, using the time-domain signal cyclic shift property of FFT, i.e. F [x (n), k]=x (n-k)_N<=> F [X (n), k]=X (n) e^-j2πkn/N, the corresponding time domain parity signal of frequency domain sequence of parity is obtained, wherein, ()_NRepresent mould N Operation, selects k=N/2 so that the sequence of parity of frequency domain produces different phase place changes：

Learnt by both the above formula, the sequence of parity of frequency-region signal can be distinguished using the cyclic shift of time-domain signal Come, therefore, by time-domain signal x^vCan be respectively in the hope of time domain parity signal with the signal after its cyclic shift：

$x_e^v=(x^v+F\&lsqb;x^v,N/2\&rsqb;)/2$

$x_o^v=(x^v-F\&lsqb;x^v,N/2\&rsqb;)/2;$

S103：To obtaining the strange signal of time domainWith time domain idol signalUsing after the time-domain cyclic shift for carrying out different length respectively Carry out time-domain equivalent SFBC coding again, each time-domain cyclic shift simultaneously carries out time-domain equivalent SFBC coding and obtains an alternative sequence Right, described time-domain equivalent SFBC coded method is：Using conjugation reversion being total to corresponding to frequency-region signal of the time-domain signal of FFT The fundamental property of yoke, i.e. FFT [x ((N-n)_N)^*X]=[(n)]^*, the strange signal of each time domainWith time domain idol signalCarry out respectively Carry out conjugation reversion after cyclic shift again and obtain corresponding encoded signal, the SFBC coding forms of frequency-region signal are expressed as：

$\left[\begin{array}{cc}{X}_e^v\left(n\right)& -{\&lsqb;X_o^v\left(n\right)\&rsqb;}^{*}\\ X_o^v\left(n\right)& {\&lsqb;X_e^v\left(n\right)\&rsqb;}^{*}\end{array}\right]$

If the strange signal of time domainWith time domain idol signalThe cyclic shift factor be respectively：WithM represents the number of times of cyclic shift, by time-domain cyclic shift operation obtain M kinds it is different when Domain signal sequence：

$x_e^{vm}=circshift\{x_e^v,\&lsqb;0,u_e^{vm}\&rsqb;\}$

$x_o^{vn}=circshift\{x_o^v,\&lsqb;0,u_o^{vn}\&rsqb;\}$

In above formula, 1≤m≤M, 1≤n≤M；Time domain parity signal respectively through after the time-domain cyclic shift of M different length again Time-domain equivalent SFBC coding is carried out, M is always obtained²Different alternative sequences pair are planted, due to the cyclic shift correspondence of time-domain signal In the phase place of frequency-region signal, then its frequency-region signal SFBC coding forms are collectively expressed as：

$\left[\begin{array}{cc}{e}^{j2{\mathrm{\&pi;u}}_e^{vm}(n-1)/N}{X}_e^{vm}\left(n\right)& -{\&lsqb;e^{j2{\mathrm{\&pi;u}}_o^{vn}(n-1)/N}{X}_o^{vn}\left(n\right)\&rsqb;}^{*}\\ e^{j2{\mathrm{\&pi;u}}_0^{vn}(n-1)/N}{X}_o^{vn}\left(n\right)& {\&lsqb;e^{j2{\mathrm{\&pi;u}}_e^{vm}(n-1)/N}{X}_o^{vm}\left(n\right)\&rsqb;}^{*}\end{array}\right]$

The corresponding time domain parity signal of V time-domain signal after time-domain cyclic shift operation again respectively through carrying out time-domain equivalent SFBC Coding, is always obtained VM²Plant alternative sequence pair；

S104：The PAPR of each alternative sequence pair is calculated, a PAPR as whole sequence pair for selecting PAPR maximum, then PAPR minima is selected in all of sequence pair as the PAPR of system, and selects PAPR minimum a pair of sequences to as transmission Sequence, the transmission sequence enters cyclic prefix unit after parallel serial conversion unit and adds Cyclic Prefix, then single through D/A conversions Launched by antenna after unit and radio frequency unit；

Described reception processing process idiographic flow is：

S201：The reception signal of antenna passes sequentially through radio frequency unit, A/D converting units, removes cyclic prefix unit and serioparallel exchange FFT demodulation is carried out after unit and obtains frequency-region signal R=[R_e, R_o], the form for receiving signal is represented：

$R_e=H_{1e}{e}^{j2{\mathrm{\&pi;u}}_e^{vm}/N}{X}_e^{vm}+H_{2e}{e}^{j2{\mathrm{\&pi;u}}_o^{vn}/N}{X}_o^{vn}+N_1$

$R_o=H_{1o}{\&lsqb;-e^{j2{\mathrm{\&pi;u}}_o^{vn}/N}{X}_o^{vn}\&rsqb;}^{*}+H_{2o}{\&lsqb;e^{j2{\mathrm{\&pi;u}}_e^{vm}/N}{X}_e^{vm}\&rsqb;}^{*}+N_2$

In formula, H_ieAnd H_ioTransmission channel gain, N are represented respectively_iInterchannel noise is represented, wherein, i=1,2, for adjacent son is carried The channel gain of ripple signal, there is H_1e=H_1o=H₁, H_2e=H_2o=H₂, therefore to frequency-region signal R=[R_e, R_o] carry out SFBC solutions Code, obtains signal [Y_e, Y_o]：

$\left\{\begin{array}{c}{Y}_e={\mathrm{\&alpha;}}^{-2}(R_e{\left(H_1\right)}^{*}+{\left(R_o\right)}^{*}{H}_2)\\ Y_o={\mathrm{\&alpha;}}^{-2}(R_e{\left(H_2\right)}^{*}-{\left(R_o\right)}^{*}{H}_1)\end{array}\right.$

In formula, α²=| H₁|²+|H₂|², [the Y for solving_e, Y_o] it is by transmitting terminal time domain parity signalHave passed through different length Cyclic shift after signal, be the equal of that the signal that phase place is obtained has been carried out by frequency-region signal；

S202：The frequency domain parity signal that SFBC decodings are obtained is carried out into signal blind Detecting：To the signal [Y for solving_e, Y_o] carry out instead To phase place, recover the cyclic shift factor and phase rotation coefficient P^v, it is multiplied by what is recovered by the reverse rotation sequence for obtaining Phase rotation coefficient P^vDetection signal is obtained, the detection signal to obtaining carries out parallel-serial conversion and base band demodulating recovers to obtain original Signal.

2. SLM methods of reduction SFBC MIMO-OFDM system peak-to-average power ratios according to claim 1, its feature exists In：The signal blind Detecting of described step S202 includes recovering the cyclic shift factor and recovery phase rotation coefficient two parts, institute The method of the recovery cyclic shift factor stated is：

To the signal [Y for solving_e, Y_o] reverse phase rotation is carried out, due to the time domain parity signal of transmitting terminalEach use VM cyclic shift factor, therefore, parity signal Y_eAnd Y_oIt is respectively necessary for VM complex multiplication and realizes that reverse phase rotates：

$Y_e^{vm}\left(n\right)=y_e\left(n\right)e^{j2{\mathrm{\&pi;u}}_e^{vm}(n-1)/N}$

$Y_o^{vm}\left(n\right)=Y_o\left(n\right)e^{j2{\mathrm{\&pi;u}}_o^{vm}(n-1)/N}$

In formula, 1≤v≤V, 1≤m≤M, 1≤vm≤VM,It is logical Above formula is crossed, parity signal respectively obtains VM reverse rotation sequence, and these reversely rotate in sequence and there will necessarily be a sequence, it All frequencies had rotated in the constellation point of modulated signal, because noise is present, frequency may deviate from original constellation Point, but all frequency its nearest neighbours constellation points from probability is minimum apart from sum, therefore, first reverse rotation sequenceWithIt is judged to from its nearest constellation point Y^Q(n), then all frequencies are calculated to constellation point Y^QThe distance of (n) it VM distance value has been respectively obtained with, the strange signal of time domain and time domain idol signal, a most narrow spacing has been selected respectively from all distances From the corresponding cyclic shift factorWithTransmitting terminal time domain parity signal the mostWithThe cyclic shift factor, be designated as u′_eWith u '_o：

$u_e^{\&prime;}=\arg \underset{1\&le;vm\&le;VM}{min}\underset{n=1}{\overset{N}{\&Sigma;}}|Y_e^{vm}\left(n\right)-Y^Q\left(n\right){|}^2$

$u_o^{\&prime;}=\arg \underset{1\&le;vm\&le;VM}{min}\underset{n=1}{\overset{N}{\&Sigma;}}|Y_o^{vm}\left(n\right)-Y^Q\left(n\right){|}^2$

In above formula, Y^Q(n) ∈ Q, Q for transmitting terminal selected modulation mode signal constellation (in digital modulation) figure, the cyclic shift factor u ' for recovering_e With u '_oCorresponding reverse rotation signalWithRespectively as the strange signal for recovering to obtain and even signal；

Described recovery phase rotation coefficient P^vMethod be：Because original frequency domain signal X is multiplied by every time phase rotation coefficient P^vAfterwards Carry out IFFT modulation time-domain signal x^vSequence of parityWithThe cyclic shift factor for usingWithDifference, therefore, pass through The cyclic shift factor u ' for obtaining_eWith u '_oThe vector at place is judging to obtain the phase rotation coefficient P that transmitting terminal is used^v。