CN103280837A

CN103280837A - Photovoltaic grid connected direct current injecting restraining method

Info

Publication number: CN103280837A
Application number: CN2013102062722A
Authority: CN
Inventors: 杨艺云; 李长佶; 司传涛; 张阁; 肖园园; 周柯; 周林; 郭珂; 郑光辉
Original assignee: Chongqing University; Electric Power Research Institute of Guangxi Power Grid Co Ltd
Current assignee: Chongqing University; Electric Power Research Institute of Guangxi Power Grid Co Ltd
Priority date: 2013-05-30
Filing date: 2013-05-30
Publication date: 2013-09-04

Abstract

The invention discloses a photovoltaic grid-connected DC injection suppression method. The method uses a PR+PI or quasi-PR+PI dual controller as a control strategy of a grid-connected current loop controller, and operates an inverter bridge switching device through feedback control. Inconsistency causes DC injection to be suppressed. On the one hand, the present invention uses a PR controller or a quasi-PR controller as a grid-connected current loop controller to realize steady-state non-difference control of an AC controlled signal of a specified frequency in a static coordinate system without adding a detection circuit. Realize unity power factor grid connection, on the other hand, use PI controller to effectively restrain the DC flow.

Description

A photovoltaic grid-connected DC injection suppression method

技术领域technical field

本发明涉及光伏并网系统中直流注入抑制领域，更具体的说，特别涉及一种利用控制方式抑制逆变桥中开关器件动作不一致带来的光伏并网直流注入问题的光伏并网直流注入抑制方法。The present invention relates to the field of DC injection suppression in photovoltaic grid-connected systems, and more specifically, to a photovoltaic grid-connected DC injection suppression that uses a control method to suppress the photovoltaic grid-connected DC injection problem caused by the inconsistency of the switching devices in the inverter bridge. method.

背景技术Background technique

并网发电是光伏领域研究和发展的热点。根据有无隔离变压器，光伏并网系统可分为隔离型和非隔离型。非隔离型光伏并网系统不需要工频变压器，在经济效益和技术上具有一定优势，提高了并网系统的整机效率，但是由于没有变压器隔离，光伏阵列和电网之间存在电气连接，带来直流注入问题。直流注入的不利影响主要表现为扰乱变压器正常运行，而电网中存在大量的配电变压器，因此直流注入现象作为光伏并网应用中实际存在的问题日益受到重视。现有直流注入抑制方法主要有检测补偿法和电容隔直法，其中检测补偿法方法实现较繁琐，且额外的直流分量检测电路增加了系统的复杂程度和成本；电容隔直法包括直接串联隔直电容和虚拟电容法，直接串联隔直电容能有效抑制直流分量，但是电容值大，并且需要设置保护电路，增加系统的控制策略和总体结构复杂度，虚拟电容法能达到电容隔直功能，但使得控制结构发生改变，系统的动态响应速度会随电容的增大而减慢，而电容越小其阻抗越大，基波压降越大，电容取值要折中考虑，使系统不能工作于最佳状态。Grid-connected power generation is a hot spot in the research and development of photovoltaic field. According to whether there is an isolation transformer, the photovoltaic grid-connected system can be divided into isolated type and non-isolated type. The non-isolated photovoltaic grid-connected system does not require a power frequency transformer, which has certain advantages in terms of economic benefits and technology, and improves the overall efficiency of the grid-connected system. However, since there is no transformer isolation, there is an electrical connection between the photovoltaic array and the grid, which brings Come DC injection problem. The adverse effect of DC injection is mainly manifested as disturbing the normal operation of transformers, and there are a large number of distribution transformers in the power grid, so the phenomenon of DC injection as a practical problem in photovoltaic grid-connected applications has been paid more and more attention. The existing DC injection suppression methods mainly include detection compensation method and capacitor DC blocking method. The detection compensation method is more complicated to implement, and the additional DC component detection circuit increases the complexity and cost of the system; Direct capacitor and virtual capacitor method, direct series DC blocking capacitor can effectively suppress the DC component, but the capacitance value is large, and a protection circuit needs to be installed, which increases the control strategy and overall structural complexity of the system. The virtual capacitor method can achieve the DC blocking function of the capacitor. However, if the control structure changes, the dynamic response speed of the system will slow down with the increase of the capacitance, and the smaller the capacitance, the greater the impedance and the greater the fundamental voltage drop. The value of the capacitance must be compromised, so that the system cannot work at its best.

发明内容Contents of the invention

由于直流注入产生的原因可归为两类：逆变桥开关器件动作不一致和检测元件的偏移误差，而前者属于前向通道上的扰动，因此可以通过反馈控制进行适当抑制，本发明的目的在于利用控制方式抑制逆变桥中开关器件动作不一致带来的直流注入问题，提供一种光伏并网直流注入抑制方法，该方法能在不增加检测电路前提下，有效抑制直流注入的现象。The causes of direct current injection can be classified into two categories: inconsistent actions of inverter bridge switching devices and offset errors of detection elements, and the former belongs to the disturbance on the forward channel, so it can be properly suppressed by feedback control, the purpose of the present invention The purpose is to use the control method to suppress the DC injection problem caused by the inconsistency of the switching devices in the inverter bridge, and provide a photovoltaic grid-connected DC injection suppression method, which can effectively suppress the DC injection phenomenon without increasing the detection circuit.

为了提供高质量的并网电流，并网滤波器选用LCL滤波器，它具有较好的高频谐波抑制功能，并针对LCL滤波器采用并网电流外环和电容电流内环双闭环控制策略，实现对谐振现象的有源阻尼，则电流外环控制器输出v_m至并网电流i_o的传递函数为：In order to provide high-quality grid-connected current, the grid-connected filter uses an LCL filter, which has a good high-frequency harmonic suppression function, and adopts a double-closed-loop control strategy for the grid-connected current outer loop and the capacitor current inner loop for the LCL filter , to realize the active damping of the resonance phenomenon, then the transfer function of the current outer loop controller output v _m to the grid current i _o is:

${G G}_{i i} ((s the s)) = = \frac{{i i}_{o o} ((s the s))}{{v v}_{m m} ((s the s))} = = \frac{{k k}_{PWM PWM}}{{L L}_{11} {L L}_{22} {C C}_{f f} {s the s}^{33} + + {k k}_{PWM PWM} {k k}_{f f} {C C}_{f f} {L L}_{22} {s the s}^{22} + + (({L L}_{11} + + {L L}_{22})) s the s}$

光伏并网系统中存在直流注入问题时，并网电流可表示为：When there is a DC injection problem in the photovoltaic grid-connected system, the grid-connected current can be expressed as:

i_o＝A₂sinω₂t+D_o i _o ＝A ₂ sinω ₂ t+D _o

其中：A₂代表并网电流幅值，ω₂代表并网电流角频率，D_o代表并网电流中的直流分量。而在理想情况下，系统电流的参考信号i_o ^*中不含有直流量，即：Among them: A ₂ represents the amplitude of the grid-connected current, ω ₂ represents the angular frequency of the grid-connected current, and D _o represents the DC component in the grid-connected current. Under ideal conditions, the reference signal i _o ^* of the system current does not contain DC, that is:

i_o ^*＝A₁sinω₁ti _o ^* ＝A ₁ sinω ₁ t

其中：A₁代表参考电流幅值，ω₁代表电网基频，i_o ^*也可看作由两部分组成，即交流部分参考信号A₁sinω₁t，直流部分参考信号为零。以PR控制器作为并网电流环控制器能够在静止坐标系下对指定频率的交流被控信号实现稳态无差控制，实现单位功率因数并网，对直流量却无法实现零稳态误差控制，即不能起到抑制直流分量的作用，而PI控制器则能弥补其在直流控制方面的不足。Among them: A ₁ represents the reference current amplitude, ω ₁ represents the fundamental frequency of the power grid, and i _o ^* can also be regarded as composed of two parts, that is, the AC part reference signal A ₁ sinω ₁ t, and the DC part reference signal is zero. Using the PR controller as the grid-connected current loop controller can realize steady-state error-free control of the AC controlled signal with a specified frequency in the static coordinate system, and realize grid-connection with unit power factor, but it cannot achieve zero steady-state error control for DC flow , that is, it cannot play the role of suppressing the DC component, while the PI controller can make up for its deficiency in DC control.

为了解决以上提出的问题，本发明采用的技术方案为：In order to solve the problems raised above, the technical solution adopted in the present invention is:

本发明所采用的光伏并网直流注入抑制方法，是以PR+PI或准PR+PI双控制器作为并网电流环控制器控制策略，通过反馈控制对逆变桥开关器件动作不一致引起直流注入进行抑制。The photovoltaic grid-connected DC injection suppression method adopted in the present invention uses the PR+PI or quasi-PR+PI dual controller as the control strategy of the grid-connected current loop controller, and the DC injection is caused by the inconsistent action of the inverter bridge switching device through feedback control to suppress.

其中，一方面以PR+PI双控制器作为并网电流环控制器，与电网电压锁相环相配合，能达到单位功率因数并网和直流注入抑制的双重功效，PR+PI双控制器传递函数为：Among them, on the one hand, the PR+PI dual controller is used as the grid-connected current loop controller, which cooperates with the grid voltage phase-locked loop to achieve the dual effects of unit power factor grid-connected and DC injection suppression. The PR+PI dual controller delivers The function is:

${G G}_{ci ci} ((s the s)) = = {G G}_{PR PR + + PI P.I.} ((s the s)) = = {k k}_{p p 11} + + \frac{{k k}_{r r} s the s}{{s the s}^{22} + + {ω ω}_{00}^{22}} + + {k k}_{p p 22} + + \frac{{k k}_{i i}}{s the s}$

式中，k_p1为PR控制器比例系数，k_p2为PI控制器比例系数，ω₀为PR控制器谐振频率，k_r为PR控制器广义积分系数，s为复频率，k_i为PI控制器积分系数。In the formula, k _p1 is the proportional coefficient of the PR controller, k _p2 is the proportional coefficient of the PI controller, ω ₀ is the resonant frequency of the PR controller, k _r is the generalized integral coefficient of the PR controller, s is the complex frequency, and _ki is the PI control Integral coefficient of the device.

而在基波频率ω₁处该控制器增益为：And the controller gain at the fundamental frequency ω ₁ is:

$| | {G G}_{ci ci} ((j j {ω ω}_{11})) | | = = \sqrt{{(({k k}_{p p 11} + + {k k}_{p p 22}))}^{22} + + {((\frac{{k k}_{r r} {ω ω}_{11}}{- - {ω ω}_{11}^{22} + + {ω ω}_{00}^{22}} - - \frac{{k k}_{i i}}{{ω ω}_{11}}))}^{22}}$

这样，因交流参考信号频率为电网基频，故控制器中谐振频率设置为电网基频，即ω₀＝ω₁，由此可知基频处控制器增益趋于无穷大，故PR+PI双控制器能达到与PR控制器单独作用时相同的正弦信号无静差控制效果，实现并网电流与电网电压同频同相控制。在频率为0Hz时，该双控制器等效为PI控制单独作用，其增益也趋于无穷，能实现并网电流中直流部分的稳态无差跟踪，使得并网电流中的直流量趋近零，达到抑制直流注入的目的。In this way, since the frequency of the AC reference signal is the fundamental frequency of the power grid, the resonant frequency in the controller is set to the fundamental frequency of the power grid, that is, ω ₀ = ω ₁ , it can be seen that the gain of the controller at the fundamental frequency tends to infinity, so the PR+PI dual control The controller can achieve the same sinusoidal signal and no static difference control effect as when the PR controller acts alone, and realize the same frequency and phase control of the grid-connected current and the grid voltage. When the frequency is 0Hz, the dual controllers are equivalent to PI control acting alone, and its gain tends to be infinite, which can realize the steady-state error-free tracking of the DC part of the grid-connected current, making the DC flow in the grid-connected current approach Zero, to achieve the purpose of suppressing DC injection.

另一方面，降低电网频率偏移对并网电流的影响，采用准PR+PI双控制器作为并网电流环控制器，与电网电压锁相环相配合，能达到单位功率因数并网和直流注入抑制的双重功效，准PR+PI双控制器传递函数为：On the other hand, to reduce the impact of grid frequency deviation on grid-connected current, the quasi-PR+PI dual controller is used as the grid-connected current loop controller, which cooperates with the grid voltage phase-locked loop to achieve unity power factor grid-connected and DC The dual effect of injection suppression, the transfer function of the quasi-PR+PI dual controller is:

${G G}_{q q - - PR PR + + PI P.I.} ((s the s)) = = {k k}_{p p 11} + + \frac{22 {k k}_{r r} {ω ω}_{C C} s the s}{{s the s}^{22} + + 22 {ω ω}_{c c} s the s + + {ω ω}_{00}^{22}} + + {k k}_{p p 22} + + \frac{{k k}_{i i}}{s the s}$

式中，k_p1为准PR控制器比例系数，k_p2为PI控制器比例系数，ω₀为准PR控制器谐振频率，k_r为准PR控制器广义积分系数，s为复频率，k_i为PI控制器积分系数，ω_c为准PR控制器截止频率。In the formula, k _p1 is the proportional coefficient of the quasi-PR controller, k _p2 is the proportional coefficient of the PI controller, ω ₀ is the resonant frequency of the quasi-PR controller, k _r is the generalized integral coefficient of the quasi-PR controller, s is the complex frequency, k _i is the integral coefficient of the PI controller, and _ωc is the cut-off frequency of the quasi-PR controller.

由于准PR控制器在谐振频率处的增益不为无穷大，但其仍能达到高增益，能以较高的精度实现对交流信号的控制，因此利用准PR+PI控制器能更好的达到上述双重功效。Since the gain of the quasi-PR controller at the resonant frequency is not infinite, it can still achieve high gain and control the AC signal with high precision, so the quasi-PR+PI controller can better achieve the above Double effect.

而上述的PR+PI双控制器或准PR+PI双控制器设计步骤一般包括：The design steps of the above-mentioned PR+PI dual controller or quasi-PR+PI dual controller generally include:

(1)、根据带宽要求选择ω_c。考虑电网频率偏差最恶劣情况，选择准PR控制器的带宽ω_c/π＝1Hz，则ω_c＝πrad/s。(1) Select ω _c according to bandwidth requirements. Considering the worst case of power grid frequency deviation, select the bandwidth ω _c /π=1Hz of the quasi-PR controller, then ω _c =πrad/s.

(2)、选择比例系数k_p1和k_p2。由于k_p1和k_p2对系统性能的影响相同，可将两个比例系数合并，通过两者综合作用的效果，选取比例系数的总取值k_pe＝k_p1+k_p2。且如果选择电流环开环截止频率与谐振频率相隔较远，则认为截止频率处双控制器增益为k_pe。(2) Select the proportional coefficients k _p1 and k _p2 . Since k _p1 and k _p2 have the same impact on system performance, the two proportional coefficients can be combined, and the total value of the proportional coefficient k _pe =k _p1 +k _p2 can be selected through the combined effect of the two. And if the cut-off frequency of the open-loop current loop is selected to be far away from the resonant frequency, it is considered that the gain of the dual controllers at the cut-off frequency is k _pe .

(3)、根据PI控制的影响范围选择k_i。在谐振频率ω₀远小于电流环截止频率ω_ci的情况下，可选择ω_i＝(0.1～0.4)ω₀。(3) Select _ki according to the influence range of PI control. When the resonant frequency ω ₀ is much lower than the cut-off frequency ω _ci of the current loop, ω _i =(0.1˜0.4)ω ₀ can be selected.

(4)、根据对控制器增益的需求选择k_r。在谐振频率附近，由于主要是双控制器的谐振部分起作用，可忽略比例控制和积分控制，k_r取值可根据经验结合对控制器增益的需求进行选择，并通过校验简化系统补偿后的相位裕量对所选取值进行调节，以达到所需要求。(4) Select k _r according to the demand for controller gain. Near the resonant frequency, since the resonant part of the dual controller is mainly active, the proportional control and integral control can be ignored. The value of _kr can be selected according to experience and the demand for controller gain, and the system compensation can be simplified by checking Adjust the chosen value for the phase margin to meet the desired requirement.

(5)、校验相位裕量。对上述选择参数后的PR+PI双控制器或准PR+PI双控制器在补偿后系统的相位裕量γ_V进行校验，看能否满足30°≤γ_V≤70°，如不满足要求，则重新选择k_r和k_i，直至满足系统设计要求。(5) Check the phase margin. Check the phase margin γ _V of the PR+PI dual controller or quasi-PR+PI dual controller after compensation after the above parameters are selected to see if it can satisfy 30° _≤γV ≤70°, if not Requirements, then re-select k _r and _ki , until the system design requirements are met.

与现有技术相比，本发明的有益效果在于：在不增加检测电路的情况下，一方面以PR控制器或准PR控制器作为并网电流环控制器能够在静止坐标系下对指定频率的交流被控信号实现稳态无差控制，实现单位功率因数并网，另一方面以PI控制器对直流量进行有效的抑制。Compared with the prior art, the beneficial effect of the present invention is that, on the one hand, the PR controller or quasi-PR controller can be used as the grid-connected current loop controller to control the specified frequency in the static coordinate system without increasing the detection circuit. The AC controlled signal realizes steady-state non-difference control and realizes unit power factor grid connection. On the other hand, the PI controller is used to effectively restrain the DC flow.

附图说明Description of drawings

图1为单相两级式光伏并网系统；Figure 1 is a single-phase two-stage photovoltaic grid-connected system;

图2为并网电流和电容电流双闭环控制框图；Figure 2 is a block diagram of double closed-loop control of grid-connected current and capacitive current;

图3为PI、PR+PI和准PR+PI三种控制器的波特图；Figure 3 is the Bode diagram of the three controllers of PI, PR+PI and quasi-PR+PI;

图4为ω_c变化时准PR+PI双控制器波特图；Figure 4 is the bode diagram of the quasi-PR+PI dual-controller when _ωc changes;

图5(a)(b)分别为k_p1、k_p2变化时准PR+PI双控制器波特图；Figure 5(a)(b) are bode diagrams of quasi-PR+PI dual controllers when k _p1 and k _p2 change respectively;

图6为k_i变化时准PR+PI双控制器波特图；Figure 6 is a bode diagram of the quasi-PR+PI dual-controller when k _i changes;

图7为k_r变化时准PR+PI双控制器波特图；Figure 7 is the bode diagram of the quasi-PR+PI dual-controller when k _r changes;

图8(a)(b)分别为直流注入抑制前并网电流和电网电压仿真波形及并网电流频谱分析图；Figure 8(a)(b) is the grid-connected current and grid voltage simulation waveforms and grid-connected current spectrum analysis diagrams before DC injection suppression;

图9(a)(b)分别为采用本发明光伏并网直流注入抑制方法后并网电流和电网电压仿真波形及并网电流频谱分析图；Fig. 9 (a) (b) is the grid-connected current and grid voltage simulation waveforms and grid-connected current spectrum analysis diagrams respectively after adopting the photovoltaic grid-connected DC injection suppression method of the present invention;

图10(a)(b)分别为直流注入抑制前并网侧电压和并网电流实验波形及并网电流FFT分析和直流含量分析结果图；Figure 10(a)(b) are the experimental waveforms of grid-connected voltage and grid-connected current before DC injection suppression, and the results of FFT analysis of grid-connected current and DC content analysis;

图11(a)(b)分别为采用本发明光伏并网直流注入抑制方法后并网侧电压和并网电流实验波形及并网电流FFT分析和直流含量分析结果图。Fig. 11(a)(b) are the experimental waveforms of grid-connected side voltage and grid-connected current and the results of grid-connected current FFT analysis and DC content analysis after adopting the photovoltaic grid-connected DC injection suppression method of the present invention.

具体实施方式Detailed ways

下面结合实施例和附图对本发明作进一步详细的描述，但本发明的实施方式不限于此。The present invention will be further described in detail below with reference to the examples and drawings, but the implementation of the present invention is not limited thereto.

本发明所提供的一种光伏并网直流注入抑制的方法，其中：单相两级式光伏并网系统结构如图1所示，V_DC为直流母线电压，由光伏阵列经DC/DC变换器作用后提供，该逆变器为单相全桥逆变器，并网滤波器采用LCL滤波器。后级控制结构框图如图2所示，为了抑制LCL滤波器的谐振峰，保证系统稳定，采用并网电流和电容电流双闭环控制结构。其中，G_ci(s)为并网电流控制器传递函数，k_f为电容电流反馈系数，k_PWM为PWM逆变桥等效而成的增益环节。由图2所示的结构图可得参考电流i_o ^*和并网电流i_o之间的传递函数为：A photovoltaic grid-connected DC injection suppression method provided by the present invention, wherein: the single-phase two-stage photovoltaic grid-connected system structure is shown in Figure 1, V _DC is the DC bus voltage, and the photovoltaic array passes through the DC/DC converter Provided after functioning, the inverter is a single-phase full-bridge inverter, and the grid-connected filter adopts an LCL filter. The block diagram of the post-stage control structure is shown in Figure 2. In order to suppress the resonance peak of the LCL filter and ensure the stability of the system, a double closed-loop control structure of grid-connected current and capacitor current is adopted. Among them, G _ci (s) is the transfer function of the grid-connected current controller, k _f is the capacitor current feedback coefficient, and k _PWM is the gain link equivalent to the PWM inverter bridge. From the structure diagram shown in Figure 2, the transfer function between the reference current i _o ^* and the grid-connected current i _o can be obtained as:

$Φ Φ ((s the s)) = = \frac{{i i}_{o o} ((s the s))}{{i i}_{o o}^{* *} ((s the s))} = = \frac{{G G}_{ci ci} ((s the s)) {k k}_{PWM PWM}}{{L L}_{11} {L L}_{22} {C C}_{f f} {s the s}^{33} + + {k k}_{PWM PWM} {k k}_{f f} {C C}_{f f} {L L}_{22} {s the s}^{22} + + (({L L}_{11} + + {L L}_{22})) s the s + + {k k}_{PWM PWM} {G G}_{ci ci} ((s the s))}$

$((11))$

$= = \frac{11}{11 + + (({L L}_{11} {L L}_{22} {C C}_{f f} {s the s}^{33} + + {k k}_{PWM PWM} {k k}_{f f} {C C}_{f f} {L L}_{22} {s the s}^{22} + + {L L}_{11} s the s + + {L L}_{22} s the s)) / / (({k k}_{PWM PWM} {G G}_{ci ci} ((s the s))))}$

由于光伏并网系统中存在直流注入问题时，其并网电流可表示为：Due to the DC injection problem in the photovoltaic grid-connected system, its grid-connected current can be expressed as:

i_o＝A₂sinω₂t+D_o (2)i _o ＝A ₂ sinω ₂ t+D _o (2)

式中：A₂代表并网电流幅值，ω₂代表并网电流角频率，D_o代表并网电流中的直流分量。而在理想情况下，系统电流的参考信号i_o ^*中不含有直流量，即并网电流为：In the formula: A ₂ represents the amplitude of the grid-connected current, ω ₂ represents the angular frequency of the grid-connected current, and D _o represents the DC component in the grid-connected current. Under ideal conditions, the reference signal i _o ^* of the system current does not contain DC, that is, the grid-connected current is:

i_o ^*＝A₁sinω₁t (3)i _o ^* ＝A ₁ sinω ₁ t (3)

式中：A₁代表参考电流幅值，ω₁代表电网基频。此时，参考信号i_o ^*也可看作由两部分组成，即交流部分参考信号为A₁sinω₁t，而直流部分参考信号为零。In the formula: A ₁ represents the reference current amplitude, ω ₁ represents the fundamental frequency of the power grid. At this time, the reference signal i _o ^* can also be regarded as composed of two parts, that is, the reference signal of the AC part is A ₁ sinω ₁ t, and the reference signal of the DC part is zero.

这样，并网电流环采用PR控制器能够在αβ静止坐标系下实现对交流被控信号的稳态无差控制，但PR控制器对直流量却无法实现零稳态误差，即不能起到抑制直流分量的作用，而PI控制器能弥补其在直流控制方面的不足，故本发明所提出的采用PR+PI双控制器作为电流环控制器，与电网电压锁相环相配合，即能达到单位功率因数并网和直流抑制的双重功效。而该PR+PI双控制器传递函数为：In this way, the grid-connected current loop adopts the PR controller to realize the steady-state error-free control of the AC controlled signal in the αβ static coordinate system, but the PR controller cannot achieve zero steady-state error for the DC flow, that is, it cannot suppress The role of the DC component, and the PI controller can make up for its lack of DC control, so the present invention uses the PR+PI dual controller as the current loop controller, and cooperates with the grid voltage phase-locked loop to achieve Unity power factor grid connection and dual effects of DC suppression. And the transfer function of the PR+PI dual controller is:

${G G}_{ci ci} ((s the s)) = = {G G}_{PR PR + + PI P.I.} ((s the s)) = = {k k}_{p p 11} + + \frac{{k k}_{r r} s the s}{{s the s}^{22} + + {ω ω}_{00}^{22}} + + {k k}_{p p 22} + + \frac{{k k}_{i i}}{s the s} - - - - - - ((44))$

同时，相比PR控制器，准PR控制器能降低电网频率偏移对并网电流的影响，本发明的方案可改进为准PR+PI双控制器的方法以更好地实现上述双重功效，该控制器传递函数为：At the same time, compared with the PR controller, the quasi-PR controller can reduce the impact of grid frequency offset on the grid-connected current, and the solution of the present invention can be improved to a quasi-PR+PI dual-controller method to better realize the above-mentioned dual effects. The controller transfer function is:

${G G}_{q q - - PR PR + + PI P.I.} ((s the s)) = = {k k}_{p p 11} + + \frac{22 {k k}_{r r} {ω ω}_{C C} s the s}{{s the s}^{22} + + 22 {ω ω}_{c c} s the s + + {ω ω}_{00}^{22}} + + {k k}_{p p 22} + + \frac{{k k}_{i i}}{s the s} - - - - - - ((55))$

其中，式(4)和(5)中，k_p1为(准)PR控制器比例系数，k_p2为PI控制器比例系数，ω₀为(准)PR控制器谐振频率，k_r为(准)PR控制器广义积分系数，s为复频率，k_i为PI控制器积分系数，ω_c为准PR控制器截止频率。Among them, in formulas (4) and (5), k _p1 is the proportional coefficient of the (quasi) PR controller, k _p2 is the proportional coefficient of the PI controller, ω ₀ is the resonant frequency of the (quasi) PR controller, k _r is the (quasi) ) PR controller generalized integral coefficient, s is the complex frequency, k _i is the integral coefficient of PI controller, ω _c is the cut-off frequency of quasi-PR controller.

而PR+PI双控制器和准PR+PI双控制器参数的设计可遵循如下原则：The design of PR+PI dual controller and quasi-PR+PI dual controller parameters can follow the following principles:

(2)、选择比例系数k_p1和k_p2。由于k_p1和k_p2对系统性能的影响相同，可将两个比例系数合并，通过两者综合作用的效果，选取比例系数的总取值k_pe，即k_pe＝k_p1+k_p2，此时认为截止频率ω_ci处双控制器的增益约等于k_pe。(2) Select the proportional coefficients k _p1 and k _p2 . Since k _p1 and k _p2 have the same impact on system performance, the two proportional coefficients can be combined, and through the combined effects of the two, the total value of the proportional coefficient k _pe is selected, that is, k _pe = k _p1 + k _p2 , where It is considered that the gain of the dual controllers at the cutoff frequency ω _ci is approximately equal to k _pe .

(3)、根据PI控制的影响范围选择k_i，需将k_i的取值设置较小以减少PI控制器的影响范围。设积分作用主要工作在0～ω_i的频率范围，为减少其对系统相位裕量的影响，在谐振频率ω₀远小于电流环截止频率ω_ci的情况下，可选择ω_i＝(0.1～0.4)ω₀。(3) Select _ki according to the scope of influence of the PI control, and the value of _ki needs to be set smaller to reduce the scope of influence of the PI controller. Assuming that the integral action mainly works in the frequency range of 0～ _ωi , in order to reduce its influence on the system phase margin, when the resonant frequency _ω0 is much smaller than the cut-off frequency _ωci of the current loop, you can choose _ωi = (0.1～ 0.4) ω ₀ .

(4)、选择k_r。在谐振频率附近，主要是双控制器的谐振部分起作用，可忽略比例控制和积分控制，k_r取值可根据经验结合对控制器增益的需求进行选择，并通过校验简化系统补偿后的相位裕量对所选初值进行调节，以达到所需要求。(4) Select k _r . Near the resonant frequency, the resonant part of the dual controller is mainly active, and the proportional control and integral control can be ignored. The value of _kr can be selected according to experience and the demand for controller gain, and the system after compensation can be simplified by checking The phase margin adjusts the selected initial value to meet the required requirements.

(5)、校验相位裕量。根据上述选择参数后的PR+PI双控制器和准PR+PI双控制器对补偿后系统的相位裕量γ_V进行校验，看能否满足30°≤γ_V≤70°，如不满足要求，则重新选择k_r和k_i，直至满足系统设计要求。(5) Check the phase margin. Check the phase margin γ _V of the compensated system according to the PR+PI dual controller and the quasi-PR+PI dual controller after the parameters are selected above to see if it satisfies 30° _≤γV ≤70°, if not Requirements, then re-select k _r and _ki , until the system design requirements are met.

在实施例中，为协助本领域技术人员更好地理解并实现本发明，下面进一步披露本实施例中准PR+PI双控制器中各参数对控制器性能的影响，以指导双控制器的设计。In the embodiment, in order to assist those skilled in the art to better understand and realize the present invention, the following further discloses the influence of each parameter in the quasi-PR+PI dual controller in this embodiment on the performance of the controller, so as to guide the performance of the dual controller design.

首先，如图3所示，比较准PR+PI控制、准PR控制和PI控制三种控制方式的频率特性。从图中可以看出，本文所采用的双控制器中的准PR控制器主要在50Hz处作用，而PI控制器对直流量和交流量都有作用，两者的控制对象相同，控制目标不同，因此在控制器设计时需要考虑两者参数之间的影响。因交流参考信号频率为电网基频，故控制器中谐振频率设置为电网基频，即ω₀＝ω₁。设定谐振频率ω₀后，该双控制器中还有五个参数k_p1、k_r、ω_c、k_p2和k_i需要确定。可设其中四个参数不变，观察第五个参数变化对系统性能的影响。在双控制器中k_p1、k_r、ω_c对系统性能的影响与在准PR控制器中相同。First, as shown in Figure 3, compare the frequency characteristics of the quasi-PR+PI control, quasi-PR control and PI control modes. It can be seen from the figure that the quasi-PR controller of the dual controllers used in this paper mainly works at 50Hz, while the PI controller works on both DC and AC. The control objects of the two are the same, but the control objectives are different , so the influence between the two parameters needs to be considered in the design of the controller. Since the frequency of the AC reference signal is the fundamental frequency of the power grid, the resonant frequency in the controller is set to the fundamental frequency of the power grid, that is, ω ₀ =ω ₁ . After setting the resonant frequency ω ₀ , there are five parameters k _p1 , k _r , ω _c , k _p2 and _ki to be determined in the dual controller. Four of the parameters can be set unchanged, and the influence of the change of the fifth parameter on the system performance can be observed. The influence of k _p1 , k _r , ω _c on the system performance in the dual controller is the same as that in the quasi-PR controller.

如图4所示，ω_c的变化影响控制器在非谐振频率处的增益，两者成正比关系，而在谐振频率处的增益不受影响。但由于受到控制器带宽的限制，对增益的影响主要体现在谐振频率附近，在离谐振频率较远的0Hz附近，控制器增益基本不会受到影响。控制器的带宽随着ω_c的增大而增大，非谐振频率处的相角也随之增大。根据相关标准规定，电力系统正常频率偏差允许值为±0.2Hz，当系统容量较小时，频率偏差可放宽到±0.5Hz。考虑最恶劣情况，可选择带宽为1Hz，则选择ω_c＝πrad/s。As shown in Figure 4, the change of ω _c affects the gain of the controller at the non-resonant frequency, and the two are proportional, while the gain at the resonant frequency is not affected. However, due to the limitation of the bandwidth of the controller, the influence on the gain is mainly reflected in the vicinity of the resonant frequency, and the gain of the controller is basically not affected near 0 Hz, which is far from the resonant frequency. The bandwidth of the controller increases with the increase of _ωc , and the phase angle at the non-resonant frequency also increases. According to relevant standards, the allowable value of the normal frequency deviation of the power system is ±0.2Hz. When the system capacity is small, the frequency deviation can be relaxed to ±0.5Hz. Considering the worst case, the selectable bandwidth is 1Hz, then choose ω _c = πrad/s.

从式(5)可知，两控制器为并联关系，并参看图5(a)、(b)所示k_p1和k_p2变化时双控制器的波特图，k_p1变化对系统的影响与k_p2相同，选取比例系数的总取值k_pe，即k_pe＝k_p1+k_p2。比例系数在整个频率段都有作用，且离谐振频率ω₀越远，比例控制作用越强，系统受谐振控制的影响也相对越弱。若选择电流环开环截止频率ω_ci与ω₀相隔较远，则可认为ω_ci处双控制器增益约为k_pe。在低于谐振频率范围内，LCL滤波器的频率特性与单L滤波器相似，而光伏并网逆变器中网侧电流的控制性能是一种低频运行特性，故在光伏并网系统中可将LCL滤波器等效成电感值为L_T＝L₁+L₂的单L滤波器进行控制器设计。可得电流环开环截止频率处的增益为：It can be known from formula (5) that the two controllers are connected in parallel, and referring to the Bode diagrams of the dual controllers when k _p1 and k _p2 are shown in Figure 5(a) and (b), the influence of k _p1 changes on the system is related to k _p2 is the same, and the total value k _pe of the proportional coefficient is selected, that is, k _pe =k _p1 +k _p2 . The proportional coefficient has an effect in the whole frequency range, and the farther away from the resonant frequency ω ₀ , the stronger the effect of the proportional control, and the weaker the influence of the system by the resonant control. If the cut-off frequency ω _ci of the open loop of the current loop is selected to be far away from ω ₀ , it can be considered that the gain of the dual controller at ω _ci is about k _pe . In the range below the resonant frequency, the frequency characteristic of the LCL filter is similar to that of the single L filter, and the control performance of the grid-side current in the photovoltaic grid-connected inverter is a low-frequency operation characteristic, so it can be used in the photovoltaic grid-connected system The LCL filter is equivalent to a single-L filter with an inductance value L _T =L ₁ +L ₂ for controller design. The gain at the open-loop cutoff frequency of the current loop can be obtained as:

$2020 lg lg | | {T T}_{oi oi} ((j j {ω ω}_{ci ci})) | | = = 2020 lg lg | | \frac{{k k}_{pe pe} {k k}_{PWM PWM}}{{L L}_{T T} {ω ω}_{ci ci}} | | = = 00$

由上式可得k_pe的计算公式为：From the above formula, the calculation formula of k _pe can be obtained as:

${k k}_{pe pe} = = \frac{{L L}_{T T} {ω ω}_{ci ci}}{{k k}_{PWM PWM}}$

现需对k_i变化对系统性能的影响进行分析。其它参数固定，k_i变化时，控制器频率特性如图6所示。从图中可知，k_i取值的变化影响了积分环节的作用范围，但是对谐振频率处的增益不影响，随着k_i的增加，积分作用的范围变大，对谐振频率处相角的影响增大，从而影响系统的相位裕量。要减少PI控制器的影响范围，需将k_i的取值设置较小。设积分作用主要工作在0～ω_i的频率范围，为减少其对系统相位裕量的影响，在谐振频率ω₀远小于电流环截止频率ω_ci的情况下，可选择ω_i＝(0.1～0.4)ω₀。Now it is necessary to analyze the impact of _ki changes on system performance. When the other parameters are fixed and k _i changes, the frequency characteristics of the controller are shown in Figure 6. It can be seen from the figure that the change of the value of _ki affects the action range of the integral link, but has no effect on the gain at the resonance frequency. With the increase of _ki , the range of the integral action becomes larger, and the effect on the phase angle at the resonance frequency The effect increases, thereby affecting the phase margin of the system. To reduce the range of influence of the PI controller, the value of _ki needs to be set smaller. Assuming that the integral action mainly works in the frequency range of 0～ _ωi , in order to reduce its influence on the system phase margin, when the resonant frequency _ω0 is much smaller than the cut-off frequency _ωci of the current loop, we can choose _ωi = (0.1～ 0.4) ω ₀ .

由于k_r的变化对接近0Hz的频率段不影响，而在谐振频率处，随着k_r增大时，谐振控制器作用增强，谐振峰处增益增大，但控制器带宽不变，谐振频率附近相角绝对值增大，将会影响系统的相位裕量，参见图7，且在谐振频率处主要为双控制器中谐振部分起作用，可忽略比例控制和积分控制，则电流环在谐振频率ω₀处的开环增益为：Because the change of k _r has no effect on the frequency range close to 0Hz, but at the resonant frequency, as k _r increases, the effect of the resonant controller is enhanced, and the gain at the resonant peak increases, but the bandwidth of the controller remains unchanged, and the resonant frequency The increase of the absolute value of the nearby phase angle will affect the phase margin of the system, see Figure 7, and at the resonant frequency, it is mainly the resonant part of the dual controller that plays a role, and the proportional control and integral control can be ignored, then the current loop is at the resonant The open loop gain at frequency _ω0 is:

$| | {T T}_{oi oi} ((j j {ω ω}_{00})) | | = = | | \frac{22 {k k}_{r r} {ω ω}_{c c} {ω ω}_{00} j j}{- - {ω ω}_{00}^{22} + + 22 {ω ω}_{c c} {ω ω}_{00} j j + + {ω ω}_{00}^{22}} \cdot \cdot \frac{{k k}_{PWM PWM}}{{L L}_{T T} {ω ω}_{00} j j} | | = = \frac{{k k}_{r r} {k k}_{PWM PWM}}{{L L}_{T T} {ω ω}_{00}}$

由上式可得k_r的计算公式为：From the above formula, the calculation formula of k _r can be obtained as:

${k k}_{r r} = = \frac{| | {T T}_{oi oi} ((j j {ω ω}_{00})) | | {L L}_{T T} {ω ω}_{00}}{{k k}_{PWM PWM}}$

k_r的取值可根据经验结合对控制器增益的需求进行选择，在已确定ω_c、k_pe和k_i的值后，k_r的取值的变化直接影响系统的相位裕量，k_r越大，相位裕量越小，因此可以通过校验简化系统补偿后的相位裕量对初值进行调节，以达到所需要求。The value of k _r can be selected based on experience and the demand for controller gain. After the values of ω _c , k _pe and _ki have been determined, the change of the value of k _r directly affects the phase margin of the system, k _r The larger the , the smaller the phase margin, so the initial value can be adjusted by checking the phase margin after the simplified system compensation to meet the required requirements.

前述控制器设计所选控制对象为简化后系统模型，但LCL滤波逆变系统中采用的有源阻尼策略会影响开环截止频率处的相角，因此需要校验补偿后原系统的相位裕量γ_V，看能否满足30°≤γ_V≤70°，如不满足要求，则重新选择k_r和k_i，直至满足系统设计要求。The control object selected in the aforementioned controller design is the simplified system model, but the active damping strategy adopted in the LCL filter inverter system will affect the phase angle at the open-loop cut-off frequency, so it is necessary to verify the phase margin of the original system after compensation γ _V , see if 30°≤γ _V ≤70° can be satisfied, if not, reselect k _r and _ki until the system design requirements are met.

本发明的正确性和可行性通过了MATLAB/Simulink软件仿真和RT-LAB平台实验的验证。The correctness and feasibility of the present invention are verified by MATLAB/Simulink software simulation and RT-LAB platform experiment.

1、采用MATLAB/Simulink对所提出的方法进行仿真研究，仿真参数为：系统功率3.84kW，开关频率10kHz，直流侧电压V_DC＝400V，电网电压220V/50Hz，并网电流额定峰值24.68A，桥侧电感3.33mH，网侧电感0.67mH，滤波电容C_f＝5μF，电容电流反馈系数k_f＝0.126，双控制器参数：k_pe＝0.063，k_r＝25，ω_c＝πrad/s，k_i＝0.2，ω₀＝62.83。1. Use MATLAB/Simulink to conduct simulation research on the proposed method. The simulation parameters are: system power 3.84kW, switching frequency 10kHz, DC side voltage V _DC = 400V, grid voltage 220V/50Hz, grid-connected current rated peak value 24.68A, Bridge-side inductance 3.33mH, grid-side inductance 0.67mH, filter capacitor C _f ＝5μF, capacitor current feedback coefficient k _f ＝0.126, dual controller parameters: k _pe ＝0.063, k _r ＝25, ω _c ＝πrad/s, k _i =0.2, ω ₀ =62.83.

(1)、首先在仿真模型中加入直流注入模拟环节但不采取直流抑制措施，检验直流注入模拟效果。在直流注入现象模拟时，采用在调制信号中加入少量直流偏置的方法，这样可以使逆变器中开关器件的通断时间不一致，从而间接模拟器件特性不同等因素造成的直流注入，且不会对模型中的其它环节造成影响。并网电流控制器采用准PR控制器，仿真结果如图8(a)所示，从图中可看出，并网电流正弦度良好，为了便于并网电流与电网电压的相位比较，将电网电压缩小5倍后送入示波器，从两者波形中可看出，两者的重合点不在电网电压过零点，说明并网电流有一定偏置。在系统达到稳定状态时对并网电流进行频谱分析，结果如图8(b)，并网电流基波幅值为24.65A，THD为0.70％，符合国标中相关要求，但是直流偏置含量为5％～6％，大于国标中1％的技术指标，这种情况下，不宜将电流并入电网。(1) First, add the DC injection simulation link in the simulation model but do not take DC suppression measures to check the DC injection simulation effect. When simulating the DC injection phenomenon, the method of adding a small amount of DC bias to the modulation signal can make the on-off time of the switching device in the inverter inconsistent, thereby indirectly simulating the DC injection caused by factors such as different device characteristics, and does not affect other parts of the model. The grid-connected current controller adopts a quasi-PR controller. The simulation results are shown in Figure 8(a). It can be seen from the figure that the grid-connected current has a good sinusoidal degree. In order to facilitate the phase comparison between the grid-connected current and the grid voltage, the grid After the voltage is reduced by 5 times, it is sent to the oscilloscope. It can be seen from the waveforms of the two that the coincidence point of the two is not at the zero crossing point of the grid voltage, indicating that the grid-connected current has a certain bias. When the system reaches a steady state, the spectrum analysis of the grid-connected current is carried out. The result is shown in Figure 8(b). The fundamental wave amplitude of the grid-connected current is 24.65A, and the THD is 0.70%. 5% to 6%, which is greater than the technical index of 1% in the national standard. In this case, it is not suitable to incorporate the current into the grid.

(2)、验证本发明所提出的双控制器直流注入抑制方法的效果，并网电流控制器采用准PR+PI双控制器，仿真结果如图9(a)所示，波形上可以看出，采用双控制器后，并网电流波形正弦度良好，且与电网电压波形在过零点重合，能够跟踪电网电压相位，系统的并网效果良好。在与未采取直流抑制措施相同的情况下对并网电流频谱分析，结果如图9(b)，与图8(b)比较能明显看出直流分量得到了较好的抑制，远小于1％的并网要求，且并网电流THD含量为0.73％，也能满足国标要求，说明系统成功实现了并网和直流抑制的双重功效。(2), verify the effect of the dual-controller DC injection suppression method proposed in the present invention, the grid-connected current controller adopts the quasi-PR+PI dual controller, the simulation result is shown in Figure 9 (a), and it can be seen from the waveform , after adopting dual controllers, the grid-connected current waveform has a good sine degree, and coincides with the grid voltage waveform at the zero-crossing point, which can track the grid voltage phase, and the grid-connected effect of the system is good. The grid-connected current spectrum is analyzed under the same conditions as without DC suppression measures, and the results are shown in Figure 9(b). Compared with Figure 8(b), it can be clearly seen that the DC component has been better suppressed, far less than 1%. The grid-connected requirements, and the grid-connected current THD content is 0.73%, which can also meet the national standard requirements, indicating that the system has successfully realized the dual functions of grid-connected and DC suppression.

2、为了验证所提出方法的可行性，在RT-LAB实验平台进行了功能验证性实验。搭建了单相并网逆变实验系统，直流侧电压由稳压电压源提供，直流电压恒定为30V，交流侧通过自耦调压器调压后与实验电路相连接，调压器将交流侧电压幅值调节为15V；采用日本三菱公司智能功率模块IPM的U、V桥臂作为单相全桥逆变桥，开关频率为8.3kHz，SPWM调制的死区时间为2μs；电流测量使用泰克A622电流探头，测量档位为100mV/A。滤波器参数为：逆变桥侧电感L₁＝5.21mH，电网侧电感L₂＝1.52mH，滤波电容C_f＝3.29μF，并网电流额定峰值2.25A。2. In order to verify the feasibility of the proposed method, a functional verification experiment was carried out on the RT-LAB experimental platform. A single-phase grid-connected inverter experiment system was built. The voltage on the DC side was provided by a regulated voltage source, and the DC voltage was constant at 30V. The AC side was regulated by an auto-coupling voltage regulator and connected to the experimental circuit. The voltage amplitude is adjusted to 15V; the U and V bridge arms of the intelligent power module IPM of Mitsubishi Corporation of Japan are used as the single-phase full-bridge inverter bridge, the switching frequency is 8.3kHz, and the dead time of SPWM modulation is 2μs; the current measurement uses Tektronix A622 Current probe, the measurement range is 100mV/A. The filter parameters are: inverter bridge side inductance L ₁ =5.21mH, grid side inductance L ₂ =1.52mH, filter capacitor C _f =3.29μF, grid-connected current rated peak value 2.25A.

(1)、未采用直流抑制方法进行实验，即并网电流控制器采用准PR控制。图10(a)为并网侧电压和并网电流波形，波形显示并网电流有明显直流偏置。对并网电流进行FFT分析，重点观察直流分量与基频分量之间的比值，结果如图10(b)所示，图中椭圆标识出的数据为并网电流中对应频率分量的有效值，计算出直流分量与基频分量之间的比值达到21％。通过对实验电路中相关部分进行检测、分析，得知该直流偏置不是传感器造成，其主要原因可能与IPM模块中器件不对称，控制信号产生模块等相关。(1) The DC suppression method is not used for experiments, that is, the grid-connected current controller adopts quasi-PR control. Figure 10(a) shows the grid-connected side voltage and grid-connected current waveforms, and the waveforms show that the grid-connected current has obvious DC bias. Carry out FFT analysis on the grid-connected current, and focus on observing the ratio between the DC component and the fundamental frequency component. The result is shown in Figure 10(b). The data marked by the ellipse in the figure is the effective value of the corresponding frequency component in the grid-connected current. It is calculated that the ratio between the DC component and the fundamental frequency component reaches 21%. Through the detection and analysis of the relevant parts in the experimental circuit, it is known that the DC bias is not caused by the sensor, and the main reason may be related to the asymmetry of the devices in the IPM module and the control signal generation module.

(2)、采用直流注入抑制方法进行实验，即采用准PR+PI双控制器作为并网电流控制器。图11(a)为实验时并网侧电压和并网电流的对比波形，从波形能看出进行直流注入抑制后，并网电流波形正弦度良好，相位上仍能较好地跟踪并网侧电压。并网电流的直流偏置明显得到了改善，对其进行FFT分析及直流含量分析的结果如图11(b)，图中右上角数据显示直流分量与基波分量之间比值为0.24％，小于国标所要求的1％的限定值。由此可知，并网控制器设计正确，且所采用的双控制器直流注入抑制方法效果明显。(2) Experiments are carried out using the DC injection suppression method, that is, the quasi-PR+PI dual controller is used as the grid-connected current controller. Figure 11(a) is the comparative waveform of the grid-connected side voltage and grid-connected current during the experiment. It can be seen from the waveform that after the DC injection is suppressed, the grid-connected current waveform has a good sine degree, and the phase can still track the grid-connected side well. Voltage. The DC bias of the grid-connected current has been significantly improved. The results of FFT analysis and DC content analysis are shown in Figure 11(b). The data in the upper right corner of the figure shows that the ratio between the DC component and the fundamental component is 0.24%, which is less than The limit value of 1% required by the national standard. It can be seen that the design of the grid-connected controller is correct, and the effect of the dual-controller DC injection suppression method is obvious.

上述实施例为本发明较佳的实施方式，但本发明的实施方式并不受上述实施例的限制，其他的任何未背离本发明的精神实质与原理下所作的改变、修饰、替代、组合、简化，均应为等效的置换方式，都包含在本发明的保护范围之内。The above-mentioned embodiment is a preferred embodiment of the present invention, but the embodiment of the present invention is not limited by the above-mentioned embodiment, and any other changes, modifications, substitutions, combinations, Simplifications should be equivalent replacement methods, and all are included in the protection scope of the present invention.

Claims

1. a grid-connected direct current injects the inhibition method, it is characterized in that: this method adopts with PR+PI or the accurate PR+PI dual controller control strategy as the grid-connected current ring controller, causes that direct current injects and suppresses the action of converter bridge switching parts device is inconsistent by FEEDBACK CONTROL;

Wherein, the transfer function of PR+PI dual controller is:

G_{ci} (s) = G_{PR + PI} (s) = k_{p 1} + \frac{k_{r} s}{s^{2} + {ω_{0}}^{2}} + k_{p 2} + \frac{k_{i}}{s};

The transfer function of accurate PR+PI dual controller is:

G_{q - PR + PI} (s) = k_{p 1} + \frac{2 k_{r} ω_{C} s}{s^{2} + 2 ω_{c} s + {ω_{0}}^{2}} + k_{p 2} + \frac{k_{i}}{s};

In the formula: k _P1Be (standard) PR controller proportionality coefficient, k _P2Be PI controller proportionality coefficient, ω ₀Be (standard) PR controller resonance frequency, k _rBe (standard) PR controller improper integral coefficient, s is complex frequency, k _iBe PI controller integral coefficient, ω _cPR controller cut-off frequency is as the criterion.

2. grid-connected direct current according to claim 1 injects the inhibition method, and it is characterized in that: the parameter of described PR+PI or accurate PR+PI dual controller specifically meets the following conditions:

At first, consideration mains frequency deviation is chosen the bandwidth omega of accurate PR controller _c/ π=1Hz, then ω _c=π rad/s;

Then, at ω ₀Much smaller than electric current loop cut-off frequency ω _CiSituation, the selection percentage coefficient k _P1, k _P2Total value k _Pe=k _P1+ k _P2Equal controller at ω _CiThe gain that the place need reach, and select k _iTo satisfy ω _i=(0.1～0.4) ω ₀

At last, select k according to phase margin _r, make the phase margin γ that compensates the back system _VSatisfy 30 °≤γ _V≤ 70 °.