[go: up one dir, main page]

CN102130883A - A method for time-frequency synchronization in TD-LTE system - Google Patents

A method for time-frequency synchronization in TD-LTE system Download PDF

Info

Publication number
CN102130883A
CN102130883A CN2011100947597A CN201110094759A CN102130883A CN 102130883 A CN102130883 A CN 102130883A CN 2011100947597 A CN2011100947597 A CN 2011100947597A CN 201110094759 A CN201110094759 A CN 201110094759A CN 102130883 A CN102130883 A CN 102130883A
Authority
CN
China
Prior art keywords
time
synchronization
signal
frequency
value
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Granted
Application number
CN2011100947597A
Other languages
Chinese (zh)
Other versions
CN102130883B (en
Inventor
陈发堂
马磊
李小文
王丹
王华华
刘宇
许彦斌
凌云志
黄武
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Chongqing University of Post and Telecommunications
CETC 41 Institute
Original Assignee
Chongqing University of Post and Telecommunications
CETC 41 Institute
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Chongqing University of Post and Telecommunications, CETC 41 Institute filed Critical Chongqing University of Post and Telecommunications
Priority to CN 201110094759 priority Critical patent/CN102130883B/en
Publication of CN102130883A publication Critical patent/CN102130883A/en
Application granted granted Critical
Publication of CN102130883B publication Critical patent/CN102130883B/en
Expired - Fee Related legal-status Critical Current
Anticipated expiration legal-status Critical

Links

Images

Landscapes

  • Synchronisation In Digital Transmission Systems (AREA)

Abstract

The invention discloses a method for resolving time synchronization and decimal frequency synchronization in a time division long-term evolution (TD-LTE) system by using weighted average and relates to the technical field of mobile communication. Aiming at the synchronization problem in the TD-LTE system in the prior art, the method comprises the following steps of: delaying an orthogonal frequency division multiplexing (OFDM) symbol sampling point for a baseband digital signal which is received at a receiving end, performing normalized autocorrelation in a time domain to generate a peak value, and obtaining a time synchronization point and decimal frequency offset; performing cross-correlation on the baseband digital signal and a locally generated primary synchronization signal (PSS) in the time domain, and obtaining higher-accuracy time synchronization point and decimal frequency offset according to a peak value which is generated trough the cross-correlation; and obtaining time-domain channel estimation through the PSS signal by using non-coherent detection, performing weighted average in time synchronization and decimal frequency synchronization which are obtained in a simplified maximum likelihood (ML) algorithm and the PSS respectively according to channel impulse response length so as to adapt to environmental change of the channel and effectively acquire stable-performance time-frequency synchronization. The invention provides a method for time synchronization and decimal frequency synchronization which adapt to channel environment.

Description

一种用于TD-LTE系统时频同步的方法A method for time-frequency synchronization in TD-LTE system

技术领域technical field

本发明涉及移动通信技术领域，具体涉及第三代移动通信长期演进系统（以下简称TD-LTE）中时间同步和小数倍频率同步的方法。The present invention relates to the technical field of mobile communication, in particular to a method for time synchronization and fractional multiple frequency synchronization in the third-generation mobile communication long-term evolution system (hereinafter referred to as TD-LTE).

背景技术Background technique

3GPP(3rd Generation Partnership Project)组织于2005年3月启动了空中技术的长期演进工作。LTE的目标是以OFDMA(Orthogonal Frequency Division Multiplexing Access)多址接入和多天线为主要技术基础。在TD-LTE系统中，下行方向上多址接入技术采用了正交频分复用OFDM(Orthogonal Frequency Division Multiplexing)技术，该技术具有高频谱效率、高峰值速率、能够很好地抵抗信道间干扰、频率选择性衰落和脉冲噪声等，当终端采用2天线接收，在20MHz的载波带宽情况下，瞬时峰值速率可以到达100Mbit/s(频谱效率为5bit/s/Hz)。但是对于OFDM技术，一方面符号的到达时间对于接收端是未知的，因此需要进行时间同步，也就是确定FFT（快速傅里叶变换）的窗口。另一方面TD-LTE系统中采用的相互正交的子载波技术，因此OFDM对载波频率偏移非常敏感，载波频率偏移主要是由于发射机和接收机之间的晶体振荡器频率不匹配和多普勒频移所引起的。载波频率偏移可以分为两部分，一是小数倍频偏(FFO)，即小于子载波间隔的部分，二是整数倍频偏(IFO)，即子载波间隔的整数部分。在OFDM系统中，整数倍频偏不会像小数倍频偏那样会破坏子载波之间的正交性，而小数部分则会造成子信道干扰，破坏各子载波间的正交性，导致了系统的误码率增加。The 3GPP (3rd Generation Partnership Project) organization started the long-term evolution of air technology in March 2005. The goal of LTE is based on OFDMA (Orthogonal Frequency Division Multiplexing Access) multiple access and multiple antennas. In the TD-LTE system, the multiple access technology in the downlink direction adopts OFDM (Orthogonal Frequency Division Multiplexing) technology, which has high spectral efficiency, high peak rate, and can well resist inter-channel Interference, frequency selective fading, and impulse noise, etc., when the terminal uses 2 antennas to receive, the instantaneous peak rate can reach 100Mbit/s (the spectral efficiency is 5bit/s/Hz) in the case of a carrier bandwidth of 20MHz. But for OFDM technology, on the one hand, the arrival time of symbols is unknown to the receiving end, so time synchronization is required, that is, to determine the window of FFT (Fast Fourier Transform). On the other hand, the mutually orthogonal sub-carrier technology used in TD-LTE system, so OFDM is very sensitive to carrier frequency offset, which is mainly due to the crystal oscillator frequency mismatch between the transmitter and receiver and caused by Doppler shift. The carrier frequency offset can be divided into two parts, one is the fractional frequency offset (FFO), which is the part smaller than the subcarrier spacing, and the other is the integer multiple frequency offset (IFO), which is the integer part of the subcarrier spacing. In an OFDM system, the integer frequency offset will not destroy the orthogonality between subcarriers like the fractional frequency offset, but the fractional part will cause subchannel interference and destroy the orthogonality between subcarriers, resulting in The bit error rate of the system increases.

在TD-LTE系统中采用帧结构类型2，适用于时分双工(Time-division duplex, TDD)模式，图1示出TDD的帧结构，其中，每个无线帧长为10ms，包括两个长度为5ms的半帧，每个半帧包括五个长度为1ms的子帧，支持5ms和10ms的上下行切换周期，在5ms周期中，子帧1和子帧6固定配置为特殊子帧，每一个特殊子帧由下行导频时隙(DwPTS)、保护导频(GP)和上行导频时隙(UpPTS)3个特殊时隙组成，其中，主同步信号(Primary Synchronization Signal, PSS)位于子帧1、6的第三个OFDM符号，辅同步信号(Secondary Synchronization Signal, SSS)位于子帧0、5的最后一个OFDM符号。PSS和SSS信号的位置相对固定，与TD-LTE系统的上下行子帧配置、小区覆盖大小等因素无关。另外，TD-LTE系统支持多种传输带宽配置，为了保证各个系统带宽下PSS和SSS位置的相对固定和检测算法的实现简化，PSS和SSS信号在频率上总是处于整个系统带宽中央1.08MHz (6个物理资源块(PRB, Physical Resource Block))的位置，图2示出长度为62的PSS序列映射至直流载波附近的62个子载波上，序列两端各有5个子载波未使用，其中，中间被打孔打掉的元素是为了避免直流载波(DC)。The frame structure type 2 is adopted in the TD-LTE system, which is applicable to the Time-division duplex (TDD) mode. Figure 1 shows the frame structure of TDD, wherein each wireless frame is 10ms long, including two lengths It is a half frame of 5ms, each half frame includes five subframes with a length of 1ms, and supports uplink and downlink switching periods of 5ms and 10ms. In the 5ms period, subframe 1 and subframe 6 are fixedly configured as special subframes, each The special subframe is composed of three special time slots: downlink pilot time slot (DwPTS), guard pilot frequency (GP) and uplink pilot time slot (UpPTS), in which the primary synchronization signal (Primary Synchronization Signal, PSS) is located in the subframe In the third OFDM symbols of 1 and 6, the Secondary Synchronization Signal (SSS) is located in the last OFDM symbols of subframes 0 and 5. The positions of the PSS and SSS signals are relatively fixed, and have nothing to do with factors such as the uplink and downlink subframe configuration and cell coverage size of the TD-LTE system. In addition, the TD-LTE system supports a variety of transmission bandwidth configurations. In order to ensure the relative fixation of the PSS and SSS positions under each system bandwidth and the simplification of the detection algorithm, the frequency of the PSS and SSS signals is always in the center of the entire system bandwidth 1.08MHz ( The positions of 6 physical resource blocks (PRB, Physical Resource Block)), Figure 2 shows that the PSS sequence with a length of 62 is mapped to 62 subcarriers near the DC carrier, and there are 5 subcarriers at both ends of the sequence that are not used, wherein, The elements that are punched out in the middle are to avoid direct current carrier (DC).

现有的用于TD-LTE系统时间同步和小数倍频率同步的最普遍的方法主要分为两类：基于循环前缀(Cycle Prefix, CP)的方法和数据辅助的方法。基于CP的方法主要是利用OFDM符号的结构特征，利用自相关的峰值进行时间同步，进而得到小数倍频偏，但是由ML算法来估计时间同步和频率同步的公式可以知道，传统ML算法计算复杂度非常高，这里的计算复杂度以计算一个OFDM符号时间同步所需要的复数乘法和复数加法的次数来度量，考虑子载波数                                               

Figure 2011100947597100002DEST_PATH_IMAGE002
，CP的长度
Figure 2011100947597100002DEST_PATH_IMAGE004
的OFDM系统，因此每个OFDM符号的实际长度就为
Figure 2011100947597100002DEST_PATH_IMAGE006
个样值。观察接收到
Figure 2011100947597100002DEST_PATH_IMAGE008
个连续样值
Figure 2011100947597100002DEST_PATH_IMAGE010
的基带信号，其中这些样值中包括一个完整的
Figure 10555DEST_PATH_IMAGE006
个样值的OFDM符号，那么计算一个OFDM符号的时间同步需要144*2048+144*2*2048=884736次复数乘法和144*2048+144*2*2048=884736次复数加法，而简化后的ML算法计算一个OFDM符号的时间同步需要36+127*6+72+12*127=2394次复数乘法和36+127*6+72+12*127=2394次复数加法，可见简化的ML算法大大减小了计算复杂度。基于数据辅助的方法主要是利用TD-LTE系统无线帧中的PSS信号良好的相关性来估计时间同步和频率同步，但是TD-LTE系统无线帧中，并不是每个子帧，每个OFDM符号都存在PSS信号，PSS信号只有子帧1和子帧6第3个OFDM符号的固定位置才会出现，因此在多普勒和噪声干扰不是很大的情况下，如果仍然用PSS来进行时间同步和频率同步的话，会增加计算复杂度、增大了存储空间和需要较长的处理时间。综上所述，现有TD-LTE系统缺少一种有效且简单的实现时间同步和小数倍频率同步的方法。The most common existing methods for TD-LTE system time synchronization and fractional multiple frequency synchronization are mainly divided into two categories: a method based on a cyclic prefix (Cycle Prefix, CP) and a data-assisted method. The CP-based method mainly uses the structural characteristics of OFDM symbols, and uses the peak value of autocorrelation for time synchronization, and then obtains the fractional multiple frequency offset. However, the formula for estimating time synchronization and frequency synchronization by the ML algorithm can be known. The complexity is very high. The computational complexity here is measured by the number of complex multiplications and complex additions required to calculate the time synchronization of an OFDM symbol, considering the number of subcarriers
Figure 2011100947597100002DEST_PATH_IMAGE002
, the length of CP
Figure 2011100947597100002DEST_PATH_IMAGE004
OFDM system, so the actual length of each OFDM symbol is
Figure 2011100947597100002DEST_PATH_IMAGE006
samples. Observation received
Figure 2011100947597100002DEST_PATH_IMAGE008
consecutive samples
Figure 2011100947597100002DEST_PATH_IMAGE010
baseband signal, where these samples include a complete
Figure 10555DEST_PATH_IMAGE006
sample OFDM symbols, then calculating the time synchronization of an OFDM symbol requires 144*2048+144*2*2048=884736 complex multiplications and 144*2048+144*2*2048=884736 complex additions, and the simplified The ML algorithm needs 36+127*6+72+12*127=2394 complex multiplications and 36+127*6+72+12*127=2394 complex additions to calculate the time synchronization of an OFDM symbol. It can be seen that the simplified ML algorithm is greatly Reduced computational complexity. The data-assisted method mainly uses the good correlation of the PSS signal in the radio frame of the TD-LTE system to estimate time synchronization and frequency synchronization, but in the radio frame of the TD-LTE system, not every subframe, every OFDM symbol There is a PSS signal, and the PSS signal will only appear at the fixed position of the third OFDM symbol in subframe 1 and subframe 6. Therefore, if Doppler and noise interference are not very large, if PSS is still used for time synchronization and frequency Synchronization increases computational complexity, increases storage space, and requires longer processing time. To sum up, the existing TD-LTE system lacks an effective and simple method for realizing time synchronization and fractional multiple frequency synchronization.

发明内容Contents of the invention

本发明针对现有技术TD-LTE系统在实现时间同步和小数倍频率同步时计算复杂，处理时间长等缺陷，本发明提出一种时间同步和小数倍频率同步方法。根据实际信道环境自适应的用简化的最大似然(ML)算法和PSS分别得到的时间同步点和小数倍频偏进行加权平均，具体包括如下步骤：The present invention aims at defects such as complicated calculation and long processing time in realizing time synchronization and fractional multiple frequency synchronization of the prior art TD-LTE system, and the present invention proposes a method for time synchronization and fractional multiple frequency synchronization. According to the actual channel environment, the weighted average of the time synchronization points and fractional frequency offsets obtained respectively by the simplified maximum likelihood (ML) algorithm and PSS is carried out, which specifically includes the following steps:

(1) 将接收端接收到的基带数字信号进行降频处理，延时一个OFDM符号抽样点数，并在时域内进行归一化自相关，基于归一化自相关产生的峰值得到时间同步点，进而根据时间同步点得到小数倍频偏估计值；(1) Perform down-frequency processing on the baseband digital signal received by the receiving end, delay one OFDM symbol sampling point, and perform normalized autocorrelation in the time domain, and obtain the time synchronization point based on the peak value generated by the normalized autocorrelation, Then, an estimated fractional multiple frequency offset is obtained according to the time synchronization point;

(2) 将基带数字信号与本地生成的主同步信号在时域内进行互相关，根据互相关产生的峰值得到精确度更高的时间同步点和小数倍频偏；(2) Cross-correlate the baseband digital signal with the locally generated main synchronization signal in the time domain, and obtain a more accurate time synchronization point and fractional frequency offset according to the peak value generated by the cross-correlation;

(3) 确定接收端主同步信号的位置，根据PSS信号采用非相干检测得到时域里的信道估计值，根据信道冲激响应长度，确定加权系数，自适应的对用步骤（1）和步骤（2）分别得到的同步时间点和小数倍频偏进行加权平均；(3) Determine the position of the main synchronization signal at the receiving end, use non-coherent detection to obtain the channel estimation value in the time domain according to the PSS signal, determine the weighting coefficient according to the length of the channel impulse response, and adaptively use steps (1) and steps (2) Weighted average of the synchronization time points and fractional frequency offsets obtained respectively;

(4) 利用所得小数倍频偏估计对接收端的基带数字信号进行小数倍频偏校正。(4) Use the obtained fractional frequency offset estimation to perform fractional frequency offset correction on the baseband digital signal at the receiving end.

通过降频处理降低复杂度，所述降频处理进一步包括：The complexity is reduced by down-frequency processing, and the down-frequency processing further includes:

(a) 将接收端接收到的基带信号采用降采样(可采用1/2、1/4或1/8降采样)，即每2、4、8个符号抽样一个符号。以下以1/4降采样为例进行说明。计算公式和公式(a) The baseband signal received by the receiving end is down-sampled (1/2, 1/4 or 1/8 down-sampling can be used), that is, one symbol is sampled every 2, 4, or 8 symbols. The following takes 1/4 downsampling as an example for illustration. Calculation formula and the formula

Figure 2011100947597100002DEST_PATH_IMAGE014
在
Figure 2011100947597100002DEST_PATH_IMAGE016
时，
Figure 2011100947597100002DEST_PATH_IMAGE018
和
Figure 2011100947597100002DEST_PATH_IMAGE020
的值并保存，作为采用递归的方法计算下一个值的初始值。其中，
Figure 2011100947597100002DEST_PATH_IMAGE024
为时域序号，
Figure 2011100947597100002DEST_PATH_IMAGE026
为CP的长度，
Figure 2011100947597100002DEST_PATH_IMAGE028
为一个OFDM符号的抽样点数，
Figure 2011100947597100002DEST_PATH_IMAGE030
为经过1/4降采样后的基带数字信号，
Figure 803062DEST_PATH_IMAGE018
为当前
Figure 582799DEST_PATH_IMAGE022
值
Figure 2011100947597100002DEST_PATH_IMAGE032
个相距为
Figure 40325DEST_PATH_IMAGE028
的样值对之间相关值之和，
Figure 854698DEST_PATH_IMAGE020
为当前
Figure 677160DEST_PATH_IMAGE022
值独立于频率偏差的能量项；
Figure 2011100947597100002DEST_PATH_IMAGE014
exist
Figure 2011100947597100002DEST_PATH_IMAGE016
hour,
Figure 2011100947597100002DEST_PATH_IMAGE018
and
Figure 2011100947597100002DEST_PATH_IMAGE020
and save the value as a recursive method to calculate the next The initial value of the value. in,
Figure 2011100947597100002DEST_PATH_IMAGE024
is the sequence number in the time domain,
Figure 2011100947597100002DEST_PATH_IMAGE026
is the length of CP,
Figure 2011100947597100002DEST_PATH_IMAGE028
is the number of sampling points for one OFDM symbol,
Figure 2011100947597100002DEST_PATH_IMAGE030
is the baseband digital signal after 1/4 downsampling,
Figure 803062DEST_PATH_IMAGE018
for the current
Figure 582799DEST_PATH_IMAGE022
value
Figure 2011100947597100002DEST_PATH_IMAGE032
a distance of
Figure 40325DEST_PATH_IMAGE028
The sum of the correlation values between the sample value pairs,
Figure 854698DEST_PATH_IMAGE020
for the current
Figure 677160DEST_PATH_IMAGE022
an energy term whose value is independent of the frequency deviation;

(b) 由终端运算能力确定滑动步长，如以滑动步长为16(或32)，采用递归的方法计算下一个

Figure 944193DEST_PATH_IMAGE022
值(也就是)对应的
Figure 2011100947597100002DEST_PATH_IMAGE036
和
Figure 2011100947597100002DEST_PATH_IMAGE038
的值，即调用以下公式计算：(b) The sliding step size is determined by the computing power of the terminal. For example, the sliding step size is 16 (or 32), and the next step is calculated recursively.
Figure 944193DEST_PATH_IMAGE022
value (i.e. )corresponding
Figure 2011100947597100002DEST_PATH_IMAGE036
and
Figure 2011100947597100002DEST_PATH_IMAGE038
The value of , which is calculated by calling the following formula:

Figure 2011100947597100002DEST_PATH_IMAGE040
和
Figure 2011100947597100002DEST_PATH_IMAGE040
and

Figure 2011100947597100002DEST_PATH_IMAGE042
，其中，
Figure 956143DEST_PATH_IMAGE036
，
Figure 687339DEST_PATH_IMAGE038
分别为下一个
Figure 680702DEST_PATH_IMAGE022
值的相关值之和和能量项，则计算每个
Figure 435032DEST_PATH_IMAGE022
值的
Figure 172044DEST_PATH_IMAGE018
只需要2次复数相乘和2次复数相加，计算每个
Figure 695429DEST_PATH_IMAGE022
值的
Figure 125273DEST_PATH_IMAGE020
只需要4次复数相乘和4次复数相加，可以得到经过简化后的方法大大的减小了传统ML算法的计算复杂度；
Figure 2011100947597100002DEST_PATH_IMAGE042
,in,
Figure 956143DEST_PATH_IMAGE036
,
Figure 687339DEST_PATH_IMAGE038
respectively for the next
Figure 680702DEST_PATH_IMAGE022
The sum of the associated values of the values and the energy term are calculated for each
Figure 435032DEST_PATH_IMAGE022
worth it
Figure 172044DEST_PATH_IMAGE018
Only 2 complex multiplications and 2 complex additions are required to calculate each
Figure 695429DEST_PATH_IMAGE022
worth it
Figure 125273DEST_PATH_IMAGE020
Only 4 complex multiplications and 4 complex additions are required, and the simplified method can greatly reduce the computational complexity of the traditional ML algorithm;

(c) 根据步骤(a)和(b)的结果，采用公式

Figure 2011100947597100002DEST_PATH_IMAGE044
，即延时归一化自相关公式，计算时间同步粗同步点
Figure 2011100947597100002DEST_PATH_IMAGE046
；(c) Based on the results of steps (a) and (b), apply the formula
Figure 2011100947597100002DEST_PATH_IMAGE044
, which is the delay normalized autocorrelation formula, and calculates the time synchronization coarse synchronization point
Figure 2011100947597100002DEST_PATH_IMAGE046
;

(d) 确定时间同步粗同步点

Figure 179948DEST_PATH_IMAGE046
后，根据步骤(b)中的滑动步长确定时间同步精同步范围
Figure 2011100947597100002DEST_PATH_IMAGE048
，即由时间同步粗同步位置向前推移16个采样点和向后推移16个采样点(或32个采样点)，并恢复步骤(a)的原始采样速率，根据公式和公式
Figure 2011100947597100002DEST_PATH_IMAGE052
计算时，
Figure 2011100947597100002DEST_PATH_IMAGE056
和
Figure 2011100947597100002DEST_PATH_IMAGE058
的值并保存，作为采用递归的方法计算下一个值的初始值。为未经降采样基带数字信号，
Figure 329583DEST_PATH_IMAGE056
为当前
Figure 58504DEST_PATH_IMAGE022
值
Figure 137319DEST_PATH_IMAGE026
个相距为
Figure 635296DEST_PATH_IMAGE028
的样值对之间相关值之和，
Figure 406943DEST_PATH_IMAGE058
为当前
Figure 419899DEST_PATH_IMAGE022
值独立于频率偏差的能量项；(d) Determining the time synchronization coarse synchronization point
Figure 179948DEST_PATH_IMAGE046
Finally, determine the time synchronization fine synchronization range according to the sliding step in step (b)
Figure 2011100947597100002DEST_PATH_IMAGE048
, that is, move forward 16 sampling points and backward 16 sampling points (or 32 sampling points) from the time synchronization coarse synchronization position, and restore the original sampling rate of step (a), according to the formula and the formula
Figure 2011100947597100002DEST_PATH_IMAGE052
calculate hour,
Figure 2011100947597100002DEST_PATH_IMAGE056
and
Figure 2011100947597100002DEST_PATH_IMAGE058
and save the value as a recursive method to calculate the next The initial value of the value. is a non-downsampled baseband digital signal,
Figure 329583DEST_PATH_IMAGE056
for the current
Figure 58504DEST_PATH_IMAGE022
value
Figure 137319DEST_PATH_IMAGE026
a distance of
Figure 635296DEST_PATH_IMAGE028
The sum of the correlation values between the sample value pairs,
Figure 406943DEST_PATH_IMAGE058
for the current
Figure 419899DEST_PATH_IMAGE022
an energy term whose value is independent of the frequency deviation;

(e) 以滑动步长为1，采用和步骤(b)相同的递归方法计算的相邻的下一个

Figure 302404DEST_PATH_IMAGE022
值(也就是
Figure 2011100947597100002DEST_PATH_IMAGE060
)对应的
Figure 2011100947597100002DEST_PATH_IMAGE062
和
Figure 2011100947597100002DEST_PATH_IMAGE064
的值。(e) With the sliding step as 1, the next adjacent one calculated by the same recursive method as in step (b)
Figure 302404DEST_PATH_IMAGE022
value (i.e.
Figure 2011100947597100002DEST_PATH_IMAGE060
)corresponding
Figure 2011100947597100002DEST_PATH_IMAGE062
and
Figure 2011100947597100002DEST_PATH_IMAGE064
value.

(f) 根据步骤(e)的结果，在时间同步精同步范围内采用公式

Figure 2011100947597100002DEST_PATH_IMAGE066
，
Figure 2011100947597100002DEST_PATH_IMAGE068
计算ML同步点
Figure 2011100947597100002DEST_PATH_IMAGE070
和小数倍频偏。(f) Based on the results of step (e), within the scope of time synchronization fine synchronization using the formula
Figure 2011100947597100002DEST_PATH_IMAGE066
,
Figure 2011100947597100002DEST_PATH_IMAGE068
Calculate ML synchronization points
Figure 2011100947597100002DEST_PATH_IMAGE070
and fractional frequency offset .

进一步，所述步骤(2)之前，还包括如下步骤：Further, before the step (2), the following steps are also included:

(g) 根据小区组内ID号

Figure 2011100947597100002DEST_PATH_IMAGE074
生成本地主同步频域信号，经过IFFT转换到时域，用
Figure 2011100947597100002DEST_PATH_IMAGE076
表示，其中
Figure 2011100947597100002DEST_PATH_IMAGE078
表示与
Figure 530254DEST_PATH_IMAGE074
一一对应的ZC根索引，N为OFDM符号抽样点数；(g) According to the ID number in the cell group
Figure 2011100947597100002DEST_PATH_IMAGE074
Generate a local master synchronous frequency domain signal, convert it to the time domain through IFFT, and use
Figure 2011100947597100002DEST_PATH_IMAGE076
said, among them
Figure 2011100947597100002DEST_PATH_IMAGE078
express with
Figure 530254DEST_PATH_IMAGE074
One-to-one corresponding ZC root index, N is the number of OFDM symbol sampling points;

(h) 采用互相关公式

Figure 2011100947597100002DEST_PATH_IMAGE080
计算接收端接收到的基带信号与本地生成的时域PSS信号的互相关；(h) Using the cross-correlation formula
Figure 2011100947597100002DEST_PATH_IMAGE080
Calculate the cross-correlation between the baseband signal received by the receiving end and the locally generated time-domain PSS signal;

(i) 根据公式

Figure 2011100947597100002DEST_PATH_IMAGE082
的峰值点计算精度更高的时间同步点
Figure 2011100947597100002DEST_PATH_IMAGE084
；(i) According to the formula
Figure 2011100947597100002DEST_PATH_IMAGE082
The peak point calculation accuracy of the time synchronization point is higher
Figure 2011100947597100002DEST_PATH_IMAGE084
;

(j) 根据时间同步位置，计算接收端接收到基带信号主同步信号所在OFDM符号的起始位置，去掉OFDM符号前的CP。采用公式

Figure 2011100947597100002DEST_PATH_IMAGE086
计算接收端带小数倍频偏的基带信号与本地生成时域主同步信号的互相关；其中
Figure 2011100947597100002DEST_PATH_IMAGE088
为接收端带小数倍频偏的基带信号，
Figure 2011100947597100002DEST_PATH_IMAGE090
为发送端主同步信号，
Figure 2011100947597100002DEST_PATH_IMAGE092
为频率偏移，
Figure 2011100947597100002DEST_PATH_IMAGE094
为高斯噪声。(j) According to the time synchronization position, calculate the starting position of the OFDM symbol where the main synchronization signal of the baseband signal is received by the receiving end, and remove the CP before the OFDM symbol. use the formula
Figure 2011100947597100002DEST_PATH_IMAGE086
Calculate the cross-correlation between the baseband signal with a fractional frequency offset at the receiving end and the locally generated time-domain primary synchronization signal; where
Figure 2011100947597100002DEST_PATH_IMAGE088
is the baseband signal with fractional frequency offset at the receiving end,
Figure 2011100947597100002DEST_PATH_IMAGE090
is the main synchronization signal at the sending end,
Figure 2011100947597100002DEST_PATH_IMAGE092
is the frequency offset,
Figure 2011100947597100002DEST_PATH_IMAGE094
is Gaussian noise.

(k) 基于最大似然公式

Figure 2011100947597100002DEST_PATH_IMAGE096
峰值点计算精确更高的小数倍频偏，其中
Figure 2011100947597100002DEST_PATH_IMAGE098
为
Figure 348168DEST_PATH_IMAGE092
的最大似然估计值。采用公式
Figure 2011100947597100002DEST_PATH_IMAGE100
得到归一化小数倍频偏，在TD-LTE系统中规定为采样时间间隔。(k) Based on maximum likelihood formula
Figure 2011100947597100002DEST_PATH_IMAGE096
The peak point calculation is more accurate and higher fractional frequency deviation, where
Figure 2011100947597100002DEST_PATH_IMAGE098
for
Figure 348168DEST_PATH_IMAGE092
The maximum likelihood estimate of . use the formula
Figure 2011100947597100002DEST_PATH_IMAGE100
Get the normalized fractional multiple frequency offset, which is specified in the TD-LTE system is the sampling time interval.

进一步，所述步骤(k)中，根据似然函数，假设频率偏移值范围为，以步长

Figure 2011100947597100002DEST_PATH_IMAGE106
为间隔取样：
Figure 2011100947597100002DEST_PATH_IMAGE108
；将所得采样点的数值代入步骤(k)中的最大似然公式得到相应的似然函数值，其最大似然函数值所对应的样值点的就是频率偏移的最大似然估计值
Figure 599152DEST_PATH_IMAGE098
，其中
Figure 2011100947597100002DEST_PATH_IMAGE110
为最大频偏。Further, in the step (k), according to the likelihood function, it is assumed that the frequency offset value range is , with the step size
Figure 2011100947597100002DEST_PATH_IMAGE106
Sampling for an interval:
Figure 2011100947597100002DEST_PATH_IMAGE108
; The numerical value of gained sampling point is substituted into the maximum likelihood formula in the step (k) to obtain the corresponding likelihood function value, and what the sample value point corresponding to its maximum likelihood function value is exactly the maximum likelihood estimated value of frequency offset
Figure 599152DEST_PATH_IMAGE098
,in
Figure 2011100947597100002DEST_PATH_IMAGE110
is the maximum frequency deviation.

进一步，所述步骤(3)之前，还包括如下步骤：Further, before the step (3), the following steps are also included:

(l) 由步骤(j)确定接收端基带信号主同步信号所在OFDM符号的起始位置后，根据主同步信号映射到直流子载波附近的62个子载波上，确定接收端主同步信号的位置,则当前PSS信号占用子载波的频域信道估计值为：(l) After determining the initial position of the OFDM symbol where the main synchronization signal of the baseband signal at the receiving end is located in step (j), map the main synchronization signal to 62 subcarriers near the DC subcarrier to determine the position of the main synchronization signal at the receiving end, Then the frequency-domain channel estimation value of the subcarrier occupied by the current PSS signal is:

，其中

Figure 2011100947597100002DEST_PATH_IMAGE114
为接收端频域PSS信号，为本地生成的频域PSS信号，对
Figure 2011100947597100002DEST_PATH_IMAGE118
进行N=2048点的IFFT，得到时域信道冲激响应：
Figure 2011100947597100002DEST_PATH_IMAGE120
； ,in
Figure 2011100947597100002DEST_PATH_IMAGE114
is the frequency-domain PSS signal at the receiver, is a locally generated frequency-domain PSS signal, for
Figure 2011100947597100002DEST_PATH_IMAGE118
Perform IFFT with N= 2048 points to get the channel impulse response in time domain:
Figure 2011100947597100002DEST_PATH_IMAGE120
;

(m) 根据步骤（l）得到的时域信道冲激响应

Figure 2011100947597100002DEST_PATH_IMAGE122
，估计峰值点位置
Figure 2011100947597100002DEST_PATH_IMAGE124
：(m) The time-domain channel impulse response obtained according to step (l)
Figure 2011100947597100002DEST_PATH_IMAGE122
, to estimate the position of the peak point
Figure 2011100947597100002DEST_PATH_IMAGE124
:

Figure 2011100947597100002DEST_PATH_IMAGE126
，并估计最大功率，定义阈值Th：
Figure 2011100947597100002DEST_PATH_IMAGE128
，其中
Figure 2011100947597100002DEST_PATH_IMAGE130
为阈值系数，
Figure 2011100947597100002DEST_PATH_IMAGE132
；
Figure 2011100947597100002DEST_PATH_IMAGE126
, and to estimate the maximum power, define the threshold Th :
Figure 2011100947597100002DEST_PATH_IMAGE128
,in
Figure 2011100947597100002DEST_PATH_IMAGE130
is the threshold coefficient,
Figure 2011100947597100002DEST_PATH_IMAGE132
;

(n) 对于

Figure 2011100947597100002DEST_PATH_IMAGE134
，从n=1开始按照n的递增顺序，检测得到第1个大于阈值Th的瞬时功率，记其位置为L start ，从n=N cp开始按照n的递减顺序，检测得到第1个大于阈值Th的瞬时功率，记其位置为L end ，则时域信道冲激响应长度为：；(n) for
Figure 2011100947597100002DEST_PATH_IMAGE134
, starting from n = 1, according to the increasing order of n , the first instantaneous power greater than the threshold Th is detected, and its position is recorded as L start , starting from n = N cp , according to the decreasing order of n , the first detected instantaneous power is greater than the threshold The instantaneous power of Th is recorded as its position as L end , then the impulse response length of the time-domain channel for: ;

(o) 采用公式，

Figure 2011100947597100002DEST_PATH_IMAGE142
将分别用ML算法自相关和用时域PSS互相关计算的时间同步点和小数倍频偏加权平均，得到TD-LTE系统的时间同步和小数倍频率同步，其中
Figure 2011100947597100002DEST_PATH_IMAGE144
为用简化后ML算法估计的时间同步点和小数倍频偏在加权平均中的比例因子，
Figure 2011100947597100002DEST_PATH_IMAGE146
为用PSS估计的时间同步点和小数倍频偏在加权平均中的比例因子，
Figure 767572DEST_PATH_IMAGE144
的大小由冲激响应长度与CP长度的大小关系确定，
Figure 2011100947597100002DEST_PATH_IMAGE148
。根据仿真，当
Figure 2011100947597100002DEST_PATH_IMAGE150
时，
Figure 353591DEST_PATH_IMAGE144
=1，当
Figure 2011100947597100002DEST_PATH_IMAGE152
时，
Figure 341139DEST_PATH_IMAGE144
=0.6，当
Figure 2011100947597100002DEST_PATH_IMAGE154
时，
Figure 565447DEST_PATH_IMAGE144
=0.2，当
Figure 2011100947597100002DEST_PATH_IMAGE156
时，=0。(o) Using the formula ,
Figure 2011100947597100002DEST_PATH_IMAGE142
The time synchronization point and fractional frequency offset calculated by ML algorithm autocorrelation and time-domain PSS cross-correlation are weighted average to obtain the time synchronization and fractional frequency synchronization of the TD-LTE system, where
Figure 2011100947597100002DEST_PATH_IMAGE144
is the scale factor of the time synchronization point and fractional frequency offset estimated by the simplified ML algorithm in the weighted average,
Figure 2011100947597100002DEST_PATH_IMAGE146
is the scaling factor of the time synchronization point and fractional frequency offset estimated by PSS in the weighted average,
Figure 767572DEST_PATH_IMAGE144
The magnitude of the impulse response length The size relationship with the CP length is determined,
Figure 2011100947597100002DEST_PATH_IMAGE148
. According to the simulation, when
Figure 2011100947597100002DEST_PATH_IMAGE150
hour,
Figure 353591DEST_PATH_IMAGE144
=1, when
Figure 2011100947597100002DEST_PATH_IMAGE152
hour,
Figure 341139DEST_PATH_IMAGE144
=0.6, when
Figure 2011100947597100002DEST_PATH_IMAGE154
hour,
Figure 565447DEST_PATH_IMAGE144
=0.2, when
Figure 2011100947597100002DEST_PATH_IMAGE156
hour, =0.

本发明利用主同步信号获取信道冲激响应，结合基于简化后的ML和主同步信号的时域同步算法，自适应调整同步估计算法参数，为TD-LTE系统提出了一种有效且简单的时间同步和小数倍频率同步的方法。与传统的分别用ML算法或主同步信号计算时频同步相比，本发明简化了时频同步的过程，大大降低了处理时间，提高了处理效率。The present invention uses the main synchronization signal to obtain the channel impulse response, combines the time domain synchronization algorithm based on the simplified ML and the main synchronization signal, and adaptively adjusts the parameters of the synchronization estimation algorithm, and proposes an effective and simple time synchronization algorithm for the TD-LTE system. Methods for synchronization and fractional frequency synchronization. Compared with the traditional calculation of time-frequency synchronization using ML algorithm or main synchronization signal respectively, the present invention simplifies the process of time-frequency synchronization, greatly reduces processing time and improves processing efficiency.

附图说明Description of drawings

图1 TD-LTE系统中帧结构类型2的帧结构示意图；Figure 1 Schematic diagram of the frame structure of frame structure type 2 in the TD-LTE system;

图2 TD-LTE系统中帧结构类型2中PSS所在频率位置的示意图；Figure 2 is a schematic diagram of the frequency position of the PSS in frame structure type 2 in the TD-LTE system;

图3 TD-LTE系统采用加权平均求时间同步和小数倍频率同步方法的流程图。Figure 3 TD-LTE system adopts the flow chart of weighted average time synchronization and fractional multiple frequency synchronization methods.

具体实施方式Detailed ways

本发明公开了一种用于TD-LTE系统时间同步和小数倍频率同步的方法，根据实际的信道环境自适应的用简化的ML算法和PSS分别得到的时间同步点和小数倍频偏进行加权平均，参考图3，包括如下步骤：The invention discloses a method for TD-LTE system time synchronization and fractional multiple frequency synchronization, according to the actual channel environment, the time synchronization point and fractional multiple frequency offset obtained respectively by using simplified ML algorithm and PSS Carry out weighted average, refer to Fig. 3, comprise the following steps:

(101) 将接收端接收到的基带信号采用降采样(可采用1/2、1/4或1/8降采样)，即每2、4、8个符号抽样一个符号。以下以1/4降采样为例进行说明。计算公式

Figure 724344DEST_PATH_IMAGE012
和公式(101) The baseband signal received by the receiving end is down-sampled (1/2, 1/4 or 1/8 down-sampling can be used), that is, one symbol is sampled every 2, 4, or 8 symbols. The following takes 1/4 downsampling as an example for illustration. Calculation formula
Figure 724344DEST_PATH_IMAGE012
and the formula

Figure 136871DEST_PATH_IMAGE014
在
Figure 164870DEST_PATH_IMAGE016
时，
Figure 346452DEST_PATH_IMAGE018
和
Figure 536125DEST_PATH_IMAGE020
的值并保存，作为采用递归的方法计算下一个抽样符号
Figure 435948DEST_PATH_IMAGE022
值的初始值。其中，为时域序号，
Figure 366044DEST_PATH_IMAGE026
为CP的长度，
Figure 726618DEST_PATH_IMAGE028
为一个OFDM符号的抽样点数，
Figure 113737DEST_PATH_IMAGE030
为经过1/4降采样后的基带数字信号，
Figure 483538DEST_PATH_IMAGE018
为当前值
Figure 984238DEST_PATH_IMAGE032
个相距为
Figure 593074DEST_PATH_IMAGE028
的样值对之间相关值之和，
Figure 766566DEST_PATH_IMAGE020
为当前
Figure 777248DEST_PATH_IMAGE022
值为0独立于频率偏差的能量项；
Figure 136871DEST_PATH_IMAGE014
exist
Figure 164870DEST_PATH_IMAGE016
hour,
Figure 346452DEST_PATH_IMAGE018
and
Figure 536125DEST_PATH_IMAGE020
and save the value as the recursive method to calculate the next sampling symbol
Figure 435948DEST_PATH_IMAGE022
The initial value of the value. in, is the sequence number in the time domain,
Figure 366044DEST_PATH_IMAGE026
is the length of CP,
Figure 726618DEST_PATH_IMAGE028
is the number of sampling points for one OFDM symbol,
Figure 113737DEST_PATH_IMAGE030
is the baseband digital signal after 1/4 downsampling,
Figure 483538DEST_PATH_IMAGE018
for the current value
Figure 984238DEST_PATH_IMAGE032
a distance of
Figure 593074DEST_PATH_IMAGE028
The sum of the correlation values between the sample value pairs,
Figure 766566DEST_PATH_IMAGE020
for the current
Figure 777248DEST_PATH_IMAGE022
An energy term whose value is 0 is independent of the frequency deviation;

(102) 由终端运算量要求确定滑动步长，如以滑动步长为16(或32)，采用递归的方法计算的下一个值(也就是

Figure 903652DEST_PATH_IMAGE034
)对应的和
Figure 480444DEST_PATH_IMAGE038
的值，即调用以下公式计算：(102) The sliding step size is determined by the terminal operation requirements. For example, the sliding step size is 16 (or 32), and the next step calculated by recursive method value (i.e.
Figure 903652DEST_PATH_IMAGE034
)corresponding and
Figure 480444DEST_PATH_IMAGE038
The value of , which is calculated by calling the following formula:

Figure 619302DEST_PATH_IMAGE040
和
Figure 619302DEST_PATH_IMAGE040
and

Figure 202730DEST_PATH_IMAGE042
，其中，
Figure 262565DEST_PATH_IMAGE036
，分别为下一个
Figure 557597DEST_PATH_IMAGE022
值的相关值之和和能量项，则计算每个
Figure 628321DEST_PATH_IMAGE022
值的只需要2次复数相乘和2次复数相加，计算每个
Figure 583825DEST_PATH_IMAGE022
值的只需要4次复数相乘和4次复数相加，可以得到经过简化后的方法大大的减小了ML算法的计算复杂度；
Figure 202730DEST_PATH_IMAGE042
,in,
Figure 262565DEST_PATH_IMAGE036
, respectively for the next
Figure 557597DEST_PATH_IMAGE022
The sum of the associated values of the values and the energy term are calculated for each
Figure 628321DEST_PATH_IMAGE022
worth it Only 2 complex multiplications and 2 complex additions are required to calculate each
Figure 583825DEST_PATH_IMAGE022
worth it Only 4 complex multiplications and 4 complex additions are required, and the simplified method can greatly reduce the computational complexity of the ML algorithm;

(103) 根据步骤(101)和(102)的结果，采用公式

Figure 356926DEST_PATH_IMAGE044
，即延时归一化自相关公式，计算时间同步粗同步点
Figure 214024DEST_PATH_IMAGE046
；(103) According to the results of steps (101) and (102), the formula
Figure 356926DEST_PATH_IMAGE044
, which is the delay normalized autocorrelation formula, and calculates the time synchronization coarse synchronization point
Figure 214024DEST_PATH_IMAGE046
;

(104) 确定时间同步粗同步点

Figure 908310DEST_PATH_IMAGE046
后，根据步骤(102)中的滑动步长确定时间同步精同步范围
Figure 559871DEST_PATH_IMAGE048
，即由时间同步粗同步位置向前推移16个采样点和向后推移16个采样点(或32个采样点)，并恢复步骤(101)的原始采样速率，根据公式
Figure 418237DEST_PATH_IMAGE050
和公式
Figure 813446DEST_PATH_IMAGE052
计算
Figure 362239DEST_PATH_IMAGE054
时，
Figure 450281DEST_PATH_IMAGE056
和
Figure 717314DEST_PATH_IMAGE058
的值并保存，作为采用递归的方法计算下一个
Figure 978531DEST_PATH_IMAGE022
值的初始值。为未经降采样基带数字信号，
Figure 640774DEST_PATH_IMAGE056
为当前
Figure 395103DEST_PATH_IMAGE022
值
Figure 132115DEST_PATH_IMAGE026
个相距为
Figure 655500DEST_PATH_IMAGE028
的样值对之间相关值之和，
Figure 898394DEST_PATH_IMAGE058
为当前
Figure 874440DEST_PATH_IMAGE022
值独立于频率偏差的能量项；(104) Determine time synchronization coarse synchronization point
Figure 908310DEST_PATH_IMAGE046
Finally, determine the time synchronization fine synchronization range according to the sliding step in step (102)
Figure 559871DEST_PATH_IMAGE048
, that is, move forward 16 sampling points and backward 16 sampling points (or 32 sampling points) from the time synchronization coarse synchronization position, and restore the original sampling rate of step (101), according to the formula
Figure 418237DEST_PATH_IMAGE050
and the formula
Figure 813446DEST_PATH_IMAGE052
calculate
Figure 362239DEST_PATH_IMAGE054
hour,
Figure 450281DEST_PATH_IMAGE056
and
Figure 717314DEST_PATH_IMAGE058
and save the value as a recursive method to calculate the next
Figure 978531DEST_PATH_IMAGE022
The initial value of the value. is a non-downsampled baseband digital signal,
Figure 640774DEST_PATH_IMAGE056
for the current
Figure 395103DEST_PATH_IMAGE022
value
Figure 132115DEST_PATH_IMAGE026
a distance of
Figure 655500DEST_PATH_IMAGE028
The sum of the correlation values between the sample value pairs,
Figure 898394DEST_PATH_IMAGE058
for the current
Figure 874440DEST_PATH_IMAGE022
an energy term whose value is independent of the frequency deviation;

(105) 以滑动步长为1，采用和步骤(102)相同的递归方法计算的相邻的下一个

Figure 415143DEST_PATH_IMAGE022
值(也就是
Figure 58614DEST_PATH_IMAGE060
)对应的
Figure 393780DEST_PATH_IMAGE062
和
Figure 185019DEST_PATH_IMAGE064
的值；(105) With the sliding step size being 1, the adjacent next
Figure 415143DEST_PATH_IMAGE022
value (i.e.
Figure 58614DEST_PATH_IMAGE060
)corresponding
Figure 393780DEST_PATH_IMAGE062
and
Figure 185019DEST_PATH_IMAGE064
value;

(106) 根据步骤(105)的结果，在时间同步精同步范围内采用公式

Figure 263833DEST_PATH_IMAGE066
，
Figure 761811DEST_PATH_IMAGE068
计算ML同步点
Figure 267878DEST_PATH_IMAGE070
和小数倍频偏
Figure 484096DEST_PATH_IMAGE072
；(106) According to the result of step (105), the formula
Figure 263833DEST_PATH_IMAGE066
,
Figure 761811DEST_PATH_IMAGE068
Calculate ML synchronization points
Figure 267878DEST_PATH_IMAGE070
and fractional frequency offset
Figure 484096DEST_PATH_IMAGE072
;

(107) 根据小区组内ID号生成本地主同步频域信号，经过IFFT转换到时域，用表示，其中

Figure 474683DEST_PATH_IMAGE078
表示与
Figure 912618DEST_PATH_IMAGE074
一一对应的ZC根索引，N为OFDM符号抽样点数；(107) According to the ID number in the cell group Generate a local master synchronous frequency domain signal, convert it to the time domain through IFFT, and use said, among them
Figure 474683DEST_PATH_IMAGE078
express with
Figure 912618DEST_PATH_IMAGE074
One-to-one corresponding ZC root index, N is the number of OFDM symbol sampling points;

(108) 采用互相关公式

Figure 333235DEST_PATH_IMAGE080
计算接收端接收到的基带信号与本地生成的时域PSS信号的互相关；(108) Using the cross-correlation formula
Figure 333235DEST_PATH_IMAGE080
Calculate the cross-correlation between the baseband signal received by the receiving end and the locally generated time-domain PSS signal;

(109) 根据公式

Figure 805804DEST_PATH_IMAGE082
的峰值点计算精度更高的时间同步点
Figure 715991DEST_PATH_IMAGE084
；(109) According to the formula
Figure 805804DEST_PATH_IMAGE082
The peak point calculation accuracy of the time synchronization point is higher
Figure 715991DEST_PATH_IMAGE084
;

(110) 根据时间同步位置，计算接收端接收到基带信号主同步信号所在OFDM符号的起始位置，去掉OFDM符号前的CP。采用公式

Figure 641222DEST_PATH_IMAGE086
计算接收端带小数倍频偏的基带信号与本地生成时域主同步信号的互相关；其中
Figure 865530DEST_PATH_IMAGE088
为接收端带小数倍频偏的基带信号，
Figure 192606DEST_PATH_IMAGE090
为发送端主同步信号，
Figure 476957DEST_PATH_IMAGE092
为频率偏移，
Figure 699603DEST_PATH_IMAGE094
为高斯噪声。(110) According to the time synchronization position, calculate the starting position of the OFDM symbol where the main synchronization signal of the baseband signal is received by the receiving end, and remove the CP before the OFDM symbol. use the formula
Figure 641222DEST_PATH_IMAGE086
Calculate the cross-correlation between the baseband signal with a fractional frequency offset at the receiving end and the locally generated time-domain primary synchronization signal; where
Figure 865530DEST_PATH_IMAGE088
is the baseband signal with fractional frequency offset at the receiving end,
Figure 192606DEST_PATH_IMAGE090
is the main synchronization signal at the sending end,
Figure 476957DEST_PATH_IMAGE092
is the frequency offset,
Figure 699603DEST_PATH_IMAGE094
is Gaussian noise.

(111) 基于最大似然公式

Figure 462023DEST_PATH_IMAGE096
峰值点计算精确更高的小数倍频偏，其中
Figure 643606DEST_PATH_IMAGE098
为
Figure 98858DEST_PATH_IMAGE092
的最大似然估计值。采用公式
Figure 998681DEST_PATH_IMAGE100
得到归一化小数倍频偏，在TD-LTE系统中规定
Figure 627108DEST_PATH_IMAGE102
为采样时间间隔；(111) Based on the maximum likelihood formula
Figure 462023DEST_PATH_IMAGE096
The peak point calculation is more accurate and higher fractional frequency deviation, where
Figure 643606DEST_PATH_IMAGE098
for
Figure 98858DEST_PATH_IMAGE092
The maximum likelihood estimate of . use the formula
Figure 998681DEST_PATH_IMAGE100
Get the normalized fractional multiple frequency offset, which is specified in the TD-LTE system
Figure 627108DEST_PATH_IMAGE102
is the sampling time interval;

(112) 由步骤(110)确定接收端基带信号主同步信号所在OFDM符号的起始位置后，根据主同步信号映射到直流子载波附近的62个子载波上，确定接收端主同步信号的位置,则当前PSS信号占用子载波的频域信道估计值为：(112) After determining the initial position of the OFDM symbol where the main synchronization signal of the baseband signal at the receiving end is determined by step (110), map the main synchronization signal to 62 subcarriers near the DC subcarrier to determine the position of the main synchronization signal at the receiving end, Then the frequency-domain channel estimation value of the subcarrier occupied by the current PSS signal is:

Figure 663197DEST_PATH_IMAGE112
，其中
Figure 289351DEST_PATH_IMAGE114
为接收端频域PSS信号，
Figure 676470DEST_PATH_IMAGE116
为本地生成的频域PSS信号，对
Figure 780692DEST_PATH_IMAGE118
进行N=2048点的IFFT，得到时域信道冲激响应：
Figure 749916DEST_PATH_IMAGE120
；
Figure 663197DEST_PATH_IMAGE112
,in
Figure 289351DEST_PATH_IMAGE114
is the frequency-domain PSS signal at the receiver,
Figure 676470DEST_PATH_IMAGE116
is a locally generated frequency-domain PSS signal, for
Figure 780692DEST_PATH_IMAGE118
Perform IFFT with N= 2048 points to get the channel impulse response in time domain:
Figure 749916DEST_PATH_IMAGE120
;

(113) 根据步骤（112）得到的时域信道冲激响应

Figure 546971DEST_PATH_IMAGE122
，估计峰值点位置
Figure 155807DEST_PATH_IMAGE124
：
Figure 329299DEST_PATH_IMAGE126
，并估计最大功率，定义阈值Th：，其中
Figure 370253DEST_PATH_IMAGE130
为阈值系数，
Figure 466385DEST_PATH_IMAGE132
；(113) The time-domain channel impulse response obtained according to step (112)
Figure 546971DEST_PATH_IMAGE122
, to estimate the position of the peak point
Figure 155807DEST_PATH_IMAGE124
:
Figure 329299DEST_PATH_IMAGE126
, and to estimate the maximum power, define the threshold Th : ,in
Figure 370253DEST_PATH_IMAGE130
is the threshold coefficient,
Figure 466385DEST_PATH_IMAGE132
;

(114) 对于，从n=1开始按照n的递增顺序，检测得到第1个大于阈值Th的瞬时功率，记其位置为L start ，从n=N cp开始按照n的递减顺序，检测得到第1个大于阈值Th的瞬时功率，记其位置为L end ，则时域信道冲激响应长度

Figure 43177DEST_PATH_IMAGE136
为：
Figure 182034DEST_PATH_IMAGE138
；(114) for , starting from n = 1, according to the increasing order of n , the first instantaneous power greater than the threshold Th is detected, and its position is recorded as L start , starting from n = N cp , according to the decreasing order of n , the first detected instantaneous power is greater than the threshold The instantaneous power of Th is recorded as its position as L end , then the impulse response length of the time-domain channel
Figure 43177DEST_PATH_IMAGE136
for:
Figure 182034DEST_PATH_IMAGE138
;

(115) 采用公式

Figure 578512DEST_PATH_IMAGE140
，
Figure 828228DEST_PATH_IMAGE142
将分别用ML算法自相关和用时域PSS互相关计算的时间同步点和小数倍频偏加权平均，得到TD-LTE系统的时间同步和小数倍频率同步，其中
Figure 813501DEST_PATH_IMAGE144
为用简化后ML算法估计的时间同步点和小数倍频偏在加权平均中的比例因子，
Figure 123260DEST_PATH_IMAGE146
为用PSS估计的时间同步点和小数倍频偏在加权平均中的比例因子，
Figure 193984DEST_PATH_IMAGE144
的大小由冲激响应长度
Figure 247391DEST_PATH_IMAGE136
与CP长度的大小关系确定，
Figure 149487DEST_PATH_IMAGE148
。根据仿真，当
Figure 630147DEST_PATH_IMAGE150
时，
Figure 922588DEST_PATH_IMAGE144
=1，当
Figure 779686DEST_PATH_IMAGE152
时，=0.6，当
Figure 938583DEST_PATH_IMAGE154
时，
Figure 983899DEST_PATH_IMAGE144
=0.2，当时，=0。(115) using the formula
Figure 578512DEST_PATH_IMAGE140
,
Figure 828228DEST_PATH_IMAGE142
The time synchronization point and fractional frequency offset calculated by ML algorithm autocorrelation and time-domain PSS cross-correlation are weighted average to obtain the time synchronization and fractional frequency synchronization of the TD-LTE system, where
Figure 813501DEST_PATH_IMAGE144
is the scale factor of the time synchronization point and fractional frequency offset estimated by the simplified ML algorithm in the weighted average,
Figure 123260DEST_PATH_IMAGE146
is the scaling factor of the time synchronization point and fractional frequency offset estimated by PSS in the weighted average,
Figure 193984DEST_PATH_IMAGE144
The magnitude of the impulse response length
Figure 247391DEST_PATH_IMAGE136
The size relationship with the CP length is determined,
Figure 149487DEST_PATH_IMAGE148
. According to the simulation, when
Figure 630147DEST_PATH_IMAGE150
hour,
Figure 922588DEST_PATH_IMAGE144
=1, when
Figure 779686DEST_PATH_IMAGE152
hour, =0.6, when
Figure 938583DEST_PATH_IMAGE154
hour,
Figure 983899DEST_PATH_IMAGE144
=0.2, when hour, =0.

(116) 根据步骤(115)得到的小数倍频偏估计对接收端的基带数字信号进行小数倍频偏校正。(116) Perform fractional frequency offset correction on the baseband digital signal at the receiving end according to the fractional frequency offset estimation obtained in step (115).

Claims (4)

1. A method for time synchronization and decimal frequency synchronization of a TD-LTE system is characterized by comprising the following steps:
(1) the receiving end carries out frequency reduction processing on the baseband digital signal, delays one OFDM symbol sampling point number, carries out normalization autocorrelation in a time domain to generate a peak value, and determines a time synchronization point according to the peak value
Figure 2011100947597100001DEST_PATH_IMAGE002
Sum fractional frequency offset estimation
Figure 2011100947597100001DEST_PATH_IMAGE004
；
(2) The baseband digital signal and the locally generated main synchronous signal are cross-correlated in a time domain to generate a peak value, and a time synchronous point with higher accuracy is obtained according to the peak value
Figure 2011100947597100001DEST_PATH_IMAGE006
And fractional frequency offset
Figure 2011100947597100001DEST_PATH_IMAGE008
；
(3) Determining the position of the main synchronization signal of the receiving end, obtaining the channel estimation value in the time domain by adopting non-coherent detection according to the PSS signal, and determining the weighting coefficient of the ML algorithm according to the channel impulse response length
Figure 2011100947597100001DEST_PATH_IMAGE010
And PSS algorithm weighting coefficients
Figure 2011100947597100001DEST_PATH_IMAGE012
By the formula
Figure 2011100947597100001DEST_PATH_IMAGE014
，
Figure 2011100947597100001DEST_PATH_IMAGE016
Obtaining a time synchronization point of a TD-LTE system
Figure 2011100947597100001DEST_PATH_IMAGE018
And decimal frequency synchronization point
Figure 2011100947597100001DEST_PATH_IMAGE020
Wherein
Figure 2011100947597100001DEST_PATH_IMAGE022
。
2. the method of claim 1, wherein the method comprises the steps of: the frequency reduction treatment in the step (1) specifically comprises: adopting 1/4 to reduce sampling according to formula
Figure 2011100947597100001DEST_PATH_IMAGE024
And formula
Figure 2011100947597100001DEST_PATH_IMAGE026
Is determined atTime of flight
Figure 2011100947597100001DEST_PATH_IMAGE030
Are at a distance of
Figure 2011100947597100001DEST_PATH_IMAGE032
Sum of correlation values between pairs of samples
Figure 2011100947597100001DEST_PATH_IMAGE034
In the field of
Figure 665402DEST_PATH_IMAGE028
Time-independent energy term of frequency deviation
Figure 2011100947597100001DEST_PATH_IMAGE036
Wherein
Figure 2011100947597100001DEST_PATH_IMAGE038
is a time domain sequence number that is,
Figure 2011100947597100001DEST_PATH_IMAGE040
which is the length of the CP,
Figure 395591DEST_PATH_IMAGE032
for the number of samples of one OFDM symbol,
Figure 2011100947597100001DEST_PATH_IMAGE042
1/4 is the baseband digital signal after down sampling;
the method of claim 1, wherein the method comprises the steps of: using a formula
Figure 2011100947597100001DEST_PATH_IMAGE044
，
Figure 2011100947597100001DEST_PATH_IMAGE046
Computing ML synchronization pointsAnd fractional frequency offset
Figure 632855DEST_PATH_IMAGE004
Wherein
Figure 2011100947597100001DEST_PATH_IMAGE048
、
Figure 2011100947597100001DEST_PATH_IMAGE050
are respectively as
Figure 2011100947597100001DEST_PATH_IMAGE052
The sum of the correlation values between pairs of time samples and an energy term independent of the frequency deviation,
Figure 2011100947597100001DEST_PATH_IMAGE054
to representFunction(s)Independent variable corresponding to maximum value
Figure 2011100947597100001DEST_PATH_IMAGE058
The value of (a) is selected,as a maximum likelihood estimate
Figure 257347DEST_PATH_IMAGE002
The sum of the correlation values between corresponding pairs of samples,
Figure 2011100947597100001DEST_PATH_IMAGE062
taking the argument of complex number.
3. The method of claim 1, wherein the method comprises the steps of: based on the maximum likelihood formula
Figure 2011100947597100001DEST_PATH_IMAGE064
Calculating a more accurate fractional frequency offset at the peak point, wherein
Figure 2011100947597100001DEST_PATH_IMAGE066
In order to be able to shift the frequency,
Figure 2011100947597100001DEST_PATH_IMAGE068
is composed of
Figure 142126DEST_PATH_IMAGE066
The maximum likelihood estimate of (a) is,
Figure 409159DEST_PATH_IMAGE038
as the time domainThe serial number of the serial number,
Figure 608059DEST_PATH_IMAGE032
for the number of samples of one OFDM symbol,
Figure 2011100947597100001DEST_PATH_IMAGE070
in order to sample the time interval between the samples,
Figure 2011100947597100001DEST_PATH_IMAGE072
generating a cross-correlation value of a time domain master synchronization signal for a baseband signal with a decimal frequency offset of a receiving end and the local part;
the method of claim 1, wherein the method comprises the steps of: using a formulaCalculating the cross-correlation between the baseband signal with decimal frequency offset at the receiving end and the locally generated time domain master synchronization signal
Figure 89987DEST_PATH_IMAGE038
Is a time domain sequence number that is,
Figure 83351DEST_PATH_IMAGE032
for the number of samples of one OFDM symbol,in order to sample the time interval between the samples,
Figure 2011100947597100001DEST_PATH_IMAGE076
a baseband signal with a fractional frequency offset for the receiving end,is a master synchronization signal of a transmitting end,
Figure 637009DEST_PATH_IMAGE066
in order to be able to shift the frequency,
Figure 2011100947597100001DEST_PATH_IMAGE080
is gaussian noise.
4. The method of claim 2, wherein the method comprises the steps of: calculating the next by using a recursive method
Figure 2011100947597100001DEST_PATH_IMAGE082
The initial value of the value.
CN 201110094759 2011-04-15 2011-04-15 Time frequency synchronization method for time division long-term evolution (TD-LTE) system Expired - Fee Related CN102130883B (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
CN 201110094759 CN102130883B (en) 2011-04-15 2011-04-15 Time frequency synchronization method for time division long-term evolution (TD-LTE) system

Applications Claiming Priority (1)

Application Number Priority Date Filing Date Title
CN 201110094759 CN102130883B (en) 2011-04-15 2011-04-15 Time frequency synchronization method for time division long-term evolution (TD-LTE) system

Publications (2)

Publication Number Publication Date
CN102130883A true CN102130883A (en) 2011-07-20
CN102130883B CN102130883B (en) 2013-07-17

Family

ID=44268775

Family Applications (1)

Application Number Title Priority Date Filing Date
CN 201110094759 Expired - Fee Related CN102130883B (en) 2011-04-15 2011-04-15 Time frequency synchronization method for time division long-term evolution (TD-LTE) system

Country Status (1)

Country Link
CN (1) CN102130883B (en)

Cited By (21)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN102256347A (en) * 2011-07-26 2011-11-23 国网信息通信有限公司 Synchronization method and device for flexible subcarrier OFDM (Frequency Division Multiplexing) system
CN102694763A (en) * 2012-05-31 2012-09-26 重庆邮电大学 Method for assessing integer frequency offset of TD-LTE system
CN102857961A (en) * 2012-09-14 2013-01-02 中国人民解放军总参谋部第五十七研究所 Time difference measuring method for communication signals with frequency shift
CN103139126A (en) * 2011-12-02 2013-06-05 上海无线通信研究中心 Universal method for achieving transmit-receive synchronization of wireless communication testing platform
CN103297995A (en) * 2012-02-28 2013-09-11 联芯科技有限公司 Acquiring method and device for main synchronizing signal timing
CN103379082A (en) * 2012-04-25 2013-10-30 马维尔国际有限公司 Time and frequency synchronization method and time and frequency synchronization device in LTE communication system
CN103546414A (en) * 2012-07-16 2014-01-29 京信通信系统（广州）有限公司 Frequency deviation estimation method and device for LTE systems
CN103873421A (en) * 2014-03-31 2014-06-18 清华大学 Symbol synchronizing method of multi-path channel
CN105992334A (en) * 2015-02-16 2016-10-05 苏州简约纳电子有限公司 Method for acquiring downlink time synchronization based on LTE (long term evolution) system
CN106063369A (en) * 2015-02-13 2016-10-26 华为技术有限公司 Baseband processing unit and base station system
CN106656453A (en) * 2017-02-05 2017-05-10 苏州维特比信息技术有限公司 Synchronous device and method in narrowband wireless communication terminal
CN107124246A (en) * 2017-06-28 2017-09-01 福州智程信息科技有限公司 A kind of TD LTE uplink and downlink timeslots match detection method
CN107359953A (en) * 2017-06-30 2017-11-17 京信通信系统（中国）有限公司 A kind of TD LTE synchronous method and synchronizer
CN108282434A (en) * 2017-01-06 2018-07-13 上海创远仪器技术股份有限公司 A kind of detection method of LTE downlinks primary synchronization signal
CN108512570A (en) * 2018-03-29 2018-09-07 北京智联安科技有限公司 Signal synchronization acquiring method, system, computer equipment and readable storage medium storing program for executing
CN109274631A (en) * 2018-12-11 2019-01-25 北京无线电测量研究所 Data symbol synchronous method based on wildcard-filter style fractional time delay filter
CN109639616A (en) * 2018-12-26 2019-04-16 东南大学 The down time-frequency synchronization method of synchronizing sequence and OFDM cyclic prefix is used in combination
CN111030959A (en) * 2019-12-26 2020-04-17 江苏科大亨芯半导体技术有限公司 NBIOT frequency domain time frequency synchronization method
CN111614593A (en) * 2020-05-07 2020-09-01 重庆金美通信有限责任公司 Synchronization method for resisting single-tone interference of OFDM system
US11038608B2 (en) 2016-11-04 2021-06-15 Huawei Technologies Co., Ltd. Frequency synchronization method and slave clock
CN113162878A (en) * 2021-04-10 2021-07-23 天津德力仪器设备有限公司 PSS symbol synchronization method applied to negative signal-to-noise ratio 5GNR system

Families Citing this family (1)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN104717722B (en) * 2015-04-01 2018-05-15 东南大学 The small station eliminated in ultra dense set network based on PSS interference finds method

Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN101079857A (en) * 2006-05-25 2007-11-28 北京泰美世纪科技有限公司 A carrier residual frequency deviation tracking method based on OFDM system
CN101651650A (en) * 2009-09-15 2010-02-17 北京天碁科技有限公司 Synchronization and frequency deviation combining evaluating method and device

Patent Citations (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN101079857A (en) * 2006-05-25 2007-11-28 北京泰美世纪科技有限公司 A carrier residual frequency deviation tracking method based on OFDM system
CN101651650A (en) * 2009-09-15 2010-02-17 北京天碁科技有限公司 Synchronization and frequency deviation combining evaluating method and device

Cited By (36)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN102256347A (en) * 2011-07-26 2011-11-23 国网信息通信有限公司 Synchronization method and device for flexible subcarrier OFDM (Frequency Division Multiplexing) system
CN102256347B (en) * 2011-07-26 2015-09-30 国网信息通信有限公司 The synchronous method of flexible sub-carrier ofdm system and device
CN103139126A (en) * 2011-12-02 2013-06-05 上海无线通信研究中心 Universal method for achieving transmit-receive synchronization of wireless communication testing platform
WO2013079028A1 (en) * 2011-12-02 2013-06-06 上海无线通信研究中心 General method for implementing synchronous transmission and reception of wireless communication test platform
CN103297995B (en) * 2012-02-28 2016-01-27 联芯科技有限公司 The acquisition methods of master sync signal timing and device
CN103297995A (en) * 2012-02-28 2013-09-11 联芯科技有限公司 Acquiring method and device for main synchronizing signal timing
CN103379082B (en) * 2012-04-25 2018-04-10 马维尔国际有限公司 Time-frequency synchronization method and device in LTE communication system
CN103379082A (en) * 2012-04-25 2013-10-30 马维尔国际有限公司 Time and frequency synchronization method and time and frequency synchronization device in LTE communication system
CN102694763B (en) * 2012-05-31 2014-12-31 重庆邮电大学 Method for assessing integer frequency offset of TD-LTE system
CN102694763A (en) * 2012-05-31 2012-09-26 重庆邮电大学 Method for assessing integer frequency offset of TD-LTE system
CN103546414A (en) * 2012-07-16 2014-01-29 京信通信系统（广州）有限公司 Frequency deviation estimation method and device for LTE systems
CN102857961A (en) * 2012-09-14 2013-01-02 中国人民解放军总参谋部第五十七研究所 Time difference measuring method for communication signals with frequency shift
CN103873421B (en) * 2014-03-31 2017-04-19 清华大学 Symbol synchronizing method of multi-path channel
CN103873421A (en) * 2014-03-31 2014-06-18 清华大学 Symbol synchronizing method of multi-path channel
CN106063369A (en) * 2015-02-13 2016-10-26 华为技术有限公司 Baseband processing unit and base station system
CN106063369B (en) * 2015-02-13 2019-05-07 华为技术有限公司 Baseband processing unit and base station system
CN105992334A (en) * 2015-02-16 2016-10-05 苏州简约纳电子有限公司 Method for acquiring downlink time synchronization based on LTE (long term evolution) system
CN105992334B (en) * 2015-02-16 2019-07-12 苏州简约纳电子有限公司 A method of it is synchronous that downgoing time being obtained based on LTE system
US11038608B2 (en) 2016-11-04 2021-06-15 Huawei Technologies Co., Ltd. Frequency synchronization method and slave clock
CN108282434A (en) * 2017-01-06 2018-07-13 上海创远仪器技术股份有限公司 A kind of detection method of LTE downlinks primary synchronization signal
CN108282434B (en) * 2017-01-06 2021-07-09 上海创远仪器技术股份有限公司 Detection method of LTE downlink primary synchronization signal
CN106656453A (en) * 2017-02-05 2017-05-10 苏州维特比信息技术有限公司 Synchronous device and method in narrowband wireless communication terminal
CN106656453B (en) * 2017-02-05 2020-08-21 浙江芯科物联科技股份有限公司 Synchronization device in narrow-band wireless communication terminal
CN107124246A (en) * 2017-06-28 2017-09-01 福州智程信息科技有限公司 A kind of TD LTE uplink and downlink timeslots match detection method
CN107359953A (en) * 2017-06-30 2017-11-17 京信通信系统（中国）有限公司 A kind of TD LTE synchronous method and synchronizer
CN108512570B (en) * 2018-03-29 2020-10-27 北京智联安科技有限公司 Signal synchronization capturing method, system, computer device and readable storage medium
CN108512570A (en) * 2018-03-29 2018-09-07 北京智联安科技有限公司 Signal synchronization acquiring method, system, computer equipment and readable storage medium storing program for executing
CN109274631A (en) * 2018-12-11 2019-01-25 北京无线电测量研究所 Data symbol synchronous method based on wildcard-filter style fractional time delay filter
CN109274631B (en) * 2018-12-11 2021-04-23 北京无线电测量研究所 Data symbol synchronization method based on all-pass fractional delay filter
CN109639616B (en) * 2018-12-26 2021-04-06 东南大学 Downlink time-frequency synchronization method by jointly using synchronization sequence and OFDM cyclic prefix
CN109639616A (en) * 2018-12-26 2019-04-16 东南大学 The down time-frequency synchronization method of synchronizing sequence and OFDM cyclic prefix is used in combination
CN111030959B (en) * 2019-12-26 2021-05-11 江苏科大亨芯半导体技术有限公司 Frequency domain time-frequency synchronization method of NB-IoT
CN111030959A (en) * 2019-12-26 2020-04-17 江苏科大亨芯半导体技术有限公司 NBIOT frequency domain time frequency synchronization method
CN111614593A (en) * 2020-05-07 2020-09-01 重庆金美通信有限责任公司 Synchronization method for resisting single-tone interference of OFDM system
CN111614593B (en) * 2020-05-07 2022-10-14 重庆金美通信有限责任公司 Synchronization method for resisting single-tone interference of OFDM system
CN113162878A (en) * 2021-04-10 2021-07-23 天津德力仪器设备有限公司 PSS symbol synchronization method applied to negative signal-to-noise ratio 5GNR system

Also Published As

Publication number Publication date
CN102130883B (en) 2013-07-17

Similar Documents

Publication Publication Date Title
CN102130883B (en) Time frequency synchronization method for time division long-term evolution (TD-LTE) system
CN102694763B (en) Method for assessing integer frequency offset of TD-LTE system
US8576810B2 (en) Method and apparatus for detecting secondary synchronization signal
US7742392B2 (en) Blind carrier frequency offset estimator based on single-OFDM-symbol PN ranging code in multi-user OFDMA uplink
KR101291859B1 (en) Methods and systems for improved timing acquisition for varying channel conditions
US11812416B2 (en) Coherent detection of large physical random access control channel (PRACH) delays
CN101374131B (en) Method and apparatus of timing synchronization, leading symbol as well as method and apparatus for generating the same
CN109391403B (en) Method and apparatus for transmission and reception of wireless signals
CN101827052B (en) Method and device for time synchronization and frequency synchronization of LTE system
CN101815042B (en) Orthogonal frequency division multiplexing (OFDM) system channel estimation method and device
EP3008849A1 (en) Filterbank-based multicarrier transmitter for transmitting a multicarrier signal
CN102143101A (en) Mirror-extended frequency domain windowing orthogonal frequency division multiple access channel estimation method
US9961655B1 (en) Method and apparatus for low complexity frequency synchronization in LTE wireless communication systems
CN102647382B (en) Integer frequency offset estimation method and device
CN101552635B (en) Method and device for capturing frequency deviation
JP2010050885A (en) Wireless terminal, base station and channel characteristic estimating method
CN101374129B (en) Method for generating synchronization sequence based on OFDM, synchronization method and system
CN102369707B (en) Method and device for eliminating co-channel interference on pilot frequency
WO2012171407A1 (en) Method and device for determining time synchronization location
CN101741800A (en) Synchronous searching method
Nasraoui et al. A simply-differential low-complexity primary synchronization scheme for 3GPP LTE systems
CN100521554C (en) Frequency domain channel estimation method based on two-value full-pass sequence protection interval filling
CN117320046A (en) CRS searching method, LTE time alignment error measuring method and user equipment
CN107276654B (en) Signal processing method and system
CN107277913B (en) timing synchronization method, device and system

Legal Events

Date Code Title Description
C06 Publication
PB01 Publication
C10 Entry into substantive examination
SE01 Entry into force of request for substantive examination
C14 Grant of patent or utility model
GR01 Patent grant
CF01 Termination of patent right due to non-payment of annual fee

Granted publication date: 20130717

Termination date: 20180415

CF01 Termination of patent right due to non-payment of annual fee