CN101854120A - A High Efficiency Multifunctional Flyback Converter - Google Patents

Abstract

The invention relates to a high-efficiency multifunctional flyback converter. The invention discloses a lossless resonant absorption circuit and integrates the operating principle of the lossless resonant absorption circuit into the operation of the flyback power converter to form a high-performance high-efficiency converter structure. In the operation of the converter, the absorption circuit can absorb leakage inductance of a transformer for energy storage at no loss to effectively suppress the over-voltage peak of a power switch. Moreover, energy stored by an absorption capacitor and the inductance of the transformer are utilized to form resonance to create zero voltage condition for the power switch to realize zero voltage switching operation. A unique self-driven synchronizing rectifier circuit in a secondary loop can further improve the efficiency of the converter.

Description

A High Efficiency Multifunctional Flyback Converter

background introduction

Field of invention

This patent belongs to an invention in the field of electric power conversion, more specifically a unique flyback power converter structure and its operation method. This type of converter can realize high-efficiency DC-DC voltage conversion or the composite function of single-stage power factor adjustment plus DC-DC voltage conversion. A low-cost, high-performance design concept is provided for power conversion devices.

Description of related fields

With the increasingly urgent demand for environmental protection, people are increasingly demanding the use of green energy in various fields. In the field of electric energy utilization, it is necessary to further improve the efficiency of electric equipment and electric energy conversion devices and use as few materials as possible. Another imperative requirement is to improve the power factor of the AC power device to reduce the transmission loss of electric energy. In our daily life, many semiconductor electronic devices need a power conversion device that converts AC power into DC power in order to obtain the DC power required for operation from the AC mains network. In this case, if an AC-DC power converter with high efficiency, low cost, low material consumption and power factor adjustment can be designed and manufactured, the extensive environmental and economic value it brings to people is obvious .

Traditional AC-DC converters currently on the market generally adopt two common approaches. One is to first use a diode rectifier bridge to convert the AC voltage into a pulsating DC, use a capacitor to filter out the pulsating ripple, and then use a DC-DC (referred to as DC-DC) converter to convert the filtered DC voltage into the required voltage value output and provide safe electrical isolation between DC output and AC input. The typical circuit structure of this approach is shown in Figure 1. In Figure 1, the rectifier bridge BG1 converts the AC voltage into a pulsating DC, and then the capacitor C1 filters out the pulsating ripple. Electronic power switch Q1, transformer TX1, secondary rectifier diode D2 and secondary filter capacitor C2 form a basic flyback DC-DC conversion circuit. R3 and C3 are a typical snubber circuit to absorb the energy stored in the primary leakage inductance of the transformer during the switching process, thereby suppressing the switching voltage peak caused by it on the electronic switch. It should be pointed out here that in addition to the flyback conversion circuit, there are other circuit structures such as forward type, push-pull type, half-bridge circuit, full-bridge circuit, etc., which can complete the DC-DC conversion function. Flyback circuits typically use the fewest components and cost the least. This circuit structure is usually used in occasions with low power and low power factor.

When the power consumption is large, it is generally necessary to add a power factor adjustment (abbreviated as PFC) circuit at the input end to achieve a higher power factor. A typical AC-DC converter with a power factor adjustment circuit is shown in Figure 2. Compared with the circuit in Figure 1, the electronic switch Q11, inductor L11, diode D11 and capacitor C11 form a power factor adjustment circuit. The function of the power factor adjustment circuit is mainly to make the envelope of the current waveform of the inductor L11 maintain a full-wave rectified sinusoidal waveform and be in phase with the rectified pulsating sinusoidal voltage through the switching operation of the electronic switch Q11. In this way, the AC current input from the AC input terminals AC1 and AC2 of the rectifier bridge BG1 naturally maintains a sinusoidal waveform and is in phase with the input voltage, so that the power factor can reach an ideal state close to 1. The electronic power switches in Figure 1 and Figure 2 are most commonly used with MOSFETs. Other types of switching devices such as bipolar transistors or IGBTs can also be substituted.

The circuits shown in Figures 1 and 2 generally have low operating efficiencies. The electronic power switches Q1 and Q11 shown in the figure both work in the hard switching working state. Especially when they are switched from off to on state, because the potential difference between the drain and the source changes from high voltage to close to zero, the high-voltage energy storage of the parasitic capacitance between the source and drain is forced to discharge through the switch itself in a very short moment. , all of its energy is consumed inside the switch tube and converted into heat energy. This will not only reduce the efficiency, but also increase the heat generation of the tube, and also generate strong electromagnetic radiation. Another loss factor is the leakage inductance energy of the transformer. When Q1 is turned on, electromagnetic energy is gradually built up with the increase of transformer TX1 primary winding PN1 current. When Q1 is turned off, the electromagnetic energy stored in the coupled inductor is coupled to the secondary winding SN2 through the magnetic circuit, and the rectifier diode D2 is turned on to supply power to the output terminal. The energy stored in the leakage inductance cannot be coupled to the secondary, so it can only be maintained through the source-drain capacitance of Q1 and transferred to the source-drain capacitance by charging. In this case, the drain voltage of Q1 may be pushed very high, even causing Q1 overvoltage breakdown. In order to suppress this voltage overshoot phenomenon, people usually have to use a snubber circuit to absorb and consume this part of leakage inductance energy. The R3 and C3 network in Figure 1 and Figure 2 is a kind of absorption circuit. There are many different snubber circuit designs in practical applications. These circuits are generally well known and therefore will not be described in detail here. In addition, the loss of the rectifier diode D2 cannot be ignored. Especially when the output is low voltage and high current, the turn-on voltage drop of D2 may account for 5% to 10% of the output voltage or even higher, resulting in a proportional power loss. On the other hand, due to the non-ideal reverse recovery characteristics of the diode, the switching loss caused by it is also considerable.

Summary of the invention

The purpose of the present invention is to overcome the disadvantages of the above-mentioned traditional conversion circuit, and provide a converter solution with high performance, low power consumption, simple circuit and low cost. The invention adopts a lossless active absorbing resonant circuit to restrain the switch overshoot voltage. And the LC resonance principle is used to make the power switch work in the soft switching state or close to the soft switching state. The invention also adopts the high-efficiency conversion circuit to realize the single-stage AC-DC voltage conversion function with power factor adjustment. Combining the advantages of the lossless active snubber circuit and the reduction of power conversion links, this single-stage multifunctional conversion circuit can achieve higher conversion efficiency with lower product cost. At the same time, the invention also adopts a simple and easy synchronous rectification circuit to replace the diode rectifier, thereby further improving the working efficiency of the converter.

Description of drawings

Figure 1 shows a typical traditional flyback converter circuit structure.

Figure 2 shows a traditional AC-DC converter structure with power factor adjustment.

Fig. 3 has described the circuit structure and principle of a high-efficiency converter with lossless active absorption resonant circuit of the present invention.

FIG. 4 describes another high-efficiency converter circuit structure and principle with a lossless active absorption resonant circuit of the present invention.

FIG. 5 describes a synchronous rectification control circuit of the present invention.

FIG. 6 illustrates another synchronous rectification control circuit of the present invention.

Detailed description of the invention

As mentioned above, the energy storage of the leakage inductance of the traditional flyback converter and the consumption absorbing circuit used to suppress the voltage spike caused by the switching process, as well as the switching loss of the power switch tube are the key factors in the working process of the flyback converter. main loss factor. The core concept of the invention is to use the resonant characteristics of controllable inductance and capacitance to form a lossless absorption circuit. On the one hand, the circuit can effectively absorb the energy of the leakage inductance of the transformer without loss and suppress the switch voltage peak when the power switch tube is turned off. At the same time, before the power switch tube is turned on, the resonance between the energy storage of the absorbing circuit and the transformer inductance can be used to generate a zero voltage condition for the power switch, thereby realizing zero voltage soft switching operation. One of its conceptual circuits is shown in Figure 3(A). In FIG. 3(A), an N-type metal oxide field effect transistor (hereinafter referred to as MOSFET) Q1 is the main power switch. The drain of Q1 is connected to the inverting end of the primary winding of the transformer TX1 (it should be noted here that the inverting end refers to the end without a phase point mark, and the naming of inverting and non-inverting phases is a relative concept, and the inverting end of the primary and the secondary The inverting terminal is the non-inverting terminal, and vice versa). The source of Q1 is connected to the negative terminal of the DC voltage rectified by the rectifier bridge BG1 through the current detection resistor R1, that is, the DC ground terminal PGND. The non-inverting terminal of the primary winding of TX1 is connected to the rectified DC positive terminal VDC+ of BG1. The series loop composed of capacitor C3 and N-type MOSFET Q3 connected across the two ends of the primary winding of TX1 is a lossless absorption circuit. The rectangular block CU1 in FIG. 3 represents the control and drive circuit. The control driving circuit has two driving output signals DR1 and DR2. DR1 is used to drive Q1, and DR2 is used to drive Q3. CU1 accepts two feedback input signals simultaneously. The input signal ISN1 accepts the Q1 source current signal sensed by R1. The input signal VS1 is the voltage VQ1D of the drain Q1D of Q1 detected by the voltage detection network composed of C4, R2, and R4. The circuit composed of C4, R2, and R4 is a differential detection network with a voltage divider function. The voltage change rate of VQ1D transmitted through C4 is divided by R2 and R4 and fed back to CU1 to control the driving circuit CU1. The secondary output of the transformer TX1 is rectified and filtered by D2 and C2 to finally generate the required DC output Vo.

Figure 3 (A) The main working signal waveform of the circuit is shown in Figure 3 (B). During the working process, when Q1 starts to conduct at to, the current in the primary winding PN1 of the transformer is gradually established through the flow loop from VDC+ through PN1, Q1 and R1 to PGND. When Q1 is turned off at t1, the energy stored in the coupled inductance of PN1 is coupled to the secondary winding SN2 of TX1 through a magnetic circuit and turns D2 on, supplying power to C2 and the load connected between Vo and GND. Since the energy stored in the PN1 leakage inductance cannot be coupled to the secondary, it will naturally charge the source-drain parasitic capacitance of MOSFET Q1 under the rule of maintaining its current continuity, and also charge C3 through the parasitic diode of Q3 Charge, and cause the voltage on these capacitors to continue to rise until the leakage current decays to zero at t2. At this time, the electromagnetic energy stored in the leakage inductance is all transferred to the parasitic capacitance of the source and drain of Q1 and the capacitance C3, and the voltage on these capacitances reaches its peak value. The voltage on C3 is negative at the top and positive at the bottom. If Q3 is turned on during this period, the charging of C3 is completed through Q3. The peak voltage on the capacitor and the magnitude of the leakage inductance energy are directly related to the size of the capacitor. Under the same leakage inductance energy, the larger the capacitance value, the smaller the peak voltage on the capacitance when the leakage inductance energy transfer ends. Because a lossless absorption circuit is used here, C3 can use a larger capacitance value, which can effectively reduce its voltage peak value, so that the drain voltage VQ1D of Q1 is also clamped at a lower level. At the same time, the capacitance of the parasitic capacitance of the source and drain of C3 and Q1 is directly proportional to the energy absorbed by them. The larger C3 is, the larger the energy ratio it absorbs, and the smaller the energy absorbed by the parasitic capacitance of Q1. The control signal waveform of Q3 is shown in Fig. 3(B). As shown in the figure, Q3 is on during the period from when Q1 is turned off at t1 to when the voltage of C3 reaches its peak value at t2, and the process of transferring leakage inductance energy to C3 is completed through Q3. This not only reduces the loss caused by the turn-on voltage drop of the parasitic diode of Q3, but more importantly, avoids the oscillation caused by the reverse recovery characteristic of the parasitic diode. When the voltage of C3 reaches the peak value, Q3 is turned off, its parasitic diode is in the reverse cut-off state, the path between C3 and PN1 is cut off, and the voltage of C3 is maintained at its peak value for the next period of time.

During t1 to t3 the energy stored in the coupled inductance of the primary winding of the transformer is continuously transferred to the secondary via D2 to power C2 and the load. At time t3, this part of the stored energy is exhausted, the secondary current is interrupted, the source-drain parasitic capacitance of Q1 starts to discharge to the primary winding of the transformer, and the voltage of Q1 drain Q1D starts to drop. At this time, Q3 starts conducting again. The energy stored in C3 starts to transfer to the primary winding PN1 of the transformer and forms a resonance. The first 1/4 period of resonance is the interval from t3 to t4, and the energy of C3 is transferred to PN1. At time t4, the voltage of C3 drops to zero and the current of PN1 reaches the maximum value. The second 1/4 period of the resonance begins at time t4, and the current of PN1 charges C3 at the beginning, and the voltage on C3 gradually builds up with the charging, and the polarity is up positive and down negative. As the voltage of C3 increases, the potential of Q1D decreases gradually accordingly. If the resonant energy is large enough, the voltage of C3 can be equal to VDC+ before t5 or t5. At this time, the parasitic diode of Q1 is turned on, and the drain potential of Q1 is close to zero, which creates conditions for the zero-voltage switching operation of Q1. . At time t5, Q3 is turned off, and Q1 is turned on subsequently, thus completing the soft switching operation of Q1 and beginning to repeat the operation process of the next cycle.

It should be noted here that the zero voltage condition of Q1 can not be created by resonance in all cases. When the resonance energy is small, the voltage of C3 cannot reach the value of VDC+ at the end of the second 1/4 cycle, so the drain voltage of Q1 cannot reach zero. In this case, the switching point of Q1 and Q3 is selected at the lowest point of VQ1D voltage, that is, the resonance valley point. In this way, although the zero-voltage switching operation cannot be achieved, the switching loss will still be much lower than the traditional operation method. In addition, in the traditional flyback converter design, how to make the leakage inductance of the transformer as small as possible is a very challenging problem. The larger the leakage inductance of the transformer, the greater the switching loss. In the case of this solution, most of the leakage inductance energy of the transformer is not consumed, but creates a zero voltage or minimum voltage switching condition for the power switch Q1 through resonance with the absorbing capacitor C3. Therefore, the requirements for the leakage inductance of the transformer are not so strict, and the design and manufacture of the transformer is relatively easier.

At the same time, it should be pointed out that for the switch control of Q3, it is only a conceptual example to monitor the drain voltage of Q1 with a detection network composed of C4, R2, and R4. Other kinds of voltage detection circuits can also be used to achieve the same purpose. In addition to sensing the drain voltage of Q1, by monitoring the source current signal of Q3, it can also be used to help control the switching operation of Q3. Since the source of Q3 is floating, it is more convenient to use a current mutual inductance transformer for its current signal detection. In order to keep the diagram simple and easy to understand, this part of the circuit is not shown in Figure 3(A). When the voltage signal obtained by VSN1 is used for control, the switching point of Q3 is selected at the zero-crossing point of the voltage change rate. Because the zero crossing point of the voltage change rate is also the zero crossing point of the current. Therefore, when the current signal is used to control, the switching point of Q3 is selected at the zero-crossing point of the current. For the output of the converter at the Vo end, it can be used as a voltage type or a current type output control. When the output voltage needs to be controlled, the output voltage at the Vo terminal is fed back to the control drive circuit CU1 to control the switching operation of Q1. And when the output current needs to be controlled, the current signal at the Vo or GND terminal is fed back to the control drive circuit CU1 to control the switching operation of Q1. The implementation methods of the above concepts are familiar to those in the industry, so they are not shown in FIG. 3(A). In addition, the switching devices Q1 and Q3 used in the circuit of FIG. 3(A) can also be replaced by a combination of a bipolar NPN transistor and an antiparallel diode. The corresponding relationship of the ports is shown in Fig. 3(C).

The circuit shown in Fig. 3(A) can not only realize the AC-DC conversion function without PFC function, but also realize the single-stage AC-DC converter with PFC function. When the PFC function is not needed, the capacitance of C1 can be selected relatively large, so that the VDC+ filtered by C1 is close to pure DC. When the PFC function is required, the capacitance of C1 is selected to be relatively small, as long as the capacitance is enough to filter out the ripple generated by the high-frequency switching operation of Q1, and there is basically no low-frequency sinusoidal pulsating voltage for the AC input rectified by BG1 filtering effect. This VDC+ basically maintains the sinusoidal pulsation waveform of sinusoidal AC after full-wave rectification. At this time, the current envelope control of Q1 and PN1 takes the sinusoidal ripple waveform of VDC+ as the reference signal, and finally through the switch control of Q1, the current input from the AC input terminals AC1 and AC2 has a sinusoidal waveform and is in phase with the input voltage to achieve a power factor. effect close to 1. Note that in this case, the secondary current of the transformer also follows the low-frequency sinusoidal pulsation waveform of VDC+ like the primary, so C2 needs to use a large-capacity capacitor to filter out these low-frequency pulsation components. If the requirement on the ripple coefficient of the output voltage is strict, it is also possible to consider adding an LC filter circuit after C2. It should be emphasized here that if there is no lossless resonant snubber circuit in Figure 3(A), the switching loss caused by the leakage inductance of the transformer TX1 will greatly reduce the performance of this single-stage PFC integrated AC-DC converter. efficiency. Due to the use of the lossless resonant snubber circuit in Figure 3(A), the energy stored in the leakage inductance of the transformer during the switching process is not consumed, but is used to reduce the switching loss of the power switch, so the overall efficiency will be much higher than that shown in Figure 3(A). The traditional two-stage conversion circuit shown in 2 provides a green AC-to-DC converter solution with high efficiency, less material use and low product cost.

Figure 4(A) shows a converter circuit using another lossless resonant snubber circuit. The circuit parts that are the same as those shown in FIG. 3(A) will not be repeated here. The main difference from Figure 3(A) is that C3 and Q3 are connected between the drain of Q1 and the primary DC power ground PGND instead of being connected across PN1. Q3 is also replaced by a P-type MOSFET from an N-type MOSFET, and a current detection resistor R3 is connected in series between the source of Q3 and PGND to feed back the current signal of Q3 to the feedback input terminal ISN2 of the control drive circuit CU1. As mentioned in Section [0012], the current feedback signal of Q3 can also be used to help control the switching operation of Q3, and the current signal on R3 in Figure 4(A) is based on PGND as the reference point, so the feedback circuit is very easy and convenient. The reason for using P-type MOSFET is because the polarity of its parasitic diode is opposite to that of N-type MOSFET, and the polarity of the parasitic diode of N-type MOSFET makes the discharge process of C3 to PN1 in the circuit of Figure 4 (A) uncontrollable. Generally speaking, it is not as convenient to use P-type MOSFET for Q3 as to use N-type MOSFET, but because the source uses PGND as the reference point, the gate drive signal does not need to float, and the drive circuit is relatively simple.

The switching control rules of Q1 and Q3 in Figure 4(A) are the same as those of the circuit in Figure 3(A). Its main signal waveform is shown in Fig. 4(B). Q1 is turned on in the interval from time to to t1, and the transformer primary winding current flows through PN1, Q1, and R1 under the action of VDC+ and increases linearly. The on-pulse width from to to t1 is determined by the PWM modulation circuit controlling the driving circuit CU1 according to the regulation requirement of the output voltage or current of the Vo terminal. At t1, Q1 is turned off, and Q3 is turned on to bypass its parasitic diode. The leakage inductance energy of PN1 charges C3 until all the energy is transferred to C3. At t2, the voltage of C3 reaches a peak value, and the current flowing through Q3 drops to zero. At this time, Q3 is turned off, and the voltage of C3 is maintained at the peak state. During this period, the energy stored in the PN1 coupling inductance is continuously coupled to the secondary side to supply power to the C2 and Vo end loads. When this part of energy is exhausted at t3, the transformer secondary winding current decays to zero. The positive and negative voltages at both ends of the primary winding PN1 and the voltage at the drain Q1D of Q1 begin to drop. At this time, Q3 is turned on again so that the inductance of C3 and PN1 can exchange energy through resonance. During the first 1/4 period of resonance between t3 and t4, the energy stored in C3 is gradually transferred to the inductor current of PN1 until VQ1D is equal to VDC+ at t4, and the current of PN1 reaches the maximum value at this time. In the second 1/4 period of resonance, the inductance current from t4 to t5PN1 reversely charges C3 through VDC+, C1, R3 and Q3. At time t5 the inductor energy in PN1 is depleted, the Q3 current drops to zero, and the Q1 drain voltage VQ1D drops to the bottom. If the resonant energy is large enough, the C3 voltage and the Q1 drain voltage can be pushed to zero, creating the conditions for the zero-voltage switching operation of Q1. When the Q1 drain voltage is pushed to zero or valley, Q3 turns off and the Q1 gate drive signal turns Q1 on, thus achieving zero voltage or minimum voltage switching operation.

As described in Section [0012], because the resonant frequency of the circuit is basically determined by the L, C and R values of the components participating in the resonance, so when the inductance parameters of the transformer, the capacitance of C3, and the parasitic capacitance of the source and drain of Q1 After the value and the resistive impedance in the resonant circuit are determined, the resonant frequency is also basically fixed. Based on this theory, the Q3 conduction time from t1 to t2 and the Q3 conduction time from t3 to t5 can also be controlled by an approximate fixed time according to the resonant frequency characteristics of the circuit. That is to say, the time when VQ1D rises from t1 to t2 reaches its peak, and the time from t3 to t5 when it reaches the valley point can be approximately determined by the resonant frequency of the circuit and is basically constant. Therefore, Q3 is between t1 and t5 The conduction time from t2 and t3 to t5 can also be set according to this time. In this case, the control driving circuit only needs to control the conduction time of Q3 through a timing circuit instead of real-time monitoring of the drain voltage VQ1D of Q1 or the source current of Q3. The timing circuit starts simultaneously with the gate-on signal of Q3 at t1 and t3, and turns off Q3 when the set time is reached at t2 and t5. It should be noted here that the inductance component participating in the resonance during the resonance process between t1 and t2 is the leakage inductance of the primary coil PN1 of TX1, and the inductance component participating in the resonance between t3 and t5 is the self-inductance of PN1, so the resonance frequency of these two intervals is different.

Also as shown in FIG. 3(A), the feedback signal loop from the output terminal Vo to the control drive circuit CU1 is not shown in FIG. 4(A). In a specific application, if the object to be controlled and adjusted is the output voltage, the Vo voltage signal is taken as feedback. If the object to be controlled and adjusted is the output current, the output current signal at the Vo terminal or the GND terminal is taken as feedback. This type of feedback circuit is familiar to those skilled in the art, so details will not be repeated here. Also the circuit shown in Figure 3(A), the circuit shown in Figure 4(A) can realize a DC-DC or AC-DC converter without PFC function, and can also control the primary winding current of the transformer through the switching operation of Q1 The winding line follows the pulsating sinusoidal waveform of the input voltage after full-wave rectification, thereby realizing a single-stage AC-DC converter with PFC function. In addition, the P-type MOSFET Q3 in Figure 4(A) can also be replaced by a combination of a bipolar PNP transistor and an antiparallel diode. As shown in Figure 4(C).

In addition to using a lossless resonant absorption circuit in the primary circuit of the transformer, the present invention can also use a synchronous rectification circuit in the secondary circuit to replace the traditional diode rectification to further improve efficiency. Taking a converter with 5V output as an example, when a Schottky diode is usually used for rectification, its forward voltage drop is about 0.3V to 0.5V. Compared with the 5V output, the loss caused by the forward voltage drop of the diode accounts for about 6% to 10% of output power. In synchronous rectification, MOSFETs are often used to replace rectifier diodes, and its conduction voltage drop can be controlled between 0.1V and 0.15V, that is, the efficiency can be increased by about 4% to 7%. At present, there are some implementation methods for synchronous rectification. One of the typical methods is to add an auxiliary drive winding on the secondary side of the transformer, and use the voltage of the auxiliary drive winding to drive the synchronous rectifier. The control circuit of this approach is relatively simple, but the transformer must add an additional winding for each synchronous rectifier. This will greatly increase the complexity and fabrication difficulty of the transformer in the case of multiple outputs. Another approach is to use an auxiliary drive transformer or photocoupler to couple the synchronous rectification control signal from the primary control drive circuit to the secondary and match the corresponding drive circuit to drive the synchronous rectifier. The cost of this approach will be relatively high, and the number of components used will be relatively large.

The synchronous rectification circuit of the present invention is shown in FIG. 5 . The generation of the synchronous rectification control signal does not require auxiliary drive windings, nor does it need to use auxiliary transformers or photoelectric devices to couple from the primary, but uses a unique circuit to directly process and generate according to the voltage and current changes of the secondary circuit. As shown in Figure 5(A), the rectifier diode D2 is replaced by an N-type MOSFET Q2, and moved from the positive output terminal to the output ground loop. This transposition is mainly to connect the source of Q2 to GND, so that the gate drive signal is easier to handle. Q5 and Q6 are PNP transistors, which form a differential circuit. R9 is connected between the emitters of Q5 and Q6 and Vo as a common current source resistance. The collector output of Q5 is divided by resistors R11 and R12 to drive the gate of Q2. The base of Q5 is driven by a voltage divider between Vo and the drain of Q2 by R5 and R6. The base of Q6 is driven by voltage division between Vo and GND, which is the source of Q2, by R7 and R8. The parameter selection of R5, R6, R7 and R8 makes the voltage division ratio of R7 slightly larger than that of R5. When the voltage at both ends of the R5, R6 loop and the R7, R8 loop are equal, the voltage on R7 is slightly greater than the voltage on R5, but when the voltage at both ends of the R5, R6 loop is higher than the R7, R8 loop by about tens of millivolts, the voltage of R5 drop above the voltage drop across R7. In this way, when the source-drain voltage of Q2 drops to zero, Q6 is preferentially turned on, Q5 is turned off, and Q2 is also in an off state. When Q1 is turned off during circuit operation, the voltage of the secondary winding of the transformer is positive up and down negative. The voltage needs to be raised to a level higher than the voltage across C2 (that is, the output voltage) by the forward conduction voltage drop of the parasitic diode of Q2 so that the secondary current can form a circulation loop. In this case, the voltage received by the voltage divider circuit composed of R5 and R6 is higher than the voltage accepted by the circuit of R7 and R8. The design of the voltage division ratio of the above two voltage divider circuits makes the voltage drop on R5 higher than the voltage drop on R7 in this case, so that Q5 is turned on and Q6 is turned off, so that the gate of Q2 obtains a positive voltage from Vo through Q2. Drive voltage, Q2 is turned on, and the secondary winding current supplies power to C2 and the load through Q2. When the secondary current is exhausted, the turn-on voltage drop of Q2 is zero, at this time, Q6 is turned on, Q5 is turned off, and Q2 is turned off, preventing the voltage of C2 from discharging through the secondary winding SN2 and Q2. When Q1 is turned on in the next switching operation cycle, the voltage of winding SN2 becomes lower positive and upper negative, Q5 cuts off more deeply, Q6 is still in the on state, Q2 is still cut off, and continues to prevent C2 from discharging until Q1 is turned off, and the circuit repeats as follows The aforementioned process performs another cycle of synchronous rectification operation.

Based on the same operating principle, the synchronous rectification circuit of the present invention can also be transformed into another structure as shown in FIG. 5(B). Compared with Fig. 5(A), Fig. 5(B) is basically its dual circuit. Q2 changes from an N-type MOS tube to a P-type MOS tube, and switches to the output positive end circuit, and Q5 and Q6 also change from PNP-type transistors to NPN-type transistors. Its working principle is the same as that in Figure 5(A), so it will not be repeated here. It should be reminded that Q5 and Q6 can also use small-signal MOSFETs or other active electronic devices to achieve the same function. Here, if the driving gain of the differential circuit composed of Q5 and Q6 is not enough, another level of non-inverting amplifier circuit can be inserted between the collector of Q5 and the gate of Q2 to increase the driving capability. Q4 shown in Figure 6(A) and Figure (B) is an example. Another possible approach is to drive the gate of Q2 through the output of the collector of Q6 through a first-stage inverting amplifier. Because the reason is obvious, it will not be repeated here.

At the same time, it should also be pointed out that the above description and related diagrams are mainly used as examples to illustrate the principles and concepts of the present invention. In practice the same concept can be implemented in different forms. Therefore, the application of this patent is not limited to the implementation methods described herein without departing from its basic concept. Also, the rights described in the ensuing declaration of rights are based on a principled concept, and can be realized with different circuits without violating the basic principles.

Claims (9)

1. A converter circuit, a rectifier bridge rectifies the AC input to DC, a capacitor across the DC output of the rectifier bridge, a flyback transformer, one end of its primary winding is connected to the positive voltage output of the rectifier bridge, The other end is connected to the positive voltage port of a main electronic switch, the negative voltage port of the electronic switch is connected to the negative voltage output port of the rectifier bridge through the current detection element, and a lossless absorption circuit is formed by connecting a capacitor and an auxiliary electronic switch in series. become. One end of the capacitor is connected in series with the positive voltage end of the auxiliary electronic switch, the other end is connected with the positive voltage output node of the rectifier bridge, the negative voltage end of the electronic switch is connected with the positive voltage end node of the main electronic switch, one consists of a capacitor and two resistors A voltage detection network formed in series, the network is connected between the positive voltage port node of the main electronic switch and the negative voltage output node of the rectifier bridge, one end of the capacitor is connected to the positive voltage port node of the main electronic switch, the other end is connected to the two A voltage divider network composed of resistors in series is connected, and the middle node of the two series resistors is connected to a feedback input terminal of a control drive circuit, which receives signals from the above-mentioned voltage detection network, the main electronic switch current detection element, and the secondary The current or voltage signal at the output terminal is feedback, and a drive signal is generated to drive the above-mentioned main electronic switch and auxiliary electronic switch, a rectifier diode, the anode of which is connected to the secondary winding of the flyback transformer and the reverse phase of the port where the primary winding is connected to the positive end of the rectifier bridge port, the cathode is connected to the positive terminal of an output filter capacitor and is the positive output terminal of the converter, and the negative terminal of the filter capacitor is connected to the other end of the secondary winding of the above-mentioned transformer and becomes the negative output terminal of the converter. 2. Another converter circuit, the other parts of which are the same as those described in (1), except that a lossless snubber circuit connected in series by a capacitor and an auxiliary electronic switch is connected across the positive voltage of the main electronic switch Between the port node and the negative voltage output terminal of the rectifier bridge, one end of the capacitor is connected to the positive voltage port node of the main electronic switch, the other end is connected to the negative voltage port of the auxiliary electronic switch, and the positive voltage port of the auxiliary electronic switch is connected to the rectifier bridge Negative voltage output port. 3. In the circuit operation described in (1) and (2), use a lossless resonant snubber circuit to absorb the voltage spike when the main electronic switch is turned off, and use the resonance formed by the snubber circuit and the transformer inductance to power the main electronic circuit. The switch realizes the operating characteristics of zero-voltage switching or minimum voltage switching. When the main electronic switch is turned off, the auxiliary electronic switch is turned on and turned off when the voltage at both ends of the main switch reaches the maximum value, so that the leakage inductance energy of the transformer is stored without loss in the On the absorption capacitor, when the transformer coupling electromagnetic energy is exhausted, the auxiliary switch is turned on again and uses the resonance characteristics of the absorption capacitor and the primary inductance of the transformer to reduce the voltage across the main switch. When the main switch voltage reaches zero or the lowest point, the auxiliary switch is turned off. switch and turns on the main switch, resulting in zero-voltage or minimum-voltage switch-on operation. 4. In the circuit operation described in (1) and (2), the operation of the auxiliary switch is controlled by monitoring the change rate of the main switch voltage or the current of the auxiliary switch. When the main switch is turned off, the auxiliary switch is turned on. When the switch voltage rises to the peak value, when its rate of change crosses zero for the first time, or when the auxiliary switch current crosses zero, the auxiliary switch is turned off, and before the main switch is turned on, the auxiliary switch is turned on when the transformer coupling current decays to zero, which When the resonance of the absorption capacitor and the primary inductance of the transformer causes the main switch voltage to drop, when the main switch voltage drops to the lowest value, the change rate crosses zero for the first time, or when the main switch voltage is zero, or when the auxiliary switch current is zero, Turn off the auxiliary switch and turn on the main switch. 5. Or in the circuit described in (1) and (2), the conduction time of the auxiliary switch is regularly controlled according to the natural resonance period of the circuit, and after the main switch is turned off, the auxiliary switch is turned on and maintained Turn on the auxiliary switch after about a quarter of the resonance period of the circuit state at the moment, and when the coupling current in the transformer winding decays to zero, the auxiliary switch is turned on and maintained at about half of the circuit state at the moment After one resonant period, the auxiliary switch is turned off, and then the main switch is turned on. 6. Using the circuits described in (1) and (2), it is possible to realize the function of a high-efficiency DC-DC converter or an AC-DC converter without power factor adjustment, and to maintain the voltage rectified by the rectifier circuit It is a full-wave sinusoidal pulse waveform and through the operation control of the main switch, the input current envelope at the AC input terminal follows the sinusoidal waveform in phase with the input power supply voltage, so that this type of single-stage conversion circuit can be used to achieve high efficiency with power factor Integrated conversion function for regulation and DC-DC conversion. 7. Using the circuits described in (1) and (2), the output voltage can be used as the control target feedback signal to control the operation of the switch tube to drive the voltage type load, and the output current can also be used as the control target feedback signal to control The operation of the switch tube is used to drive the current-type load. At the same time, the product of the output voltage and the output current can be used as the control target feedback signal to control the operation of the switch tube to drive the power-type load. 8. A synchronous rectification circuit, using an N-type MOSFET as a synchronous rectification device, its source is connected to the output ground or negative terminal, the drain is connected to the transformer secondary winding and the primary winding is connected to the same phase port of the input voltage positive terminal, Or use a P-type MOSFET as a synchronous rectification device, its source is connected to the positive voltage output port, the drain is connected to the secondary winding and the primary winding is connected to the positive input voltage port, and a differential circuit is used to control the synchronization The gate drive of the rectifier tube, one input terminal of the differential circuit obtains the divided voltage signal from both ends of the secondary winding of the transformer, and the other input terminal obtains the divided voltage signal from the positive and negative ends of the output voltage, the output signal of the differential circuit or The amplified output signal is used to control the switching operation of the synchronous rectifiers. 9. In the circuit operation described in (8), when the transformer secondary winding has current to supply power to the output, the output of the differential control circuit makes the synchronous rectifier conduction, when the transformer secondary winding current is zero, or the secondary When the winding voltage is in the reverse direction, that is, when the bypass diode of the synchronous rectification device tends to be reverse-biased, the output of the differential control circuit makes the synchronous rectifier tube cut off, preventing the reverse discharge of the output capacitor.

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