CN101488939B

CN101488939B - Method, device and receiver for realizing symbol synchronization in broadband wireless communication system

Info

Publication number: CN101488939B
Application number: CN200910078616XA
Authority: CN
Inventors: 李晔; 秦龙; 胡宏杰; 安东尼·宋
Original assignee: Huawei Technologies Co Ltd
Current assignee: Huawei Technologies Co Ltd
Priority date: 2009-02-27
Filing date: 2009-02-27
Publication date: 2011-08-03
Anticipated expiration: 2029-02-27
Also published as: CN101488939A

Abstract

A method, device and receiving method for realizing mark synchronization in bandwidth wireless communication system, mainly including: firstly, acquiring multi-path positions of a channel in a communication system and power information of each path, and calculating total signal to interference and noise ratios of all sub-carrier marks corresponding to a starting position in a quick Fourier transformation FFT integral internal in a pre-setting starting position set, wherein there is a plurality of starting positions of FFT integral internal in the pre-setting starting position set; then setting thestarting position of the integral internal corresponding to the greatest total signal to interference and noise ratio in the total signal to interference and noise ratios acquired by calculating as amark timing position, performing a synchronization process to a received signal. The mark synchronization technology based on the greatest total signal to interference and noise ratio standard cannotbe influenced by the channel estimation error, and has excellent performance robustness and advantage of simplicity and practicality under the condition that the channel delay spread is greater than CP length.

Description

Method, device and receiver for realizing symbol synchronization in broadband wireless communication system

技术领域technical field

本发明涉及无线通信领域，尤其涉及宽带无线通信系统中实现符号同步的技术。The invention relates to the field of wireless communication, in particular to a technology for realizing symbol synchronization in a broadband wireless communication system.

背景技术Background technique

以OFDM(Orthogonal Frequency Division Multiplex，正交频分复用)系统为例，在相应的宽带系统中，存在着无线信道时延扩展带来的ISI(inter-symbol interference，符号间干扰)，包括IBI(Inter block interference，块间干扰)和ICI(Inter carrier interference，载波间干扰)。为此，OFDM系统通过插入CP(Cyclic Prefix，循环前缀)来对此加以克服，但前提是CP要足够长，要大于信道最大时延扩展长度，否则，就会导致系统中存在相应的IBI和ICI。Taking the OFDM (Orthogonal Frequency Division Multiplex) system as an example, in the corresponding broadband system, there are ISI (inter-symbol interference, inter-symbol interference) caused by wireless channel delay expansion, including IBI (Inter block interference, inter-block interference) and ICI (Inter carrier interference, inter-carrier interference). For this reason, the OFDM system overcomes this by inserting a CP (Cyclic Prefix, cyclic prefix), but the premise is that the CP must be long enough and greater than the maximum channel delay extension length, otherwise, there will be corresponding IBI and IBI in the system. ICI.

进一步地，参照图1所示的单径信道对上述涉及到的同步的影响进行具体说明。x_n(t)表示第n个OFDM符号，符号周期长度为T_s，CP长度为T_G，则整个符号长度为T＝T_s+T_G。接收端为了在频域进行均衡，需要通过FFT(FastFourier Transform，快速傅里叶变换)将基带接收信号变换到频域，FFT积分长度为T_s，由符号同步提供积分区间的起始位置τ。基于OFDM的基本原理可知，对图1所示的“Case(i)”，由于积分起始位置在CP范围内，不存在IBI和ICI，而对“Case(ii)”和“Case(iii)”，则同时存在IBI和ICI。Further, with reference to the single-path channel shown in FIG. 1 , the impact on the above-mentioned synchronization will be described in detail. x _n (t) represents the nth OFDM symbol, the symbol period length is T _s , the CP length is T _G , and the entire symbol length is T=T _s +T _G . In order to perform equalization in the frequency domain, the receiving end needs to transform the baseband received signal to the frequency domain through FFT (FastFourier Transform). The FFT integration length is T _s , and the start position τ of the integration interval is provided by symbol synchronization. Based on the basic principle of OFDM, it can be seen that for "Case (i)" shown in Figure 1, since the starting position of the integration is within the range of CP, there is no IBI and ICI, while for "Case (ii)" and "Case (iii) ”, then IBI and ICI exist at the same time.

针对存在的IBI和ICI，目前可以通过选择合适的积分区间的起始位置的方式，最小化由于CP长度不足所导致的IBI和ICI的影响。比如，可以通过最大化所有子载波符号总信干噪比(SINR)来获得该准则下最优的积分区间起始位置，即设计起始位置τ的估计为： $\hat{τ} = \underset{τ}{\arg \max} {SINR (τ, C)},$ 其中，C表示信道信息，具体实施时所能够利用的信道信息是接收机获得的信道估计值，即利用完整的信道信息，通过最大化信干噪比的方式实现符号同步，该符号同步方法以下称为动态同步技术。For the existing IBI and ICI, the impact of IBI and ICI caused by insufficient CP length can be minimized by selecting an appropriate starting position of the integration interval. For example, the optimal starting position of the integration interval under this criterion can be obtained by maximizing the total signal-to-interference-noise ratio (SINR) of all subcarrier symbols, that is, the estimation of the design starting position τ is: $\hat{τ} = \underset{τ}{\arg \max} {SINR (τ, C)},$ Among them, C represents channel information. The channel information that can be used in the specific implementation is the channel estimation value obtained by the receiver, that is, using the complete channel information to achieve symbol synchronization by maximizing the signal-to-interference-noise ratio. The symbol synchronization method is as follows Called Dynamic Sync.

在实现本发明过程中，发明人发现现有动态同步技术至少存在如下问题：In the process of realizing the present invention, the inventor found that the existing dynamic synchronization technology has at least the following problems:

在上述动态同步技术中，需要利用信道信息C，即通常需要在频域进行信道估计后才能获得信道估计值，而变换到频域之前，首先需要进行符号同步，这说明信道估计精度又依赖于OFDM符号同步的精度，两者之间相互耦合。如此，在符号同步之前无法获得较理想的信道因子估计，进而导致无法获得最大信干噪比意义上最优的积分起始位置(也称为符号同步位置)，即信道估计误差将直接影响到信干噪比SINR估计的准确性，并进而影响到符号同步性能。In the above-mentioned dynamic synchronization technology, the channel information C needs to be used, that is, the channel estimation value can only be obtained after channel estimation in the frequency domain, and before transforming to the frequency domain, symbol synchronization needs to be performed first, which shows that the channel estimation accuracy depends on The accuracy of OFDM symbol synchronization, and the mutual coupling between the two. In this way, an ideal channel factor estimate cannot be obtained before symbol synchronization, resulting in the inability to obtain the optimal integration starting position (also called symbol synchronization position) in the sense of maximum SINR, that is, the channel estimation error will directly affect the The accuracy of SINR estimation, and then affect the symbol synchronization performance.

另外，在CP不足的情况下，信道估计本身也会受到IBI和ICI的影响，难以保证很高的准确性，进一步降低了动态同步算法的鲁棒性。In addition, in the case of insufficient CP, the channel estimation itself will also be affected by IBI and ICI, and it is difficult to guarantee high accuracy, which further reduces the robustness of the dynamic synchronization algorithm.

同样，除OFDM系统外，上述问题也存在于采用SC-FDE(single carrierfrequency domain equalization，频域均衡的单载波)技术的无线通信系统中。Similarly, in addition to OFDM systems, the above problems also exist in wireless communication systems using SC-FDE (single carrier frequency domain equalization, frequency domain equalized single carrier) technology.

发明内容Contents of the invention

本发明提供了一种宽带无线通信系统中实现符号同步的方法、装置及接收机，以准确确定相应的积分区间的起始位置，进而实现精确的符号同步。The invention provides a method, a device and a receiver for realizing symbol synchronization in a broadband wireless communication system, so as to accurately determine the starting position of a corresponding integration interval, and then realize precise symbol synchronization.

一种宽带无线通信系统中实现同步的方法，包括：A method for realizing synchronization in a broadband wireless communication system, comprising:

获得通信系统中的信道的多径位置和各径功率信息；Obtain the multipath position and power information of each path of the channel in the communication system;

根据所述信道的多径位置和各径功率信息，计算对预设起始位置集合内的快速傅里叶变换FFT积分区间起始位置所对应的所有子载波符号的总信干噪比，其中，所述预设起始位置集合内的FFT积分区间起始位置包含多个；According to the multipath position of the channel and the power information of each path, calculate the total signal-to-interference-noise ratio of all subcarrier symbols corresponding to the start position of the fast Fourier transform FFT integration interval in the preset start position set, wherein , the starting positions of the FFT integration interval in the set of preset starting positions include multiple;

将计算得到的总信干噪比中的最大总信干噪比所对应的积分区间起始位置作为符号定时位置，对接收信号进行同步处理。The starting position of the integration interval corresponding to the maximum total signal-to-interference-noise ratio in the calculated total signal-to-interference-noise-noise ratio is used as the symbol timing position, and the received signal is synchronously processed.

一种宽带无线通信系统中实现同步的装置，包括：A device for realizing synchronization in a broadband wireless communication system, comprising:

信道各径位置和功率获取单元，用于获取通信系统中的信道的多径位置和各径功率信息；The position and power acquisition unit of each path of the channel is used to acquire the multipath position and power information of each path of the channel in the communication system;

总信干噪比计算单元，用于根据所述信道各径位置和功率获取单元获得信道的多径位置和各径功率信息，计算对预设起始位置集合内的FFT积分区间起始位置所对应的所有子载波符号总信干噪比，其中，所述预设起始位置集合内的FFT积分区间起始位置包含多个；The total signal-to-interference-noise ratio calculation unit is used to obtain the multipath position and power information of each path of the channel according to the position of each path of the channel and the power acquisition unit, and calculate the starting position of the FFT integration interval in the preset starting position set The corresponding total signal-to-interference-noise ratio of all subcarrier symbols, wherein, the FFT integration interval starting positions in the preset starting position set include multiple;

符号同步执行单元，用于将所述总信干噪比计算单元计算得到的总信干噪比中的最大总信干噪比所对应的积分区间起始位置作为符号定时位置，对接收信号进行同步处理。The symbol synchronization execution unit is used to use the starting position of the integration interval corresponding to the maximum total signal-to-interference-noise ratio in the total signal-to-interference-noise ratio calculated by the total signal-to-interference and noise ratio calculation unit as the symbol timing position, and perform Synchronous processing.

一种接收机，包括上述宽带无线通信系统中实现同步的装置。A receiver, including the above-mentioned device for realizing synchronization in the broadband wireless communication system.

由上述本发明的实施例提供的技术方案可以看出，其在实现符号同步的过程中，仅需要根据信道多径位置和各径功率信息来计算预置区间内所有FFT积分区间起始位置对应的所有子载波总信干噪比，并将最大总信干噪比对应的起始位置作为符号定时位置，从而使得符号同步不受信道估计误差的影响，在信道时延扩展大于CP长度的场景下，具有很好的性能鲁棒性和简单实用的优势。It can be seen from the technical solutions provided by the above-mentioned embodiments of the present invention that in the process of realizing symbol synchronization, it is only necessary to calculate the correspondence between the starting positions of all FFT integration intervals in the preset interval according to the channel multipath position and the power information of each path. The total signal-to-interference-noise ratio of all subcarriers, and the starting position corresponding to the maximum total signal-to-interference-noise ratio is used as the symbol timing position, so that the symbol synchronization is not affected by the channel estimation error, and in the scenario where the channel delay extension is greater than the CP length It has the advantages of good performance robustness and simplicity and practicality.

附图说明Description of drawings

为了更清楚地说明本发明实施例的技术方案，下面将对实施例描述中所需要使用的附图作简单地介绍，显而易见地，下面描述中的附图仅仅是本发明的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动性的前提下，还可以根据这些附图获得其他的附图。In order to more clearly illustrate the technical solutions of the embodiments of the present invention, the following will briefly introduce the accompanying drawings that need to be used in the description of the embodiments. Obviously, the accompanying drawings in the following description are only some embodiments of the present invention. For Those of ordinary skill in the art can also obtain other drawings based on these drawings without any creative effort.

图1为现有技术中单径信道上不同符号定时位置导致IBI和ICI的示意图；FIG. 1 is a schematic diagram of IBI and ICI caused by different symbol timing positions on a single-path channel in the prior art;

图2为本发明实施例提供的实现同步操作的过程示意图；FIG. 2 is a schematic diagram of a process for realizing a synchronous operation provided by an embodiment of the present invention;

图3为本发明实施例提供的符号同步装置的结构示意图；FIG. 3 is a schematic structural diagram of a symbol synchronization device provided by an embodiment of the present invention;

图4为本发明实施例提供的装置实施例一及其所在系统的示意图；Fig. 4 is a schematic diagram of the device embodiment 1 provided by the embodiment of the present invention and the system in which it is located;

图5为本发明实施例提供的装置实施例二及其所在系统的示意图。Fig. 5 is a schematic diagram of Embodiment 2 of the device provided by the embodiment of the present invention and the system in which it is located.

具体实施方式Detailed ways

下面将结合本发明实施例中的附图，对本发明实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例仅仅是本发明一部分实施例，而不是全部的实施例。基于本发明中的实施例，本领域普通技术人员在没有做出创造性劳动前提下所获得的所有其他实施例，都属于本发明保护的范围。The following will clearly and completely describe the technical solutions in the embodiments of the present invention with reference to the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only some, not all, embodiments of the present invention. Based on the embodiments of the present invention, all other embodiments obtained by persons of ordinary skill in the art without making creative efforts belong to the protection scope of the present invention.

本发明实施例利用信道多径位置结合功率信息实现相应的符号同步，具体可以称为静态同步方法。The embodiments of the present invention utilize channel multipath positions combined with power information to implement corresponding symbol synchronization, which may be specifically referred to as a static synchronization method.

本发明实施例提供的在宽带通信系统中实现符号同步的技术方案中，首先，获得无线信道的多径位置及其功率信息；之后，根据相应信道的多径位置及其功率信息，对预设区间内的所有FFT积分区间起始位置估计对应的所有子载波符号总信干噪比，并将最大总信干噪比所对应的起始位置作为符号同步位置

对接收信号进行符号同步。In the technical solution for realizing symbol synchronization in the broadband communication system provided by the embodiment of the present invention, firstly, the multipath position and power information of the wireless channel are obtained; then, according to the multipath position and power information of the corresponding channel, the preset Estimate the total SINR of all subcarrier symbols corresponding to the starting position of all FFT integration intervals in the interval, and use the starting position corresponding to the maximum total SINR as the symbol synchronization position

Symbol synchronization is performed on the received signal.

下面首先针对CP不足情形下的接收信号模型，推导出FFT积分区间起始位置τ对应的所有子载波符号总信干噪比SINR(τ)估计方法：In the following, for the received signal model in the case of insufficient CP, the estimation method of the total signal-to-interference-noise ratio SINR(τ) of all subcarrier symbols corresponding to the starting position τ of the FFT integration interval is derived:

根据SINR(τ)的定义可得：According to the definition of SINR(τ), we can get:

$SINR SINR ((τ τ)) = = \frac{{P P}_{s the s} ((τ τ))}{{P P}_{ICI ICI} ((τ τ)) + + {P P}_{IBI IBI} ((τ τ)) + + {P P}_{AWGN AWGN}} - - - - - - ((11))$

可见，若要获得SINR(τ)值，则需要具体推导出有用信号的功率P_s(τ)，IBI功率P_IBI(τ)、ICI功率P_ICI(τ)和噪声功率P_AWGN(τ)。It can be seen that to obtain the SINR(τ) value, it is necessary to specifically derive the useful signal power P _s (τ), IBI power P _IBI (τ), ICI power P _ICI (τ) and noise power P _AWGN (τ).

下面对相应的接收信号模型进行分析，以便于获得相应的P_s(τ)、P_IBI(τ)、P_ICI(τ)和P_AWGN(τ)。The corresponding received signal model is analyzed below in order to obtain corresponding P _s (τ), P _IBI (τ), P _ICI (τ) and P _AWGN (τ).

如图1所示，OFDM符号长度为T＝T_s+T_G，其中T_s为FFT积分区间长度，T_G为CP长度。设OFDM调制信号为：As shown in Figure 1, the OFDM symbol length is T=T _s +T _G , where T _s is the length of the FFT integration interval, and T _G is the length of the CP. Let the OFDM modulated signal be:

$x x ((t t)) = = \underset{n no}{Σ Σ} {x x}_{n no} ((t t - - nT n)) - - - - - - ((22))$

其中，n表示OFDM符号序号。基于图1所示的单径接收信号分析不同符号同步τ(积分区间起始位置)下，子载波符号所受干扰的影响。Wherein, n represents the serial number of the OFDM symbol. Based on the single-path received signal shown in FIG. 1 , the impact of interference on the subcarrier symbols under different symbol synchronization τ (starting position of the integration interval) is analyzed.

在第n个OFDM符号周期上进行DFT(Discrete Fourier Transform，离散傅里叶变换)积分可得到对应的子载波符号：DFT (Discrete Fourier Transform, Discrete Fourier Transform) integration is performed on the nth OFDM symbol period to obtain the corresponding subcarrier symbols:

${s the s}_{n no} ((k k,, τ τ)) = = \frac{11}{{T T}_{S S}} {&Integral; &Integral;}_{00}^{{T T}_{S S}} x x ((t t - - nT n + + τ τ)) {e e}^{- - j j 22 π π {f f}_{k k} t t} dt dt - - - - - - ((33))$

其中f_k为第k子载波对应的频率。可见，上式(3)与积分区间起始位置τ有关，在图1所示的3种case下τ的影响是不同的，具体分析可得相应的子载波符号为：Where f _k is the frequency corresponding to the kth subcarrier. It can be seen that the above formula (3) is related to the starting position τ of the integration interval. The influence of τ is different under the three cases shown in Figure 1. The corresponding subcarrier symbols can be obtained by specific analysis:

${s the s}_{n no} ((k k,, τ τ)) = = \{\begin{matrix} {α α}_{00} ((τ τ)) {e e}^{j j 22 π π {f f}_{k k} τ τ} {s the s}_{n no} [[k k]] + + {Σ Σ}_{l l &NotEqual; &NotEqual; 00} {α α}_{11} ((τ τ)) {e e}^{j j 22 π π {f f}_{k k} τ τ} {s the s}_{n no} [[l l + + k k]] + + {e e}_{n no} ((k k,, τ τ)) & ifτ ifτ < < 00 orτ orτ > > {T T}_{G G},, \\ {e e}^{j j 22 π π {f f}_{k k} τ τ} {s the s}_{n no} [[k k]] & if if 00 \leq \leq τ τ \leq \leq {T T}_{G G},, \end{matrix} - - - - - - ((44))$

其中，α_l(τ)表示不同子载波间干扰(ICI)系数，l是对当前子载波造成ICI的子载波和当前子载波的间隔，k是子载波序号，s_n[k]为第n个OFDM符号块中的第k子载波发送符号。与通常的多普勒频偏或者扩展带来的ICI不同的是，该α_l(τ)的取值与τ的值相关，即：Among them, α _l (τ) represents the inter-subcarrier interference (ICI) coefficient, l is the interval between the subcarrier that causes ICI to the current subcarrier and the current subcarrier, k is the sequence number of the subcarrier, and s _n [k] is the nth The k-th subcarrier in an OFDM symbol block transmits symbols. Different from the ICI caused by the usual Doppler frequency offset or extension, the value of α _l (τ) is related to the value of τ, namely:

${α α}_{l l} ((τ τ)) = = \{\begin{matrix} 11 + + \frac{τ τ}{{T T}_{s the s}},, & if l if l = = 00 andτ andτ < < 00,, \\ \frac{{e e}^{j j 22 πlΔfτ πlΔfτ} - - 11}{j j 22 πl πl},, & ifl ifl &NotEqual; &NotEqual; 00 andτ andτ < < 00,, \\ 11 - - \frac{τ τ - - {T T}_{G G}}{{T T}_{s the s}},, & ifl ifl = = 00 andτ andτ > > {T T}_{G G},, \\ \frac{{e e}^{- - j j 22 πlΔf πlΔf {{τ τ - - {T T}_{G G}}}} - - 11}{j j 22 πl πl},, & ifl ifl &NotEqual; &NotEqual; 00 andτ andτ > > {T T}_{G G},, \end{matrix} - - - - - - ((55))$

式(4)中的e_n(k，τ)表示第k子载波符号中由IBI导致的干扰(以下简称IBI分量)：e _n (k, τ) in formula (4) represents the interference caused by IBI in the kth subcarrier symbol (hereinafter referred to as IBI component):

${e e}_{n no} ((k k,, τ τ)) = = \{\begin{matrix} \frac{11}{{T T}_{S S}} {&Integral; &Integral;}_{00}^{- - τ τ} {x x}_{n no - - 11} ((t t + + T T + + τ τ)) {e e}^{- - j j 22 π π {f f}_{k k} t t} dt dt,, & τ τ < < 00,, \\ \frac{11}{{T T}_{S S}} {&Integral; &Integral;}_{T T - - τ τ}^{{T T}_{S S}} {x x}_{n no + + 11} ((t t - - T T + + τ τ)) {e e}^{- - j j 22 π π {f f}_{k k} t t} dt dt,, & τ τ > > {T T}_{G G} \end{matrix} - - - - - - ((66))$

可见，IBI只存在于case II和case III两种场景下，即τ＜0和τ＞T_G下，在case II场景下，相应的IBI来自于前一个OFDM符号，在case III场景下，相应的IBI来自于后一个OFDM符号。It can be seen that IBI only exists in two scenarios of case II and case III, that is, τ<0 and τ>T _G. In case II scenario, the corresponding IBI comes from the previous OFDM symbol. In case III scenario, the corresponding The IBI comes from the latter OFDM symbol.

为了得到大时延扩展信道下的接收信号模型，下面将上述单径情况下的分析扩展到多径信道下：In order to obtain the received signal model under the extended channel with large delay, the analysis in the above single-path case is extended to the multi-path channel as follows:

首先，给出多径无线信道响应为：First, the multipath wireless channel response is given as:

$h h ((t t)) = = \underset{i i}{Σ Σ} {γ γ}_{i i} δ δ ((t t - - {τ τ}_{i i})) - - - - - - ((77))$

其中，τ_i为第i径的时延，γ_i为第i径的衰落因子，δ表示冲击函数。Among them, τ _i is the time delay of the i-th path, γ _i is the fading factor of the i-th path, and δ represents the impact function.

接收信号为信道和发送信号的卷积：The received signal is the convolution of the channel and the transmitted signal:

$y the y ((t t)) = = \underset{i i}{Σ Σ} {γ γ}_{i i} x x ((t t - - {τ τ}_{i i})) + + n no ((t t)) - - - - - - ((88))$

为了进行频域均衡，对当前第n符号的接收信号进行DFT(离散傅里叶变换)，得到频域子载波符号为：In order to perform frequency domain equalization, DFT (discrete Fourier transform) is performed on the received signal of the current nth symbol, and the obtained frequency domain subcarrier symbols are:

${\overset{^^}{s the s}}_{n no} ((k k,, τ τ)) = = \frac{11}{{T T}_{S S}} {&Integral; &Integral;}_{00}^{{T T}_{S S}} u u ((t t - - nT n + + τ τ)) {e e}^{- - j j 22 π π {f f}_{k k} t t} dt dt$

$= = \underset{i i}{Σ Σ} {γ γ}_{i i} ((\frac{11}{{T T}_{S S}} {&Integral; &Integral;}_{00}^{{T T}_{S S}} x x ((t t - - nT n + + τ τ - - {τ τ}_{i i})) {e e}^{- - j j 22 π π {f f}_{k k} t t} dt dt)) + + \overset{^^}{n no} ((k k,, τ τ)),,$

$= = \underset{i i}{Σ Σ} {γ γ}_{i i} {s the s}_{n no} ((k k,, τ τ - - {τ τ}_{i i})) + + \overset{^^}{n no} ((k k,, τ τ)),, - - - - - - ((99))$

其中，

对应着第n个OFDM符号的第k子载波符号，

为噪声部分。可见，信号部分可以表示为多个单径信道频域接收信号部分的叠加，叠加系数为对应径的衰落因子。in,

Corresponding to the kth subcarrier symbol of the nth OFDM symbol,

is the noise part. It can be seen that the signal part can be expressed as the superposition of multiple single-path channel frequency domain received signal parts, and the superposition coefficient is the fading factor of the corresponding path.

进一步，由(4)可得：Further, from (4) we can get:

${\overset{^^}{s the s}}_{n no} ((k k,, τ τ)) = = ((\underset{{τ τ}_{i i} > > τor τor < < τ τ - - {T T}_{G G}}{Σ Σ} {γ γ}_{i i} {α α}_{00} ((τ τ - - {τ τ}_{i i})) {e e}^{- - j j 22 π π {f f}_{k k} {τ τ}_{i i}} + + \underset{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ}{Σ Σ} {γ γ}_{i i} {e e}^{- - j j 22 π π {f f}_{k k} {τ τ}_{i i}})) {e e}^{j j 22 π π {f f}_{k k} τ τ} {s the s}_{n no} [[k k]]$

$+ + \underset{l l &NotEqual; &NotEqual; 00}{Σ Σ} ((\underset{{τ τ}_{i i} > > τor τor < < τ τ - - {T T}_{G G}}{Σ Σ} {γ γ}_{i i} {α α}_{l l} ((τ τ - - {τ τ}_{i i})) {e e}^{j j 22 π π {f f}_{k k} ((τ τ - - {τ τ}_{i i}))})) {s the s}_{n no} [[l l + + k k]]$

$+ + \underset{{τ τ}_{i i} > > τor τor < < τ τ - - {T T}_{G G}}{Σ Σ} {γ γ}_{i i} {e e}_{n no} ((k k,, τ τ - - {τ τ}_{i i})) + + \overset{^^}{n no} ((k k,, τ τ))$

$= = H h (({f f}_{k k},, τ τ)) {e e}^{j j 22 π π {f f}_{k k} τ τ} {s the s}_{n no} [[k k]] + + \underset{l l &NotEqual; &NotEqual; 00}{Σ Σ} {I I}_{l l} (({f f}_{k k},, τ τ)) {s the s}_{n no} [[l l + + k k]] + + {\overset{^^}{e e}}_{n no} ((k k,, τ τ)) + + \overset{^^}{n no} ((k k,, τ τ)),, - - - - - - ((1010))$

上式共有四项，第一项为有用信号部分，第二项ICI部分，第三项为IBI部分，第四项为噪声部分，由推导过程可知：There are four items in the above formula, the first item is the useful signal part, the second item is the ICI part, the third item is the IBI part, and the fourth item is the noise part, it can be known from the derivation process:

其中有用信号部分的频域信道因子为：The frequency domain channel factor of the useful signal part is:

$H h (({f f}_{k k},, τ τ)) = = \underset{{τ τ}_{i i} > > τor τor < < τ τ - - {T T}_{G G}}{Σ Σ} {γ γ}_{i i} {α α}_{00} ((τ τ - - {τ τ}_{i i})) {e e}^{- - j j 22 π π {f f}_{k k} {τ τ}_{i i}} + + \underset{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ}{Σ Σ} {γ γ}_{i i} {e e}^{- - j j 22 π π {f f}_{k k} {τ τ}_{i i}};;$

ICI系数为：The ICI coefficient is:

${I I}_{l l} (({f f}_{k k},, τ τ)) = = \underset{{τ τ}_{i i} > > τor τor < < τ τ - - {T T}_{G G}}{Σ Σ} {γ γ}_{i i} {α α}_{l l} ((τ τ - - {τ τ}_{i i})) {e e}^{j j 22 π π {f f}_{k k} ((τ τ - - {τ τ}_{i i}))};;$

表示与第k子载波符号相距l的子载波符号带来的ICI系数。Indicates the ICI coefficient brought by the subcarrier symbol that is l away from the kth subcarrier symbol.

IBI功率为：The IBI power is:

${\overset{^^}{e e}}_{n no} ((k k,, τ τ)) = = \underset{{τ τ}_{i i} > > τor τor < < τ τ - - {T T}_{G G}}{Σ Σ} {γ γ}_{i i} {e e}_{n no} ((k k,, τ τ - - {τ τ}_{i i}));;$

公式(10)中的其它变量可参见之前的推导过程。基于以上四项，可以分别估计得到所有子载波符号总功率P_s(τ)(即有用信号的总功率)，当前OFDM符号块所受IBI总功率P_IBI(τ)、所有子载波符号所受ICI功率P_ICI(τ)和所有子载波符号噪声总功率P_AWGN(τ)。Other variables in formula (10) can refer to the previous derivation process. Based on the above four items, the total power P _s (τ) of all subcarrier symbols (that is, the total power of the useful signal), the total power P IBI (τ) of the current OFDM symbol block, and the total power P _IBI (τ) of all subcarrier symbols can be estimated respectively. The ICI power P _ICI (τ) and the total symbol noise power of all subcarriers P _AWGN (τ).

与现有技术不同的是，本发明实施例中采用平均功率来代替瞬时功率，从而在计算P_s(τ)、P_IBI(τ)和P_ICI(τ)中，可以避免采用瞬时的各径信道因子。具体实现时，可以通过采用E|γ_i|²代替|γ_i|²实现的。而一般根据无线信道的基本特性，可以假设各径信道因子(或者各径信道响应)γ_i是相互独立的，且满足零均值和方差 $σ_{i}^{2} = E {| γ_{i} |}^{2},$ 其中的E表示对|γ_i|²(各径信道因子模平方)的求期望运算，由(10)式可得所有子载波符号总功率P_s(τ)为：Different from the prior art, in the embodiment of the present invention, the average power is used instead of the instantaneous power, so that in the calculation of P _s (τ), P _IBI (τ) and P _ICI (τ), it is possible to avoid using the instantaneous channel factor. In specific implementation, it can be realized by using E|γ _i | ² instead of |γ _i | ² . Generally, according to the basic characteristics of wireless channels, it can be assumed that the channel factors (or channel responses) γ _i of each path are independent of each other and satisfy zero mean and variance $σ_{i}^{2} = E. {| γ_{i} |}^{2},$ where E represents the expectation operation for |γ _i | ² (the square of the channel factor modulus of each path), and the total power P _s (τ) of all subcarrier symbols can be obtained from formula (10):

${P P}_{s the s} ((τ τ)) = = {E E.}_{C C} {{{P P}_{s the s} ((τ τ,, C C))}} = = {σ σ}_{s the s}^{22} \underset{k k}{Σ Σ} {E E.}_{C C} {{{| | H h (({f f}_{k k} - - τ τ)) | |}^{22}}}$

$= = {σ σ}_{s the s}^{22} \underset{k k}{Σ Σ} ((\underset{{τ τ}_{i i} > > τor τor < < τ τ - - {T T}_{G G}}{Σ Σ} E E. {{{| | {γ γ}_{i i} | |}^{22}}} {| | {α α}_{00} ((τ τ - - {τ τ}_{i i})) | |}^{22} + + \underset{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ}{Σ Σ} E E. {{{| | {γ γ}_{i i} | |}^{22}}}))$

$= = K K {σ σ}_{s the s}^{22} ((\underset{{τ τ}_{i i} > > τ τ}{Σ Σ} {σ σ}_{i i}^{22} {((11 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}}))}^{22} + + \underset{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ}{Σ Σ} {σ σ}_{i i}^{22} + + \underset{{τ τ}_{i i} < < τ τ - - {T T}_{G G}}{Σ Σ} {σ σ}_{i i}^{22} {((11 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}}))}^{22})) - - - - - - ((1111))$

其中，E_C{·}表示对随机信道因子求期望(进行求期望运算)，σ_s ²为发送符号功率，σ_i ²为第i径的功率。基于推导可知，通过对信道各径信道因子(或者称各径信道响应)的求期望处理，避免了现有技术中需要利用信道因子进行运算所带来的缺陷。Among them, E _C {·} represents the expectation of the random channel factor (perform the expectation operation), σ _s ² is the transmitted symbol power, and σ _i ² is the power of the i-th path. Based on the derivation, it can be seen that through the expectation processing of channel factors (or channel responses of each path) of each path of the channel, the defects caused by the need to use channel factors for calculation in the prior art are avoided.

对IBI和ICI干扰项进行类似的处理可得：Similar treatment of IBI and ICI interference terms can be obtained:

${P P}_{IBI IBI} ((τ τ)) = = {E E.}_{C C} {{{P P}_{IBI IBI} ((τ τ,, C C))}}$

$= = \underset{k k}{Σ Σ} \underset{{τ τ}_{i i} > > τ τ}{Σ Σ} {E E.}_{C C} {{{| | {γ γ}_{i i} | |}^{22}}} E E. {{{| | {e e}_{n no} ((k k,, τ τ - - {τ τ}_{i i})) | |}^{22}}} + + \underset{k k}{Σ Σ} \underset{{τ τ}_{i i} < < τ τ - - {T T}_{G G}}{Σ Σ} {E E.}_{C C} {{{| | {γ γ}_{i i} | |}^{22}}} E E. {{{| | {e e}_{n no} ((k k,, τ τ - - {τ τ}_{i i})) | |}^{22}}}$

$= = K K {σ σ}_{s the s}^{22} ((\underset{{τ τ}_{i i} > > τ τ}{Σ Σ} {σ σ}_{i i}^{22} \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}} + + \underset{{τ τ}_{i i} < < τ τ - - {T T}_{G G}}{Σ Σ} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}})),, - - - - - - ((1212))$

${P P}_{ICI ICI} ((τ τ)) = = {E E.}_{C C} {{{P P}_{ICI ICI} ((τ τ,, C C))}} = = K K {σ σ}_{s the s}^{22} \underset{l l &NotEqual; &NotEqual; 00}{Σ Σ} {E E.}_{C C} {{{| | {I I}_{l l} (({f f}_{k k},, τ τ)) | |}^{22}}}$

$= = K K {σ σ}_{s the s}^{22} \underset{l l &NotEqual; &NotEqual; 00}{Σ Σ} \underset{{τ τ}_{i i} > > τor τor < < τ τ - - {T T}_{G G}}{Σ Σ} {E E.}_{C C} {{{| | {γ γ}_{i i} | |}^{22}}} {| | {α α}_{l l} ((τ τ - - {τ τ}_{i i})) | |}^{22}$

$= = K K {σ σ}_{s the s}^{22} ((\underset{{τ τ}_{i i} > > τ τ}{Σ Σ} {σ σ}_{i i}^{22} \underset{l l &NotEqual; &NotEqual; 00}{Σ Σ} {| | \frac{{e e}^{j j 22 πlΔf πlΔf (({τ τ}_{i i} - - τ τ))}}{j j 22 πl πl} | |}^{22} + + \underset{{τ τ}_{i i} > > τ τ - - {T T}_{G G}}{Σ Σ} {σ σ}_{i i}^{22} \underset{l l &NotEqual; &NotEqual; 00}{Σ Σ} {| | \frac{{e e}^{- - j j 22 πlΔf πlΔf ((τ τ - - {τ τ}_{i i} - - {T T}_{G G}))} - - 11}{j j 22 πl πl} | |}^{22}))$

$= = K K {σ σ}_{s the s}^{22} ((\underset{{τ τ}_{i i} > > τ τ}{Σ Σ} {σ σ}_{i i}^{22} \underset{l l &NotEqual; &NotEqual; 00}{Σ Σ} \frac{{sin sin}^{22} ((πlΔf πlΔf (({τ τ}_{i i} - - ι ι))))}{{((πl πl))}^{22}} + + \underset{{τ τ}_{i i} < < τ τ - - {T T}_{G G}}{Σ Σ} {σ σ}_{i i}^{22} \underset{l l &NotEqual; &NotEqual; 00}{Σ Σ} \frac{{sin sin}^{22} ((πlΔf πlΔf ((τ τ - - {τ τ}_{i i} - - {T T}_{G G}))))}{{((πl πl))}^{22}}))$

$= = K K {σ σ}_{s the s}^{22} ((\underset{{τ τ}_{i i} > > τ τ}{Σ Σ} {σ σ}_{i i}^{22} \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}} ((11 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}})) + + \underset{{τ τ}_{i i} < < τ τ - - {T T}_{G G}}{Σ Σ} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}} ((11 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}})))) - - - - - - ((1313))$

最后分析一下噪声功率，每个子载波上的噪声功率为：Finally, analyze the noise power. The noise power on each subcarrier is:

$E E. {| | \overset{^^}{n no} ((k k,, τ τ)) | |}^{22} = = E E. {| | \frac{11}{{T T}_{S S}} {&Integral; &Integral;}_{00}^{{T T}_{S S}} n no ((t t - - nT n + + τ τ)) {e e}^{- - j j 22 π π {f f}_{k k} t t} dt dt | |}^{22} = = \frac{{N N}_{o o}}{{T T}_{S S}} - - - - - - ((1414))$

故，所有子载波符号噪声总功率为：Therefore, the total symbol noise power of all subcarriers is:

${P P}_{AWGN AWGN} = = \underset{k k}{Σ Σ} E E. {| | \overset{^^}{n no} ((k k,, τ τ)) | |}^{22} = = \frac{{KN KN}_{o o}}{{T T}_{S S}} - - - - - - ((1515))$

综上可得频域K个子载波上的基于平均功率获得的总信干噪比为：In summary, the total signal-to-interference-noise ratio obtained based on the average power on the K subcarriers in the frequency domain is:

$SINR SINR ((τ τ)) = = \frac{{P P}_{88} ((τ τ))}{{P P}_{ICI ICI} ((τ τ)) + + {P P}_{IBI IBI} ((τ τ)) + + {P P}_{AWGN AWGN}}$

$= = \frac{K K {σ σ}_{s the s}^{22} {Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} {((11 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}}))}^{22} + + {Σ Σ}_{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ} {σ σ}_{i i}^{22} + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} {((11 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}}))}^{22}}{K K {σ σ}_{s the s}^{22} (({Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}} ((22 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}})) + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}} ((22 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}})))) + + \frac{{KN KN}_{o o}}{{T T}_{S S}}}$

$= = \frac{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} {((11 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}}))}^{22} + + {Σ Σ}_{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ} {σ σ}_{i i}^{22} + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} {((11 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}}))}^{22}}{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}} ((22 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}})) + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}} ((22 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}})) + + \frac{11}{ρ ρ}},, - - - - - - ((1616))$

其中 $ρ = \frac{T_{s} σ_{s}^{2}}{N_{0}}$ 为频域子载波---xxxx符号信噪比，N₀为噪声功率谱密度。in $ρ = \frac{T_{the s} σ_{the s}^{2}}{N_{0}}$ is the frequency-domain subcarrier --- xxxx symbol signal-to-noise ratio, and N ₀ is the noise power spectral density.

以上推导适用于接收机没有采用基于块判决反馈的IBI消除方法。The above derivation is applicable to the case where the receiver does not adopt the IBI cancellation method based on block decision feedback.

针对采用块判决反馈进行IBI消除的接收机，下面分析其SINR(τ)的估计方法：For a receiver that uses block decision feedback for IBI cancellation, the estimation method of its SINR(τ) is analyzed as follows:

假设通过块判决反馈IBI消除处理，可以完全消除前一个符号对当前符号的IBI，则公式(12)中相应地去掉该部分IBI对应的项，可得：Assuming that the IBI of the previous symbol to the current symbol can be completely eliminated through the block decision feedback IBI elimination process, then the item corresponding to this part of the IBI is correspondingly removed in formula (12), and it can be obtained:

${\overset{&OverBar; &OverBar;}{P P}}_{IBI IBI} ((τ τ)) = = K K {σ σ}_{s the s}^{22} \underset{{τ τ}_{i i} < < τ τ - - {T T}_{G G}}{Σ Σ} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}} - - - - - - ((1717))$

此时，频域K个子载波上的基于平均功率获得的总信干噪比为：At this time, the total SINR obtained based on the average power on K subcarriers in the frequency domain is:

$\overset{&OverBar; &OverBar;}{SINR SINR} ((τ τ)) = = \frac{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} {((11 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}}))}^{22} + + {Σ Σ}_{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ} {σ σ}_{i i}^{22} + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} {((11 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}}))}^{22}}{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}} ((22 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}})) + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}} ((22 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}})) + + \frac{11}{ρ ρ}} - - - - - - ((1818))$

将区间[-τ_min，τ_max]上最大的SINR(τ)或者最大SINR(τ)(取决于接收机是否进行了基于块判决反馈的IBI消除)对应的τ作为符号定时

即：Use τ corresponding to the maximum SINR(τ) or the maximum SINR(τ) (depending on whether the receiver performs IBI cancellation based on block decision feedback) on the interval [-τ _min , τ _max ] as the symbol timing

Right now:

$\hat{τ} = \underset{τ}{\arg \max} {SINR (τ)}$ 或者 $\hat{τ} = \underset{τ}{\arg \max} {\overset{&OverBar;}{SINR} (τ)} - - - (19)$ $\hat{τ} = \underset{τ}{\arg \max} {SINR (τ)}$ or $\hat{τ} = \underset{τ}{\arg \max} {\overset{&OverBar;}{SINR} (τ)} - - - (19)$

可见，计算干扰的平均功率时，可能存在两种情况：情况a，接收机不进行基于判决反馈的块间干扰IBI消除，所有子载波符号的总信干噪比可以称为第一总信干噪比；情况b，接收机进行基于判决反馈的块间干扰IBI消除，所有子载波符号的总信干噪比为第二总信干噪比。It can be seen that when calculating the average power of interference, there may be two cases: Case a, the receiver does not perform inter-block interference IBI cancellation based on decision feedback, and the total SINR of all subcarrier symbols can be called the first total SINR Noise ratio; in case b, the receiver performs IBI elimination based on decision feedback, and the total SINR of all subcarrier symbols is the second total SINR.

而具体实施过程中，可以将[-τ_min，τ_max]按照所需要的符号定时分辨率量化成有限个符号同步位置{τ_m}_m集合(或称为离散化处理)，然后分别按照(16)式计算对应的SINR(τ_m)或按照(18)式计算对应的SINR(τ_m)，然后将最大的SINR(τ_m)或者最大SINR(τ)所对应的τ_m作为符号定时

即：In the specific implementation process, [-τ _min , τ _max ] can be quantized into a limited set of symbol synchronization positions {τ _m } _m according to the required symbol timing resolution (or discretization processing), and then according to ( Formula 16) calculates the corresponding SINR(τ _m ) or calculates the corresponding SINR(τ _m ) according to formula (18), and then takes the maximum SINR(τ _m ) or the τ _m corresponding to the maximum SINR(τ) as the symbol timing

Right now:

$\hat{τ} = \underset{τ_{m}}{\arg \max} {SINR (τ_{m})}$ 或者 $\hat{τ} = \underset{τ_{m}}{\arg \max} {\overset{&OverBar;}{SINR} (τ_{m})} - - - (20)$ $\hat{τ} = \underset{τ_{m}}{\arg \max} {SINR (τ_{m})}$ or $\hat{τ} = \underset{τ_{m}}{\arg \max} {\overset{&OverBar;}{SINR} (τ_{m})} - - - (20)$

本领域普通技术人员可以在上述推导的基础上，进一步对SINR(τ)或者SINR(τ)的计算公式进行化简，比如对公式(16)，可以基于简单的数学原理，化简为：Those of ordinary skill in the art can further simplify the calculation formula of SINR(τ) or SINR(τ) on the basis of the above derivation. For example, formula (16) can be simplified based on simple mathematical principles:

$SINR SINR ((τ τ)) = = \frac{{Σ Σ}_{iτ iτ} {σ σ}_{i i}^{22} + + \frac{11}{ρ ρ}}{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}} ((22 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}})) + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}} ((22 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}})) + + \frac{11}{ρ ρ}} - - 11 - - - - - - ((21 twenty one))$

类似简单化简或变形均被包括在本发明要求保护的范畴之内。Similar simplifications or modifications are included in the scope of protection of the present invention.

为便于对本发明实施例的理解，下面将结合附图对本发明实施例的具体应用过程进行详细说明。In order to facilitate the understanding of the embodiments of the present invention, the specific application process of the embodiments of the present invention will be described in detail below in conjunction with the accompanying drawings.

如图2A所示，本发明实施例提供的静态同步方案在具体应用过程中，可以包括以下步骤：As shown in Figure 2A, the static synchronization solution provided by the embodiment of the present invention may include the following steps in a specific application process:

步骤21，获得信道的各径位置(即多径位置)和各径功率信息{τ_i，σ_i ²}；Step 21, obtain the position of each path of the channel (that is, the position of multipath) and the power information of each path {τ _i , σ _i ² };

本步骤中，可以通过某种多径搜索或者跟踪方法获取信道的各径位置和各径功率信息。In this step, the position and power information of each path of the channel may be obtained through a certain multipath search or tracking method.

步骤22，基于获取的信道的各径位置和各径功率信息{τ_i，σ_i ²}，计算预设起始位置集合内的FFT积分区间起始位置所对应的所有子载波符号的总信干噪比，其中，预设起始位置集合内的FFT积分区间起始位置包含多个；Step 22, based on the obtained channel position and power information {τ _i , σ _i ² } of each path, calculate the total signal of all subcarrier symbols corresponding to the starting position of the FFT integration interval in the preset starting position set Interference-to-noise ratio, wherein the FFT integration interval starting positions in the preset starting position set include multiple;

例如，可以对预设区间[-τ_min，τ_max]对应集合中的所有τ，计算对应的所有子载波符号总信干噪比SINR(τ)。For example, for all τ in the set corresponding to the preset interval [-τ _min , τ _max ], the corresponding total signal-to-interference-noise ratio SINR(τ) of all subcarrier symbols may be calculated.

可以理解的是，在总信干噪比计算中，需要使用有用信号功率、干扰和噪声功率。本发明实施例中，可以采用平均功率代替现有技术中采用的瞬时功率，这样，可以进一步等效为利用信道各径信道因子的平均功率(对应着各径信道因子模平方的期望)，以代替信道因子本身来进行总信干噪比计算。其中，所谓“期望”，也可以理解为期望运算值。It can be understood that in the calculation of the total SINR, useful signal power, interference and noise power need to be used. In the embodiment of the present invention, the average power can be used to replace the instantaneous power used in the prior art. In this way, it can be further equivalent to using the average power of the channel factor of each path of the channel (corresponding to the expectation of the square of the channel factor of each path), with The total SINR calculation is performed instead of the channel factor itself. Wherein, the so-called "expectation" can also be understood as an expected operation value.

具体可以包括：利用所述信道的多径位置和各径功率信息，计算有用信号的平均功率、干扰的平均功率和噪声的平均功率；根据计算得到的所述有用信号的平均功率、干扰的平均功率和噪声的平均功率，计算对应的各子载波符号的总信干噪比。Specifically, it may include: calculating the average power of the useful signal, the average power of the interference, and the average power of the noise by using the multipath position and power information of each path of the channel; Calculate the total signal-to-interference-noise ratio of each subcarrier symbol corresponding to the average power of power and noise.

根据前述实施例的推导过程，相应的，频域K个子载波符号总信干噪比计算公式为：According to the derivation process of the foregoing embodiment, correspondingly, the calculation formula of the total signal-to-interference-noise ratio of K subcarrier symbols in the frequency domain is:

$SINR SINR ((τ τ)) = = \frac{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} {((11 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}}))}^{22} + + {Σ Σ}_{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ} {σ σ}_{i i}^{22} + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} {((11 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}}))}^{22}}{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}} ((22 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}})) + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}} ((22 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}})) + + \frac{11}{ρ ρ}} - - - - - - ((22 twenty two))$

或者or

$\overset{&OverBar; &OverBar;}{SINR SINR} ((τ τ)) = = \frac{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} {((11 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}}))}^{22} + + {Σ Σ}_{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ} {σ σ}_{i i}^{22} + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} {((11 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}}))}^{22}}{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}} ((22 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}})) + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}} ((22 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}})) + + \frac{11}{ρ ρ}} - - - - - - ((23 twenty three))$

如前所述，采用哪个公式取决于接收机是否进行了基于块判决反馈的IBI消除，没有采用则按照第一个公式，否则采用第二个公式。As mentioned above, which formula to use depends on whether the receiver has performed IBI cancellation based on block decision feedback, and if it is not used, then the first formula is used; otherwise, the second formula is used.

对于预设区间[-τ_min，τ_max]，具体实施中，可以将区间离散化。For the preset interval [-τ _min , τ _max ], in specific implementation, the interval can be discretized.

步骤23，在计算所得的所有SINR(τ)或SINR(τ)中搜索最大值，将最大的SINR(τ)或者SINR(τ)对应的τ作为符号定时位置

即FFT积分区间的起始位置的；Step 23, search for the maximum value among all calculated SINR(τ) or SINR(τ), and use the maximum SINR(τ) or τ corresponding to SINR(τ) as the symbol timing position

That is, the starting position of the FFT integration interval;

具体地，符号定时位置

估计值可以表示为：

\hat{τ} = \underset{τ}{\arg \max} {SINR (τ)}

或者

\hat{τ} = \underset{τ}{\arg \max} {\overset{&OverBar;}{SINR} (τ)} .

Specifically, the symbol timing position

Estimates can be expressed as:

\hat{τ} = \underset{τ}{\arg \max} {SINR (τ)}

or

\hat{τ} = \underset{τ}{\arg \max} {\overset{&OverBar;}{SINR} (τ)} .

步骤24，根据获得符号定时位置

执行接收机中的符号同步操作。Step 24, obtain the symbol timing position according to

Perform symbol synchronization operations in the receiver.

本步骤具体实现时，即根据定时位置(定时估计值)

确定FFT区间的起始位置。When this step is specifically implemented, that is, according to the timing position (timing estimated value)

Determine the starting position of the FFT interval.

本发明实施例提供的同步方案可以但不限于应用于多载波或者采用频域均衡的单载波系统中，如OFDM或SC-FDE系统等。在长时延信道导致CP长度不足的场景下，本发明实施例可以通过选择最优的符号定时位置，来最大化接收信号频域所有子载波总信干噪比，获得比采用通常的假设CP足够长的同步技术更优的接收性能。同时，与背景技术中所介绍的现有技术方案相比，本发明实施例无需利用信道因子本身进行运算(或者说信道响应)，可以避免信道估计误差对符号同步过程的不利影响或对特定系统接收机在符号同步前信道因子不可得的困难，从而更为简单实用，鲁棒性能更好。The synchronization solution provided by the embodiments of the present invention may be applied to, but not limited to, multi-carrier or single-carrier systems using frequency domain equalization, such as OFDM or SC-FDE systems. In the scenario where the length of the CP is insufficient due to a long-delay channel, the embodiment of the present invention can maximize the total SINR of all subcarriers in the frequency domain of the received signal by selecting the optimal symbol timing position, and obtain the Sufficiently long synchronization technology for better reception performance. At the same time, compared with the prior art solutions introduced in the background art, the embodiment of the present invention does not need to use the channel factor itself for calculation (or channel response), which can avoid the adverse impact of channel estimation errors on the symbol synchronization process or the specific system It is difficult for the receiver to obtain the channel factor before the symbol synchronization, so it is simpler and more practical, and the robust performance is better.

本领域普通技术人员可以理解实现上述实施例方法中的全部或部分流程，是可以通过计算机程序来指令相关的硬件来完成，所述的程序可存储于一计算机可读取存储介质中，该程序在执行时，可包括如上述各方法的实施例的流程。其中，所述的存储介质可为磁碟、光盘、只读存储记忆体(Read-Only Memory，ROM)或随机存储记忆体(Random Access Memory，RAM)等。Those of ordinary skill in the art can understand that all or part of the processes in the methods of the above embodiments can be implemented through computer programs to instruct related hardware, and the programs can be stored in a computer-readable storage medium. During execution, it may include the processes of the embodiments of the above-mentioned methods. Wherein, the storage medium may be a magnetic disk, an optical disk, a read-only memory (Read-Only Memory, ROM) or a random access memory (Random Access Memory, RAM), etc.

本发明实施例还提供了一种宽带无线通信系统中实现同步的装置，其具体实现结构如图3所示，可以包括：The embodiment of the present invention also provides a device for implementing synchronization in a broadband wireless communication system, and its specific implementation structure is shown in Figure 3, which may include:

(1)信道各径位置和功率获取单元301，用于获取通信系统中的信道的多径位置和功率信息。上述的信道的多径位置和功率信息，可以理解为信道的部分信息。(1) Channel position and power acquisition unit 301, configured to acquire multipath position and power information of channels in the communication system. The above-mentioned multipath position and power information of the channel may be understood as partial information of the channel.

(2)总信干噪比计算单元302，用于根据上述信道各径位置和功率获取单元301获得信道的多径位置和各径功率信息，计算对预设起始位置集合内的FFT积分区间起始位置所对应的所有子载波符号总信干噪比，其中，所述预设起始位置集合内的FFT积分区间起始位置包含多个；进一步地，该总信干噪比计算单元302在计算所述总信干噪比过程中，具体可以利用信道各径信道因子的平均功率代替信道因子本身进行信干噪比计算，其中，信道各径信道因子的平均功率对应各径信道因子模平方的期望运算值；(2) The total signal-to-interference-noise ratio calculation unit 302 is used to obtain the multipath position of the channel and the power information of each path according to the above-mentioned channel position and power acquisition unit 301, and calculate the FFT integration interval in the preset starting position set The total signal to interference and noise ratio of all subcarrier symbols corresponding to the starting position, wherein, the starting position of the FFT integration interval in the preset starting position set includes multiple; further, the total signal to interference and noise ratio calculation unit 302 In the process of calculating the total SINR, the average power of the channel factors of each path of the channel can be used instead of the channel factor itself to calculate the SINR, wherein the average power of the channel factors of each path of the channel corresponds to the channel factor modulus of each path expected value of squared;

相应的总信干噪比计算单元302具体用于根据无线信道的多径位置和多径功率信息{τ_i，σ_i ²}，对预设起始位置集合[-τ_min，τ_max]或者量化后的起始位置集合中的FFT积分区间起始位置τ计算对应的第一总信干噪比或第二总信干噪比；其中，所述第一总信干噪比为接收机没有进行基于判决反馈的块间干扰IBI消除时计算获得的所有子载波符号总信干噪比，所述第二总信干噪比为接收机进行了基于判决反馈的IBI消除时计算获得的所有子载波符号总信干噪比； ^The corresponding total signal-to-interference-noise ratio calculation _unit 302 is specifically configured to calculate the preset starting position set [ _{-τ min} _, τ _max ] or Calculate the corresponding first total signal-to-interference-noise ratio or the second total signal-to-interference-noise ratio for the FFT integration interval starting position τ in the quantized starting position set; wherein, the first total signal-to-interference-noise ratio is that the receiver does not The total signal-to-interference-noise ratio of all subcarrier symbols calculated when the decision feedback-based inter-block interference IBI cancellation is performed, and the second total signal-to-interference-noise ratio is the total signal-to-interference-noise ratio of all subcarrier symbols calculated when the receiver performs decision feedback-based IBI cancellation The total signal-to-interference-noise ratio of the carrier symbol;

上述第一总信干噪比的计算公式具体可以为：The calculation formula of the above-mentioned first total signal-to-interference-noise ratio may specifically be:

$SINR SINR ((τ τ)) = = \frac{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} {((11 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}}))}^{22} + + {Σ Σ}_{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ} {σ σ}_{i i}^{22} + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} {((11 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}}))}^{22}}{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}} ((22 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}})) + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}} ((22 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}})) + + \frac{11}{ρ ρ}};;$

上述第二总信干噪比的计算公式具体可以为：The calculation formula of the above-mentioned second total signal-to-interference-noise ratio may specifically be:

$\overset{&OverBar; &OverBar;}{SINR SINR} ((τ τ)) = = \frac{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} {((11 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}}))}^{22} + + {Σ Σ}_{τ τ - - {T T}_{G G} \leq \leq {τ τ}_{i i} \leq \leq τ τ} {σ σ}_{i i}^{22} + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} {((11 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}}))}^{22}}{{Σ Σ}_{{τ τ}_{i i} > > τ τ} {σ σ}_{i i}^{22} \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}} ((22 - - \frac{{τ τ}_{i i} - - τ τ}{{T T}_{S S}})) + + {Σ Σ}_{{τ τ}_{i i} < < τ τ - - {T T}_{G G}} {σ σ}_{i i}^{22} \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}} ((22 - - \frac{τ τ - - {τ τ}_{i i} - - {T T}_{G G}}{{T T}_{S S}})) + + \frac{11}{ρ ρ}},,$

其中，在上述两公式中，τ_i为第i径的时延位置，τ为在预设区间内的所有需搜索的FFT积分区间起始位置，σ_i ²为第i径的功率，T_s为多载波系统或者采用频域均衡的单载波系统符号周期长度，T_G为循环前缀CP长度，ρ为频域子载波符号信噪比。Among them, in the above two formulas, τ _i is the time delay position of the i-th path, τ is the starting position of all the FFT integration intervals to be searched in the preset interval, σ _i ² is the power of the i-th path, T _s is the symbol period length of a multi-carrier system or a single-carrier system using frequency-domain equalization, T _G is the CP length of the cyclic prefix, and ρ is the signal-to-noise ratio of sub-carrier symbols in the frequency domain.

(3)符号同步执行单元303，用于将上述总信干噪比计算单元302计算得到的总信干噪比中的最大总信干噪比所对应的积分区间起始位置作为符号定时位置，对接收信号进行同步处理；(3) The symbol synchronization execution unit 303 is used to use the starting position of the integration interval corresponding to the maximum total SINR in the total SINR calculated by the above-mentioned total SINR calculation unit 302 as the symbol timing position, Synchronize the received signal;

对应的，在符号同步执行单元303中，具体可以在上述总信干噪比计算单元302计算得到的SINR(τ)或SINR(τ)中搜索最大值，并将最大的SINR(τ)或者SINR(τ)对应的τ作为所述符号定时

，即相应的符号定时位置

估计值为：Correspondingly, in the symbol synchronization execution unit 303, specifically, the maximum value may be searched in the SINR(τ) or SINR(τ) calculated by the above-mentioned total SINR calculation unit 302, and the maximum SINR(τ) or SINR (τ) corresponds to τ as the symbol timing

, that is, the corresponding symbol timing position

The estimated value is:

$\hat{τ} = \underset{τ}{\arg \max} {SINR (τ)}$ 或者 $\hat{τ} = \underset{τ}{\arg \max} {\overset{&OverBar;}{SINR} (τ)} .$ $\hat{τ} = \underset{τ}{\arg \max} {SINR (τ)}$ or $\hat{τ} = \underset{τ}{\arg \max} {\overset{&OverBar;}{SINR} (τ)} .$

在该装置中，各单元具体采用的处理过程前面方法实施例中已经描述，在此不再详述。In the device, the processing procedures specifically adopted by each unit have been described in the previous method embodiments, and will not be described in detail here.

可选地，本发明实施例中，相应的宽带无线通信系统可以包括多载波通信系统或采用频域均衡的单载波通信系统等。Optionally, in this embodiment of the present invention, the corresponding broadband wireless communication system may include a multi-carrier communication system or a single-carrier communication system using frequency domain equalization, or the like.

进一步地，本发明实施例还提供了一种接收机，其具体结构可以包括上述宽带无线通信系统中实现同步的装置。Further, an embodiment of the present invention also provides a receiver, and its specific structure may include the above-mentioned device for implementing synchronization in the broadband wireless communication system.

在本发明实施例中，提供了可以在CP长度不足时，通过符号同步来最大化所有频域子载波总信干噪比的技术方案，或者称为通过最大化所有频域子载波总的信干噪比来进行符号同步的技术方案，分别对预设区间内的所有积分区间起始位置估计对应的总的信干噪比，然后取最大信干噪比所对应的起始位置作为符号同步位置

在估计总信干噪比的过程中，只需要利用无线信道各径位置信息即可，而无需利用完整的信道信息。而且，相应的无线信道各径位置信息可以由信道估计模块或专门的信道径位置搜索模块来提供，从而使得本发明实施例的应用更加容易。In the embodiment of the present invention, a technical solution is provided that can maximize the total SINR of all frequency-domain subcarriers through symbol synchronization when the CP length is insufficient, or it is called maximizing the total signal-to-interference-noise ratio of all frequency-domain subcarriers. The technical solution for symbol synchronization based on the interference-to-noise ratio is to estimate the corresponding total signal-to-interference-noise ratio for the starting positions of all integration intervals in the preset interval, and then take the starting position corresponding to the maximum signal-to-interference-noise ratio as the symbol synchronization Location

In the process of estimating the total signal-to-interference-noise ratio, only the position information of each path of the wireless channel needs to be used, and the complete channel information does not need to be used. Moreover, the corresponding wireless channel path location information can be provided by the channel estimation module or a dedicated channel path location search module, thus making the application of the embodiments of the present invention easier.

为便于对本发明实施例提供的通信系统中实现同步的装置有进一步理解，下面将结合附图对该装置的具体应用于接收机中的实例进行说明。In order to facilitate a further understanding of the device for realizing synchronization in the communication system provided by the embodiment of the present invention, an example of the specific application of the device in a receiver will be described below with reference to the accompanying drawings.

实施例一Embodiment one

在该实施例一中，提供了一种基于上述静态同步方案实现同步的接收机。如图4所示，在相应的OFDM发送接收机中，调制后的发送符号{s_n[k]_k＝0 ^K-1}_n＝-∞ ^+∞经过S/P(串并转换)、IFFT(反FFT)变换、P/S(并串转换)后，插入CP，在插入CP后再经过D/A(数模)变换处理，并经过等效基带信道后得到接收信号y(t)。其后，相应的接收信号y(t)经过接收机进行A/D(模数转换)等处理，其中包括：多径搜索模块(可以和前述装置实施例中的信道各径位置和功率获取单元相类比)、符号定时信息估计模块和符号定时模块，相应的多径搜索模块基于较长时间的搜索或者跟踪，提供信道各径位置τ_i和各径能量σ_i ²，然后由符号定时信息估计模块根据信道各径位置τ_i和各径能量σ_i ²采用上述本发明实施例提供的静态同步方案估计得到符号定时信息

最后，由符号定时模块根据该定时信息执行符号同步。In the first embodiment, a receiver for realizing synchronization based on the above static synchronization scheme is provided. As shown in Figure 4, in the corresponding OFDM transceiver, the modulated transmitted symbol {s _n [k] _k=0 ^K-1 } _n=-∞ ^+∞ undergoes S/P (serial-to-parallel conversion), IFFT After (inverse FFT) transformation and P/S (parallel-to-serial conversion), insert CP, after inserting CP, undergo D/A (digital-to-analog) conversion processing, and obtain the received signal y(t) after passing through the equivalent baseband channel. Thereafter, the corresponding received signal y(t) is processed by the receiver through A/D (analog-to-digital conversion), which includes: a multipath search module (which can be the same as the channel position and power acquisition unit in the aforementioned device embodiments By analogy), the symbol timing information estimation module and the symbol timing module, the corresponding multipath search module provides the position τ _i and the energy σ _i ² of each path of the channel based on the long-term search or tracking, and then estimates from the symbol timing information The module obtains the symbol timing information by estimating the static synchronization scheme provided by the embodiment of the present invention according to the position τ _i of each path of the channel and the energy of each path σ _i ²

Finally, symbol synchronization is performed by the symbol timing module based on the timing information.

实施例二Embodiment two

在该实施例二中，提供了另一种基于上述静态同步方案实现同步的接收机。如图5所示，在相应的接收机中，与图4所示的接收机的不同之处在于：由信道估计模块得到的频域信道估计值经过IDFT(逆DFT)模块后得到时域信道因子，相应的多径信息提取模块基于该时域信道因子采用类似多径搜索模块中部分功能对应的处理，提取出信道各径位置τ_i和各径功率σ_i ²，并提供给符号定时信息估计模块，该符号定时信息估计模块采用本发明实施例提供的静态同步方案便可以估计得到符号定时信息

。接着，便可以由符号定时模块根据该定时信息对接收信号执行符号同步，获得当前符号FFT积分区间，在此过程中同时进行了去CP处理。此后的FFT变换及频域处理等属于OFDM解调的典型装置，故不再详述。该接收机尤其适用于导频子载波块状连续分布的OFDM系统或单载波系统。In the second embodiment, another receiver that realizes synchronization based on the above static synchronization scheme is provided. As shown in Figure 5, in the corresponding receiver, the difference from the receiver shown in Figure 4 is that the frequency domain channel estimation value obtained by the channel estimation module passes through the IDFT (inverse DFT) module to obtain the time domain channel factor, the corresponding multipath information extraction module uses the corresponding processing similar to some functions in the multipath search module based on the time-domain channel factor to extract the channel position τ _i and the power σ _i ² of each path, and provide them to the symbol timing information An estimation module, the symbol timing information estimation module can estimate the symbol timing information by using the static synchronization scheme provided by the embodiment of the present invention

. Then, the symbol timing module can perform symbol synchronization on the received signal according to the timing information to obtain the FFT integration interval of the current symbol, and de-CP processing is performed at the same time during this process. The subsequent FFT transformation and frequency domain processing are typical devices for OFDM demodulation, so they will not be described in detail. The receiver is especially suitable for an OFDM system or a single-carrier system in which the pilot frequency subcarriers are continuously distributed in blocks.

本发明实施例中提供的大时延信道中的静态同步技术方案，不再需要精确的信道因子，从而降低了信道估计误差对同步性能的影响，并可以降低实现同步过程的复杂程度。本发明实施例特别适用于在OFDM或SC-FDE系统中由于多径时延扩展长度大于CP长度导致CP不足时的符号同步，其具有较低的复杂度和性能鲁棒性。The static synchronization technical solution in the long-delay channel provided in the embodiment of the present invention no longer needs an accurate channel factor, thereby reducing the impact of channel estimation errors on synchronization performance and reducing the complexity of the synchronization process. The embodiment of the present invention is particularly suitable for symbol synchronization when the CP is insufficient due to the multipath delay extension length being greater than the CP length in an OFDM or SC-FDE system, and has low complexity and performance robustness.

以上所述，仅为本发明较佳的具体实施方式，但本发明的保护范围并不局限于此，任何熟悉本技术领域的技术人员在本发明揭露的技术范围内，可轻易想到的变化或替换，都应涵盖在本发明的保护范围之内。因此，本发明的保护范围应该以权利要求的保护范围为准。The above is only a preferred embodiment of the present invention, but the scope of protection of the present invention is not limited thereto. Any person skilled in the art within the technical scope disclosed in the present invention can easily think of changes or Replacement should be covered within the protection scope of the present invention. Therefore, the protection scope of the present invention should be determined by the protection scope of the claims.

Claims

1. A method for synchronization in a broadband wireless communication system, comprising:

obtaining multi-path position and power information of each path of a channel in a communication system;

calculating the total signal-to-interference-and-noise ratio of all sub-carrier symbols corresponding to the initial position of the Fast Fourier Transform (FFT) integration interval in a preset initial position set according to the multi-path position of the channel and the power information of each path, wherein the number of the initial positions of the FFT integration interval in the preset initial position set is multiple;

taking the initial position of an integral interval corresponding to the maximum total signal-to-interference-and-noise ratio in the total signal-to-interference-and-noise ratios obtained by calculation as a symbol timing position, and carrying out synchronous processing on a received signal, wherein the symbol timing position is the initial position of an FFT integral interval;

in the calculation of the total signal to interference and noise ratio, the average power of the channel factors of each channel is used for replacing the channel factors to calculate the signal to interference and noise ratio.

2. The method of claim 1, wherein the calculating the total sir of all the sub-carrier symbols corresponding to the start position of the integration interval according to the multipath position of the channel and the power information of each path comprises:

according to the multi-path position and each path power information of the wireless channel

For a preset starting position set [ -tau [)_min，τ_max]Or calculating a corresponding first total signal to interference and noise ratio or a second total signal to interference and noise ratio at the starting position tau of the FFT integral interval in the quantized starting position set; the first total signal-to-interference-and-noise ratio is the total signal-to-interference-and-noise ratio of all sub-carrier symbols obtained by calculation when the receiver does not perform decision feedback-based inter-block interference (IBI) cancellation, and the second total signal-to-interference-and-noise ratio is the total signal-to-interference-and-noise ratio of all sub-carrier symbols obtained by calculation when the receiver performs decision feedback-based IBI cancellation.

3. The method of claim 2,

the calculation formula of the first total signal to interference plus noise ratio is as follows:

the calculation formula of the second total signal to interference plus noise ratio is as follows:

<math><mrow><mover><mi>SINR</mi><mo>&OverBar;</mo></mover><mrow><mo>(</mo><mi>τ</mi><mo>)</mo></mrow><mo>=</mo><mfrac><mrow><msub><mi>Σ</mi><mrow><msub><mi>τ</mi><mi>i</mi></msub><mo>></mo><mi>τ</mi></mrow></msub><msubsup><mi>σ</mi><mi>i</mi><mn>2</mn></msubsup><msup><mrow><mo>(</mo><mn>1</mn><mo>-</mo><mfrac><mrow><msub><mi>τ</mi><mi>i</mi></msub><mo>-</mo><mi>τ</mi></mrow><msub><mi>T</mi><mi>S</mi></msub></mfrac><mo>)</mo></mrow><mn>2</mn></msup><mo>+</mo><msub><mi>Σ</mi><mrow><mi>τ</mi><mo>-</mo><msub><mi>T</mi><mi>G</mi></msub><mo>≤</mo><msub><mi>τ</mi><mi>i</mi></msub><mo><</mo><mi>τ</mi></mrow></msub><msubsup><mi>σ</mi><mi>i</mi><mn>2</mn></msubsup><mo>+</mo><msub><mi>Σ</mi><mrow><msub><mi>τ</mi><mi>i</mi></msub><mo><</mo><mi>τ</mi><mo>-</mo><msub><mi>T</mi><mi>G</mi></msub></mrow></msub><msubsup><mi>σ</mi><mi>i</mi><mn>2</mn></msubsup><msup><mrow><mo>(</mo><mn>1</mn><mo>-</mo><mfrac><mrow><mi>τ</mi><mo>-</mo><msub><mi>τ</mi><mi>i</mi></msub><mo>-</mo><msub><mi>T</mi><mi>G</mi></msub></mrow><msub><mi>T</mi><mi>S</mi></msub></mfrac><mo>)</mo></mrow><mn>2</mn></msup></mrow><mrow><msub><mi>Σ</mi><mrow><msub><mi>τ</mi><mi>i</mi></msub><mo>></mo><mi>τ</mi></mrow></msub><msubsup><mi>σ</mi><mi>i</mi><mn>2</mn></msubsup><mfrac><mrow><msub><mi>τ</mi><mi>i</mi></msub><mo>-</mo><mi>τ</mi></mrow><msub><mi>T</mi><mi>S</mi></msub></mfrac><mrow><mo>(</mo><mn>1</mn><mo>-</mo><mfrac><mrow><msub><mi>τ</mi><mi>i</mi></msub><mo>-</mo><mi>τ</mi></mrow><msub><mi>T</mi><mi>S</mi></msub></mfrac><mo>)</mo></mrow><mo>+</mo><msub><mi>Σ</mi><mrow><msub><mi>τ</mi><mi>i</mi></msub><mo><</mo><mi>τ</mi><mo>-</mo><msub><mi>T</mi><mi>G</mi></msub></mrow></msub><msubsup><mi>σ</mi><mi>i</mi><mn>2</mn></msubsup><mfrac><mrow><mi>τ</mi><mo>-</mo><msub><mi>τ</mi><mi>i</mi></msub><mo>-</mo><msub><mi>T</mi><mi>G</mi></msub></mrow><msub><mi>T</mi><mi>S</mi></msub></mfrac><mrow><mo>(</mo><mn>2</mn><mo>-</mo><mfrac><mrow><mi>τ</mi><mo>-</mo><msub><mi>τ</mi><mi>i</mi></msub><mo>-</mo><msub><mi>T</mi><mi>G</mi></msub></mrow><msub><mi>T</mi><mi>S</mi></msub></mfrac><mo>)</mo></mrow><mo>+</mo><mfrac><mn>1</mn><mi>ρ</mi></mfrac></mrow></mfrac><mo>,</mo></mrow></math>

wherein, tau_iIs the time delay position of the ith path, tau is the starting position of all FFT integration intervals to be searched in a preset interval,is the power of the ith path, T_sSymbol period length, T, for multi-carrier systems or single carrier systems employing frequency domain equalization_GIs the cyclic prefix CP length, and ρ is the frequency domain subcarrier symbol signal-to-noise ratio.

4. The method according to any of claims 1 to 3, wherein the broadband wireless communication system comprises a multi-carrier communication system or a single-carrier communication system employing frequency domain equalization.

5. An apparatus for synchronization in a broadband wireless communication system, comprising:

a channel each path position and power obtaining unit, which is used for obtaining the multi-path position and each path power information of the channel in the communication system;

a total signal to interference plus noise ratio calculation unit, configured to calculate, according to the multipath positions and the path power information of the channels obtained by the channel path position and power obtaining unit, total signal to interference plus noise ratios of all subcarriers corresponding to an FFT integration interval starting position in a preset starting position set, where the FFT integration interval starting position in the preset starting position set includes multiple FFT integration interval starting positions, and the average power of channel factors of each path of the channels is used to replace the channel factor itself to perform signal to interference plus noise ratio calculation;

and the symbol synchronization execution unit is used for taking the initial position of the integral interval corresponding to the maximum total signal-to-interference-and-noise ratio in the total signal-to-interference-and-noise ratios calculated by the total signal-to-interference-and-noise ratio calculation unit as a symbol timing position, and carrying out synchronization processing on the received signals, wherein the symbol timing position is the initial position of the FFT integral interval.

6. The apparatus according to claim 5, wherein the total signal to interference plus noise ratio calculation unit is specifically configured to: according to the multi-path position and each path power information of the wireless channelFor a preset starting position set [ -tau [)_min，τ_max]Or calculating a corresponding first total signal to interference and noise ratio or a second total signal to interference and noise ratio at the starting position tau of the FFT integral interval in the quantized starting position set; the first total signal-to-interference-and-noise ratio is the total signal-to-interference-and-noise ratio of all sub-carrier symbols obtained by calculation when the receiver does not perform decision feedback-based inter-block interference (IBI) cancellation, and the second total signal-to-interference-and-noise ratio is the total signal-to-interference-and-noise ratio of all sub-carrier symbols obtained by calculation when the receiver performs decision feedback-based IBI cancellation.

7. The apparatus of claim 6,

wherein, tau_iIs the time delay position of the ith path, tau is the starting position of all FFT integration intervals to be searched in a preset interval,

is the power of the ith path, T_sSymbol period length, T, for multi-carrier systems or single carrier systems employing frequency domain equalization_GIs the cyclic prefix CP length, and ρ is the frequency domain subcarrier symbol signal-to-noise ratio.

8. The apparatus of claim 5, 6 or 7, wherein the broadband wireless communication system comprises a multi-carrier communication system or a single-carrier communication system employing frequency domain equalization.

9. A receiver comprising means for performing synchronization in a broadband wireless communication system according to any one of claims 5 to 8.