CN101122804A - A low dropout linear regulator - Google Patents
A low dropout linear regulator Download PDFInfo
- Publication number
- CN101122804A CN101122804A CNA2007101214826A CN200710121482A CN101122804A CN 101122804 A CN101122804 A CN 101122804A CN A2007101214826 A CNA2007101214826 A CN A2007101214826A CN 200710121482 A CN200710121482 A CN 200710121482A CN 101122804 A CN101122804 A CN 101122804A
- Authority
- CN
- China
- Prior art keywords
- ldo
- stage
- variable
- current
- voltage
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
- 239000003990 capacitor Substances 0.000 claims description 13
- 230000002441 reversible effect Effects 0.000 claims description 9
- 239000003381 stabilizer Substances 0.000 claims description 7
- 230000004044 response Effects 0.000 abstract description 20
- 230000003071 parasitic effect Effects 0.000 abstract description 15
- 230000007246 mechanism Effects 0.000 abstract description 13
- 230000003068 static effect Effects 0.000 abstract 1
- 238000010586 diagram Methods 0.000 description 23
- 230000008859 change Effects 0.000 description 9
- 238000005070 sampling Methods 0.000 description 9
- 230000002829 reductive effect Effects 0.000 description 8
- 230000000694 effects Effects 0.000 description 7
- 238000000034 method Methods 0.000 description 7
- 230000001052 transient effect Effects 0.000 description 5
- 230000008901 benefit Effects 0.000 description 4
- 230000010363 phase shift Effects 0.000 description 4
- 238000005516 engineering process Methods 0.000 description 3
- 230000008569 process Effects 0.000 description 3
- 230000009467 reduction Effects 0.000 description 3
- 238000004458 analytical method Methods 0.000 description 2
- 230000033228 biological regulation Effects 0.000 description 2
- 238000004519 manufacturing process Methods 0.000 description 2
- 230000004048 modification Effects 0.000 description 2
- 238000012986 modification Methods 0.000 description 2
- 238000006243 chemical reaction Methods 0.000 description 1
- 230000000052 comparative effect Effects 0.000 description 1
- 230000008878 coupling Effects 0.000 description 1
- 238000010168 coupling process Methods 0.000 description 1
- 238000005859 coupling reaction Methods 0.000 description 1
- 230000007547 defect Effects 0.000 description 1
- 238000009792 diffusion process Methods 0.000 description 1
- 230000002401 inhibitory effect Effects 0.000 description 1
- 239000011229 interlayer Substances 0.000 description 1
- 229910052751 metal Inorganic materials 0.000 description 1
- 239000002184 metal Substances 0.000 description 1
- 230000001105 regulatory effect Effects 0.000 description 1
- 230000000452 restraining effect Effects 0.000 description 1
- 239000004065 semiconductor Substances 0.000 description 1
- 238000005728 strengthening Methods 0.000 description 1
- 230000001629 suppression Effects 0.000 description 1
- 229910052715 tantalum Inorganic materials 0.000 description 1
- GUVRBAGPIYLISA-UHFFFAOYSA-N tantalum atom Chemical compound [Ta] GUVRBAGPIYLISA-UHFFFAOYSA-N 0.000 description 1
Images
Landscapes
- Amplifiers (AREA)
Abstract
一种低压差线性稳压器，通过在误差放大器的输出端引入可变跨导的gm/C滤波器电路模块，实现了高输出电压精度、高稳定性、高电源电压抑制比等性能指标。滤波器电路模块的可变跨导机制，降低了稳压器在空载和轻载时的静态功耗，并且能够产生一个频率随稳压器负载电流增加而向高频移动的寄生极点，从而拓展了稳压器的单位增益带宽，增强了稳压器在重度负载及满载时的稳定性。此外，稳压器环路中采用的可变电流高增益放大级，增加了稳压器内部节点的压摆率，提高了稳压器的响应性能。
A low-dropout linear voltage regulator, by introducing a variable transconductance gm/C filter circuit module at the output of the error amplifier, realizes performance indicators such as high output voltage accuracy, high stability, and high power supply voltage rejection ratio. The variable transconductance mechanism of the filter circuit module reduces the static power consumption of the regulator at no load and light load, and can generate a parasitic pole whose frequency moves to high frequency as the load current of the regulator increases, thereby The unity-gain bandwidth of the regulator is extended, which enhances the stability of the regulator at heavy load and full load. In addition, the variable current high-gain amplifier stage used in the regulator loop increases the slew rate of the internal nodes of the regulator and improves the response performance of the regulator.
Description
Technical field
The present invention relates to a kind of low pressure difference linear voltage regulator, particularly a kind of variable transconductance gm/C filter circuit module and variable current high-gain amplifier stage of utilizing realized the high stability high precision high power supply voltage rejection ratio low pressure difference linear voltage regulator of response fast.
Background technology
Electric power management circuit is one of important module during portable electronic is used, and it generally includes switching power circuit and low pressure difference linear voltage regulator (LDO, Low-Dropout Voltage Regulator) circuit etc.The conversion efficiency of switching power circuit can reach more than 90% usually, but its chip area is big, and output voltage ripple is big, and needs to use magnetic cell; Therefore characteristics such as and that the LDO circuit has a chip area is little, and output voltage ripple is little are widely used in and are the high performance analog circuit supply.
The high performance analog circuit has proposed very high requirement to the output voltage precision of LDO.The output voltage precision mainly comprises the precision of output voltage steady-state value and the precision of output voltage instantaneous value.The former useable linear regulation, load regulation and the indexs such as steady-state value rate of change of output voltage under different process, supply voltage, temperature characterize, and indexs such as latter's disposable load transient response characteristic are weighed.For improving the precision of output voltage steady-state value, have the high precision except requiring the employed reference voltage of LDO, also need to improve the precision of feedback factor in the feedback network and the loop gain of LDO circuit.And the precision of raising output voltage instantaneous value, the slew rate (Slew Rate) that needs to increase the unity gain bandwidth (UGF) of LDO loop usually and increase internal node in the LDO loop.But loop gain is high more, UGF is high more, and the stability of LDO system is just poor more, and its frequency compensation difficulty is just big more.For improving the slew rate of internal node (the particularly node of power tube grid end among the LDO), need increase to be connected to the amplifier stage of this node or the quiescent current of buffer stage usually, thereby increased the quiescent dissipation of LDO, influenced the application of LDO in portable product.
At present, all integrated RF circuit and other high performance circuit very responsive in nearly all portable set to noise, this just requires LDO to have excellent supply voltage noise inhibiting ability, this index uses supply-voltage rejection ratio (PSRR, Power Supply Rejection Ratio) to weigh usually.The PSRR of LDO is high more, and it is strong more to the supply voltage capability of restraining noise, and corresponding output voltage ripple is just more little.
Fig. 1 has provided existing first kind of typical LDO theory diagram.This circuit is made up of voltage reference circuit 101, error amplifier 102, driving buffer stage 108, power tube 104, dividing potential drop sampling network 105, output capacitance 106 and load 107.Voltage reference circuit 101 produce one constant, be not subjected to the reference voltage V of condition variable effects such as supply voltage, temperature substantially
REFThe output voltage V of LDO
OUTAfter 105 samplings of dividing potential drop sampling network, produce feedback voltage V
FBReference voltage V
REFAnd feedback voltage V
FBBe connected to two input ends of error amplifier 102 respectively, its difference produces a control signal after error amplifier 102 amplifies, this control signal is used for regulating the duty of power tube 104, thereby guarantees the output voltage V of LDO through driving the grid end that buffer stage 108 is passed to power tube 104
OUTStill is nominal value in supply voltage, working temperature, loading condition when changing.Drive buffer stage 108 and realize by voltage follower that usually its effect is the frequency that improves power tube 104 grid place parasitic poles (being designated as Pgate), to guarantee the stability of LDO loop.
The major defect of circuit shown in Figure 1 is that loop gain is lower, so the precision of LDO output voltage steady-state value is not high, and the PSRR performance of low-frequency range can't satisfy the requirement of portable set.
For improving performance, Fig. 2 has described existing second kind of typical LDO theory diagram.The difference of this circuit and circuit shown in Figure 1 is: this circuit is at the output terminal of the error amplifier 102 of circuit shown in Figure 1 and drive between the input end of buffer stage 108 and increased the low gain stage 202 that is used for improving the PSRR performance, and remainder is identical with Fig. 1.
Though circuit shown in Figure 2 has increased low gain stage 202, because its gain is lower, drive buffer stage 108 simultaneously and (for example can not provide gain, driving buffer stage realizes with voltage follower, its gain is 1), therefore the loop gain of whole LDO is still lower, makes circuit shown in Figure 2 also can't satisfy the requirement of high performance system to LDO output voltage precision.In addition, increase a parasitic poles in the feedback control loop that is introduced in LDO of low gain stage 202, thereby reduced the stability of loop.
Fig. 3 has provided existing the third typical LDO theory diagram.The difference of this circuit and circuit shown in Figure 1 is: the driving buffer stage 108 that adopts fixed current high-gain amplifier stage 109 to replace among Fig. 1, remainder is identical with Fig. 1.Owing to adopted fixed current high-gain amplifier stage 109, realized high loop gain, thereby effectively increased the precision of output voltage steady-state value.But the introducing of fixed current high-gain amplifier stage 109 makes the parasitic poles (Pgate) that is positioned at power tube 104 grid ends be positioned at low frequency, therefore need compensate this loop.Normal a kind of compensation method of adopting utilizes output capacitance 106
COUTWith the equivalent series resistance (R on it
ESR) low frequency Z at zero point of generation
ESROffset low-frequency pole Pgate.There is following shortcoming in this compensation method: the equivalent series resistance of output capacitance is difficult to accurately to determine and changes with the variation of conditions such as temperature, makes accurately to offset with limit Pgate the zero point that is produced, thereby reduced the stability of loop; For producing low frequency Z at zero point
ESR, need bigger R
ESRThereby, reduced the load transient response performance of LDO; For producing low frequency Z at zero point
ESR, need to use the bigger tantalum electric capacity of capacitance, thereby increased the area and the cost of circuit board; The low frequency Z at zero point that is produced
ESRPromptly allow to guarantee the steady operation of LDO circuit, use the LDO circuit of this method design can to reduce because of the existence of low-frequency pole Pgate in the PSRR of Mid Frequency performance.
Problem for the Mid Frequency PSRR poor performance that solves Fig. 3 circuit, the 4th kind of typical LDO theory diagram shown in Figure 4 introduced the fixedly filter circuit module 301 of mutual conductance between the input end of dividing potential drop sampling network 105 and error amplifier 102 on the basis of circuit shown in Figure 3.The filter circuit module 301 of described fixedly mutual conductance is made up of transconductance stage gm (this transconductance stage provides bias current by the fixing bias current sources of current value) and resistance R, the capacitor C of fixedly mutual conductance, and the input end in the same way of transconductance stage gm and output terminal are respectively the input end and the output terminal of the filter circuit module 301 of fixedly mutual conductance.The fixing filter circuit module 301 of the mutual conductance Z at zero point that can to produce a frequency be 1/RC
CBy the value of R and C rationally is set, can obtain a position can exactly determined low frequency zero point and it is used for offsetting low-frequency pole Pgate.This compensation method mainly contains two advantages: overcome the dependence of Fig. 3 circuit to the type and size of output capacitance, thereby improved stability and the load transient response performance of LDO, reduced system cost; Low frequency Z at zero point
CExistence can improve LDO in the PSRR of Mid Frequency performance.
Yet there are several shortcomings clearly in the circuit among Fig. 4:
1, the introducing of circuit module 301 has influenced the quiescent point of LDO, has reduced the precision of the LDO loop feedback factor.Introduce before the module 301, feedback factor is R2/ (R1+R2).This factor is two ratios of the resistance of coupling mutually, so its precision is higher.After introducing module 301, feedback factor becomes filter circuit module 301 input ends of fixedly mutual conductance to the DC gain of output terminal and the product of R2/ (R1+R2).Because the DC of module 301 gain (this gain is gm*ro/ (gm*ro+1), and wherein ro is the output impedance of transconductance stage gm) is a value with technology, supply voltage, temperature change, make the precision of feedback factor of LDO reduce.Because the output voltage steady-state value precision of LDO is directly related with the precision of feedback factor, so after introducing the filter circuit module 301 of fixedly mutual conductance, the output voltage precision of LDO reduces.
2, fixedly the introducing of the filter circuit module 301 of mutual conductance has reduced the stability of LDO loop, particularly in the stability of load of LDO severe and full load.For the output voltage that guarantees LDO is always steady state value from zero load in load in fully loaded variation range, need the feedback factor of LDO not change with loading condition, that is fixedly the DC gain of the filter circuit module 301 of mutual conductance does not change with loading condition, so the transconductance stage gm in the filter circuit module 301 of fixedly mutual conductance need use fixing bias current.LDO is applied in the portable product usually, for the quiescent current that reduces its consumption to prolong the serviceable life of battery, need reduce the bias current of transconductance stage gm as far as possible.And fixedly the filter circuit module 301 of mutual conductance when producing 1/RC at zero point, the parasitic poles Pc that to produce a frequency again be gm/C.The bias current of gm is more little, and the position of this limit is low more.The LDO band carries big more, and its UGF is wide more, and parasitic poles Pc is big more to the stability influence of loop.
3, be the influence of the parasitic poles Pc in the filter circuit module 301 that reduces fixedly mutual conductance, need to reduce the DC gain and the UGF of loop LDO stability.As previously mentioned, loop gain is low more, and the output voltage precision is low more; UGF is narrow more, and transient response speed is slow more.
4, between dividing potential drop sampling network 105 and error amplifier 102 input ends, introduce the fixedly filter circuit module 301 of mutual conductance, reduced LDO to the squelch performance of output voltage (directly the be added to input end of error amplifier 102 and amplify of the noise that module 301 produces, thereby increased the output voltage noise of LDO) through error amplifier 102.
5, for to satisfy system's requirement to quiescent dissipation when zero load, the bias current of employed fixed current high-gain amplifier stage 109 can not be excessive, and this has just limited the slew rate of internal node, thereby limited the response speed of LDO.Need to prove that this shortcoming also is present in the circuit shown in Figure 3.
Summary of the invention
Technology of the present invention is dealt with problems and is: the shortcoming that can not realize high output voltage precision, high stability, high power supply voltage rejection ratio at existing LDO circuit simultaneously, by in the LDO circuit, introducing the filter circuit module of variable transconductance, provide the low differential voltage linear voltage stabilizer circuit of the high PSRR of a kind of high-precision and high-stability.In addition, owing to adopted variable transconductance filter circuit module and variable current high-gain amplifier stage, make LDO circuit of the present invention also have the characteristics of quick response, low quiescent current.
Technical solution of the present invention is: a kind of low pressure difference linear voltage regulator, comprise voltage reference circuit, error amplifier, high-gain amplifier stage and power tube, the reference voltage that voltage reference circuit produces is connected to the reverse input end of error amplifier or input end in the same way, the output terminal of high-gain amplifier stage is connected to the grid end of power tube, the source end of power tube and drain terminal are respectively as the voltage input end and the voltage output end of voltage stabilizer of the present invention, and the voltage of voltage output end directly or be connected to another input end that does not link to each other with voltage reference circuit of error amplifier after the dividing potential drop sampling, it is characterized in that: between the input end of the output terminal of error amplifier and high-gain amplifier stage, also comprise one by the variable transconductance level, the filter circuit module that resistance and electric capacity are formed, wherein the input end in the same way of variable transconductance level and output terminal are respectively the input end and the output terminal of filter circuit module, the two ends of resistance are connected to the reverse input end and the output terminal of variable transconductance level respectively, one end of electric capacity links to each other with the reverse input end of transconductance stage, and the other end of electric capacity links to each other with earth potential or set potential.
The bias current sources of variable transconductance level is made of the variable bias current source of a fixed bias current source and a size of current variation in the described filter circuit module.
The size of current in described variable bias current source is proportional to the load current of voltage stabilizer of the present invention.
Described high-gain amplifier stage is a variable current high-gain amplifier stage, and the electric current that flows through this high-gain amplifier stage is the electric current of variable size.
The electric current that flows through described high-gain amplifier stage increases with the increase of voltage stabilizer load current.
The electric current that flows through described high-gain amplifier stage is proportional to the bias current of described filter circuit module.
Described high-gain amplifier stage comprises at least one gain per stage level or buffer stage amplifier.
The present invention's advantage compared with prior art is: a kind of low pressure difference linear voltage regulator, by introducing filter circuit module 302 and the variable current high-gain amplifier stage of forming by variable transconductance level gm, resistance and electric capacity 103, performances such as high output voltage precision, high stability, high power supply voltage rejection ratio, response fast, low quiescent current, low output noise have been realized.
1, high output voltage precision.The use of variable current high-gain amplifier stage 103, the gain when having significantly improved the gain, particularly underloading of LDO loop and intermediate part load, thus improved the precision of output voltage steady-state value; Circuit module 302 is positioned at the output terminal of error amplifier 102, has overcome the shortcoming that conventional filter circuit module 301 shown in Figure 4 influences the LDO quiescent point, reduces the feedback factor precision, has guaranteed voltage accuracy.
2, high stability.Filter circuit module 302 produces the Zc at zero point that can accurately control (by the value of R, C rationally is set, can be placed low frequency this zero point) that frequency is 1/RC, is used to offset loop medium and low frequency limit (for example Pgate) this zero point, has overcome ESR (Z at zero point
ESR) system stability that causes because of the frequency location out of true reduces; The variable transconductance mechanism of filter circuit module 302 can produce a frequency increases the parasitic poles Pc (its frequency is gm/C) that raises with the LDO load current, be in severe load and full load at LDO, its UGF widens, move to high frequency and parasitic poles Pc is also corresponding, thereby guarantee the stability of LDO in whole loading range.
3, high power supply voltage rejection ratio.Low frequency Zc at zero point is used for offsetting the reduction of the PSRR performance that low-frequency pole (for example Pgate) caused, and has strengthened the PSRR of Mid Frequency, and ESR does not possess this function zero point; The variable transconductance mechanism of filter circuit module 302 not only has the effect of stability when strengthening the LDO heavy duty, and helps improving the DC and the low-frequency gain of loop, thus the PSRR performance when improving the DC of loop and low frequency.
4, response fast.The variable transconductance mechanism of filter circuit module 302 has been expanded the UGF of LDO, thereby has shortened the response time of loop; The variable current principle of high-gain amplifier stage 103 significantly increased the slew rate at LDO power tube grid place when large current load, and this index normally restricts the primary factor of voltage stabilizer response speed.
5, low quiescent current.The size of current and the load current of filter circuit module 302 and high-gain amplifier stage 103 change on year-on-year basis, and when LDO was in zero load and slight load, these two module consumed current were very little.Because LDO overwhelming majority time service is in zero load and light condition, therefore the quiescent current that reduces under this state has important meaning.
6, low output noise.Filter circuit module 302 is between the input end of the output terminal of error amplifier 102 and variable current high-gain amplifier stage 103, overcome conventional circuit module 301 and introduced the shortcoming of noise, thereby reduced the output noise of LDO at the input end of error amplifier 102.
Description of drawings
Fig. 1 is existing first kind of typical LDO theory diagram;
Fig. 2 is existing second kind of typical LDO theory diagram;
Fig. 3 is existing the third typical LDO theory diagram;
Fig. 4 is existing the 4th kind of typical LDO theory diagram;
Fig. 5 is a kind of typical LDO theory diagram of the present invention;
Fig. 6 is a kind of connected mode in variable bias current source in the LDO theory diagram of the present invention shown in Figure 5;
Fig. 7 is a kind of circuit implementation of the filter circuit module 302 of variable transconductance in the present invention's LDO theory diagram shown in Figure 5;
Fig. 8 is a kind of circuit implementation of the present invention's LDO theory diagram shown in Figure 5;
Fig. 9 is a kind of circuit implementation of typical LDO theory diagram shown in Figure 4;
Figure 10 is the contrast synoptic diagram of Fig. 8 and circuit output voltage steady-state value precision shown in Figure 9;
Figure 11 is the contrast synoptic diagram of Fig. 8 and circuit amplitude versus frequency characte shown in Figure 9 and phase-frequency characteristic;
Figure 12 is the contrast synoptic diagram of Fig. 8 and circuit power voltage rejection ratio characteristic shown in Figure 9.
Embodiment
Fig. 5 has provided a kind of typical LDO theory diagram of the present invention.This LDO is made up of the filter circuit module 302 of voltage reference circuit 101, error amplifier 102, variable transconductance, variable current high-gain amplifier stage 103 (electric current that flows through this grade increases with the increase of LDO load current), power tube 104, dividing potential drop sampling network 105, output capacitance 106 and load 107.The difference of itself and circuit shown in Figure 3 is: replace fixed current high-gain amplifier stages 109 with variable current high-gain amplifier stage 103, and added the filter circuit module 302 of variable transconductance between the input end of the output terminal of error amplifier 102 and variable current high-gain amplifier stage 103.
The filter circuit module 302 of described variable transconductance is made up of transconductance stage gm, resistance R and the capacitor C of variable transconductance, the input end and the output terminal of the filter circuit module 302 that the input end in the same way of the transconductance stage gm of variable transconductance and output terminal are respectively variable transconductance, the two ends of resistance R are connected to the reverse input end and the output terminal of transconductance stage respectively, one end of capacitor C links to each other other end ground connection with the reverse input end of the transconductance stage gm of variable transconductance.Give a kind of implementation of the transconductance stage gm of variable transconductance among Fig. 5: promptly on the basis of the transconductance stage gm of the fixedly mutual conductance that original use fixed bias current source lb realizes, increasing a size of current is proportional to the variable bias current source 311 of LDO load current (its current value is k*lout, wherein lout is the load current of LDO, and k is generally the scale factor much smaller than 1).Need to prove that according to the design conditions of side circuit, fixed bias current source lb and variable bias current source 311 also might be connected (for example, when the input difference of described transconductance stage gm is nmos pass transistor to pipe) between transconductance stage gm and the ground (GND).But two kinds of connected modes have the identical effect that required variable bias current is provided for described transconductance stage gm.
The Zc at zero point that it is 1/RC that described module 302 can produce a frequency (, can be placed low frequency this zero point) and the parasitic poles Pc (its frequency is gm/C) that frequency changes with the LDO load current by the value of R, C rationally is set.Described zero point, Zc mainly contained two effects: the first, and be used to offset the negative that high-gain LDO loop medium and low frequency limit (for example, being positioned at the low-frequency pole Pgate at power tube 104 grid places) produces and move, thereby strengthen the stability of feedback control loop; The second, can be used to this zero point offset the supply-voltage rejection ratio performance reduction that low-frequency pole Pgate causes, thereby increase the PSRR of LDO at Mid Frequency.
Need to prove the low frequency Z at zero point that passes through output capacitance and the generation of the equivalent series resistance on it described in Fig. 3
ESRThough, also have the function that strengthens loop stability, do not have and improve the function of LDO at the PSRR of Mid Frequency.
The filter circuit module 301 and the fixed current high-gain amplifier stage 109 of the fixedly mutual conductance of using with Fig. 4 are compared, and the filter circuit module 302 of the variable transconductance that uses among Fig. 5 and variable current high-gain amplifier stage 103 have the following advantages:
1, the filter circuit module 302 of variable transconductance is between the input end of the output terminal of error amplifier 102 and variable current high-gain amplifier stage 103, this position can not reduce the precision of the LDO loop feedback factor, introduces the LDO output voltage precision reduction that module 301 is caused among Fig. 4 thereby eliminated.
2, the use of the filter circuit module 302 of variable transconductance can significantly strengthen the stability of LDO loop, particularly in the stability of load of LDO severe and full load.The transconductance stage gm of variable transconductance can adjust its bias current (its total bias current value is lb+k*lout) according to the change dynamics of LDO load current, when LDO is in light condition, lout=0, therefore the bias current of the transconductance stage gm of variable transconductance is lb, increase along with LDO load current lout, variable bias current part (k*lout) corresponding increase in total bias current of the transconductance stage gm of variable transconductance makes the also corresponding increase of mutual conductance of transconductance stage gm of variable transconductance.And the increase of mutual conductance can make the parasitic poles Pc (its frequency can be represented with gm/C) in the module 302 move to high frequency.Because the UGF of LDO broadens with the increase of load current, therefore be in heavy duty and full load at LDO, it is big that the influence of parasitic poles becomes.Therefore parasitic poles Pc with the increase of load current the corresponding characteristics that move to high frequency, guaranteed that LDO also can steady operation at full load.
3, the variable transconductance mechanism of using in the filter circuit module 302 of variable transconductance can effectively reduce the quiescent dissipation of filter circuit module 302 when LDO is in zero load and slight load.And filter circuit module 301 is used is fixedly mutual conductance mechanism, can need to use bigger fixed bias current lb usually at the full load steady operation for guaranteeing LDO shown in Figure 4, the power consumption of filter circuit module 301 when this has just increased the LDO zero load.
4, because filter circuit module 302 is connected to the output terminal of error amplifier 102, the introducing of this module can obviously not reduce LDO to output voltage Noise Suppression performance.And place the filter circuit module 301 between dividing potential drop sampling network 105 and error amplifier 102 input ends to understand the noiseproof feature that significantly reduces LDO among Fig. 4.
5, the variable transconductance mechanism of filter circuit module 302 helps improving the DC and the low-frequency gain of loop, thereby improves the PSRR performance of LDO when DC and low frequency.In addition, this mechanism has also been expanded the UGF of LDO, thereby shortens the response time of loop.
6, differently with fixed current high-gain amplifier stage 109 be that the variable current in the high-gain amplifier stage 103 can significantly increase the slew rate at LDO power tube grid place when large current load.Owing to will drive big load current, the size of power tube is very big, and its grid end stray capacitance is also very big, so the slew rate index of this node normally determines the primary factor of voltage stabilizer response speed speed.
In circuit shown in Figure 6, transistor Mps has provided a kind of circuit implementation in the variable bias current source 311 in Fig. 5 theory diagram of the present invention.The grid end of transistor Mps links to each other with the grid end of power tube 104, and the source end of transistor Mps links to each other (all being connected to power supply potential VDD) with the source end of power tube 104, and the drain terminal of transistor Mps is connected among the transconductance stage gm of variable transconductance the difference input to the source end of pipe.The breadth length ratio of transistor Mps is k times (wherein k is usually much smaller than 1) of the breadth length ratio of power tube 104, because two transistors have identical gate source voltage, under the insignificant situation of channel length modulation effect, the electric current that flows through Mps is k times (being k*lout) flowing through the electric current of power tube 104.
Fig. 7 has provided a kind of circuit of the filter circuit module 302 of variable transconductance in the LDO theory diagram shown in Figure 5 and has realized.Filter circuit module 302 is made up of transistor M11～M14, fixed bias current source lb, variable bias current source 311 and resistance R, the capacitor C of utilizing transistor Mps to realize, and wherein transistor M11～M14, current source lb and transistor Mps have constituted the transconductance stage gm with variable transconductance.The drain terminal of grid end, M12 and the M14 of M11, the grid end of M12 are respectively input end in the same way, output terminal and the reverse input end of the transconductance stage gm of variable transconductance; The two ends of resistance R are connected to reverse input end and the output terminal of the transconductance stage gm of variable transconductance respectively; One end ground connection of capacitor C, the other end links to each other with the reverse input end of the transconductance stage gm of variable transconductance.Need to prove that an end of capacitor C ground connection also can be connected to VDD or other set potential.The output terminal of input end in the same way of the transconductance stage gm of variable transconductance is the input end and the output terminal of respective filter circuit module 302 respectively.The grid end of transistor Mps links to each other with the grid end of power tube 104, thereby provides the variable bias current that changes with the LDO loading condition for the transconductance stage gm of variable transconductance.
Fig. 8 is a kind of circuit implementation of the present invention's LDO theory diagram shown in Figure 5.The filter circuit module 302 of variable transconductance is made up of transistor M11～M14, fixed bias current source lb, variable bias current source 311 and resistance R, the capacitor C of utilizing transistor Mps to realize.Need to prove that introduced the low-pass filter of being made up of resistance R 1 and capacitor C 1 between the grid end of the grid end of transistor Mps and power tube 104, its effect is to guarantee the work of LDO loop stability.Variable current high-gain amplifier stage 103 is made up of transistor M5～M8, and this amplifier stage is the push-pull type output stage, and the grid end of M6 is connected to the output terminal of the filter circuit module 302 of variable transconductance, and the grid end of M8 is connected to the grid end of transistor M13, M14.The electric current that flows through high-gain amplifier stage 103 is proportional to the bias current (lb+k*lout) of filter circuit module 302, therefore high-gain amplifier stage 103 consumed current also increase along with the increase of LDO load current, thereby the power consumption when being convenient to reduce the unloaded and underloading of LDO, and can accelerate the response speed of LDO.
For example, when emulation, the reference voltage of choosing LDO is 0.6V, and the preset value of output voltage is 1V, and input voltage is 1.5V, and the variation range of load current is 0～100mA.With this understanding, the electric current of fixed bias current source lb is 2 μ A, when LDO is in light condition, filter circuit module 302 consumed current are 2 μ A, variable current high-gain amplifier stage 103 consumed current are that (electric current that wherein flows through M6 place branch road is 4 μ A to 5 μ A, the electric current that flows through M8 place branch road is 1 μ A), so filter circuit module 302 and variable current high-gain amplifier stage 103 consumed current are 7 μ A; When LDO fully loaded (lout=100mA), filter circuit module 302 consumed current are 12 μ A (wherein variable current k*lout is 10 μ A), variable current high-gain amplifier stage 103 consumed current are 30 μ A (electric current that wherein flows through M6 place branch road is 24 μ A, and the electric current that flows through M8 place branch road is 6 μ A).
Circuit shown in Figure 8 has following characteristics:
1, the low frequency that produces of filter circuit module 302 is used for offsetting the negative that loop medium and low frequency limit produced zero point and moves, and improves LDO in the PSRR of Mid Frequency performance, has strengthened the stability of feedback control loop;
2, filter circuit module 302 is between the input end of the output terminal of error amplifier 102 and high-gain amplifier stage 103, makes the precision height of LDO output voltage steady-state value, and output noise is low;
3, filter circuit module 302 employed variable transconductance mechanism have strengthened the stability of LDO loop at severe load and full load;
4, the variable transconductance mechanism of using in the filter circuit module 302 and the variable current mechanism of high-gain amplifier stage 103 effectively reduce the quiescent dissipation of LDO when unloaded and underloading, thereby prolong the battery serviceable life of portable product;
5, the employed bias current sources that changes with loading condition in the filter circuit module 302 can improve the transient response performance of LDO.
6, the variable current mechanism of high-gain amplifier stage 103 has been improved the slew rate of LDO, has accelerated response speed.
Need to prove that the filter circuit module 302 employed transconductance stage gm among the present invention can be the transconductance stage of the known any routine of those of ordinary skill in the art, and are not limited to the circuit structure that Fig. 7 enumerates.Filter circuit module 302 employed capacitor C among the present invention can be any type of electric capacity that integrated circuit fabrication process can be realized, for example mos capacitance, poly-poly electric capacity, metal capacitance etc.; And resistance R also can be any type of resistance that integrated circuit fabrication process can be realized, for example, diffusion resistance, interlayer resistance, sheet resistance, poly resistance, is operated in resistance that the metal-oxide-semiconductor of linear zone forms etc.
Need to prove that equally employed error amplifier 102 and variable current high-gain amplifier stage 103 can be the amplifiers of the known any routine of those of ordinary skill in the art among the present invention, and are not limited to the circuit structure that Fig. 8 enumerates.
Be further to set forth the present invention's advantage compared with prior art, a kind of circuit that Fig. 9 has provided typical LDO schematic block circuit diagram shown in Figure 4 is realized and its a kind of circuit realization with principle of the invention block diagram shown in Figure 8 is compared.Need to prove that for ease of comparing, the error amplifier 102 that uses among Fig. 9, power tube 104, dividing potential drop sampling network 105, output capacitance 106 and load 107 are identical with Fig. 8.When emulation, the reference voltage of choosing LDO equally is 0.6V, and the preset value of output voltage is 1V, and input voltage is 1.5V, and the variation range of load current is 0～100mA.
In Fig. 9, fixed span waveguide filter circuit module 301 is made up of transistor M11～M14, fixed bias current source lb (its electric current is 4 μ A) and resistance R, capacitor C.Fixed current high-gain amplifier stage 109 is made up of transistor M5～M8, and its current sinking is 10 μ A (electric current that wherein flows through M6 place branch road is 8 μ A, and the electric current that flows through M8 place branch road is 2 μ A).
When zero load, filter circuit module 301 among Fig. 9 and fixed current high-gain amplifier stage 109 consumed current are 14 μ A, be the twice (latter only is 7 μ A) of circuit median filter circuit module 302 shown in Figure 8 and variable current high-gain amplifier stage 103 current sinkings, the quiescent dissipation when LDO therefore shown in Figure 9 is unloaded is bigger.
At full load, the electric current of fixed current high-gain amplifier stage 109 output stages (M6 place branch road) only is 8 μ A among Fig. 9, and the electric current of variable current high-gain amplifier stage 103 output stages (M6 place branch road) among Fig. 8 is 24 μ A, makes the latter at for the former 3 times of the slew rate of power tube 104 grid place nodes.Therefore, the variable current mechanism of variable current high-gain amplifier stage 103 of the present invention in the quiescent dissipation, has also effectively been accelerated the full load response speed when reducing the LDO zero load.
Figure 10 has provided the situation of change of output voltage steady-state value in whole operating temperature range of LDO circuit shown in Figure 8 and LDO circuit shown in Figure 9.Curve 1 is corresponding to the temperature variant curve of output voltage steady-state value that adopts prior art LDO circuit (Fig. 9) among Figure 10, and curve 2 is corresponding to the temperature variant curve of output voltage steady-state value of LDO circuit of the present invention (Fig. 8).As can be seen from the figure, when temperature when-40 ℃ change to 125 ℃, adopt the LDO output voltage steady-state value of prior art to change to 1.014V from 1.006V, its changing value is 8mV, corresponding output voltage precision is 0.8%; LDO output voltage steady-state value of the present invention changes to 1V from 1.0008V, and its changing value is 0.8mV, and corresponding output voltage precision is 0.08%.This comparative illustration, than prior art (circuit shown in Figure 9) filter circuit module 301 is placed the way of error amplifier 102 input ends, the present invention can make the output voltage steady-state value reduce an order of magnitude with variation of temperature by the output terminal that filter circuit module 302 is placed error amplifier 102.
It can also be seen that the LDO circuit output voltage steady-state value of employing prior art shown in Figure 9 (1.006V～1.014V) and the difference of 6mV～14mV is arranged between the predeterminated voltage (1V) from Figure 10.The reason that this difference occurs is: filter circuit module 301 places the input end of error amplifier 102, its DC gain is gm*ro/ (gm*ro+1), wherein ro is the output impedance of transconductance stage gm, this gain varies with temperature and changes, the feedback factor that causes LDO varies with temperature and changes, thereby making that the output voltage steady-state value is corresponding changes.In addition, even under fixed temperature, because the DC of filter circuit module 301 gain also is not equal to 1 (this gain just is approximately equal to 1), so the not strict R2/ (R1+R2) that equals of the feedback factor of LDO, causes to have occurred the difference of 6mV at least between output voltage steady-state value and the predeterminated voltage.And in LDO circuit of the present invention shown in Figure 8, owing to eliminated the influence of filter circuit module to feedback factor, the output voltage steady-state value only is 0.8mV with variation of temperature, and the difference between output voltage steady-state value and the predeterminated voltage is less than 1mV.
Figure 11 has provided the amplitude versus frequency characte and the phase-frequency characteristic curve contrast synoptic diagram of the LDO circuit of LDO circuit of the present invention shown in Figure 8 and employing prior art shown in Figure 9.Its simulated conditions is input voltage 1.5V, output voltage 1V, load current 100mA.Curve 1 and curve 3 are respectively the amplitude versus frequency characte and the phase-frequency characteristic curve of circuit shown in Figure 8, and curve 2 and curve 4 are respectively the amplitude versus frequency characte and the phase-frequency characteristic curve of circuit shown in Figure 9.
The UGF of circuit shown in Figure 8 is 3.6MHz (corresponding to an A point in the curve 1), and the phase shift at the UGF place is 123 degree (corresponding phase nargin is 57 degree, shown in B point in the curve 3); The UGF of circuit shown in Figure 9 only is 1.3MHz (corresponding to a C point in the curve 2), and the phase shift at the UGF place is 123 degree (corresponding phase nargin is 57 degree, shown in D point in the curve 4).As seen, the UGF of LDO of the present invention is far above the UGF that adopts prior art LDO, so the loop response speed of LDO of the present invention also is higher than the loop response speed of the LDO that adopts prior art.If the UGF of circuit shown in Figure 9 is designed to the 3.6MHz identical with the present invention, then its phase shift becomes 147 degree (corresponding phase nargin is 33 degree, shown in E point in the curve 4), makes loop stability reduce.
The pole-zero analysis result has also verified the correctness of theoretical analysis.In LDO of the present invention shown in Figure 8, when load current was 100mA, limit (Pgate) frequency at power tube 104 grid places was 146KHz, and the frequency of the parasitic poles Pc of filter circuit module 302 is 6.78MHz; And the position of two limits of LDO correspondence shown in Figure 9 is respectively 46.5KHz and 3.89MHz.The reason that this phenomenon occurs is: during the severe load, total bias current of Fig. 8 median filter circuit module 302 (12 μ A) makes that much larger than the bias current (4 μ A) of Fig. 9 median filter circuit module 301 the former parasitic poles position (6.78MHz) is higher 1.74 times than the latter (3.89MHz); The output stage electric current (8 μ A) of the output stage electric current of variable current high-gain amplifier stage 103 among Fig. 8 (24 μ A) fixed current high-gain amplifier stage 109 in Fig. 9, make that the former output impedance is low, so the position (146KHz) of its limit Pgate is also high 3.1 times than the latter (46.5KHz).And the raising of Pgate frequency help reducing LDO than low-frequency range (for example, the phase shift of 100KHz～1MHz), the raising of Pc frequency helps expanding the UGF of LDO, and improves the phase margin of LDO, thereby has strengthened loop stability.
Figure 12 has provided the PSRR performance comparison of LDO circuit shown in Figure 8 and LDO circuit shown in Figure 9.Curve 1 is the PSRR characteristic of LDO circuit of the present invention when load current is 1mA, and curve 2 is the existing PSRR characteristic of LDO circuit when load current is 1mA.Its quiescent current is little when underloading, loop gain is high owing to circuit of the present invention, makes its PSRR performance index be much better than LDO circuit shown in Figure 9 when DC and low frequency.
When variable current high-gain amplifier stage 103 had the above amplifier of two-stage, identical when its principle that adopts filter circuit module 302 to improve the LDO performances has only the one-level amplifier with above-mentioned variable current high-gain amplifier stage 103 repeated no more herein.
Though in the specific embodiment of the present invention related concrete LDO circuit is described, only be to be used for illustrating content of the present invention to the description that these physical circuits carried out.Under the prerequisite that does not break away from the principle of the invention, can also make the variation and the modification of various equivalences to example of the present invention, but its modification will drop in the scope of claim of the present invention all.Therefore the present invention is widely.
Claims (7)
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CNB2007101214826A CN100495281C (en) | 2007-09-07 | 2007-09-07 | A low dropout linear regulator |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CNB2007101214826A CN100495281C (en) | 2007-09-07 | 2007-09-07 | A low dropout linear regulator |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CN101122804A true CN101122804A (en) | 2008-02-13 |
| CN100495281C CN100495281C (en) | 2009-06-03 |
Family
ID=39085157
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CNB2007101214826A Expired - Fee Related CN100495281C (en) | 2007-09-07 | 2007-09-07 | A low dropout linear regulator |
Country Status (1)
| Country | Link |
|---|---|
| CN (1) | CN100495281C (en) |
Cited By (40)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN101540542B (en) * | 2009-03-18 | 2010-12-01 | 浙江大学 | A single-inductance switch DC voltage converter and four-mode control method |
| CN101916128A (en) * | 2010-08-18 | 2010-12-15 | 北京大学 | A method for improving the output power supply rejection ratio of a bandgap reference source and a corresponding circuit |
| CN102323846A (en) * | 2010-04-05 | 2012-01-18 | 英特赛尔美国股份有限公司 | Accurate current reference with coupling tolerance of high PSRR |
| CN102566637A (en) * | 2010-12-31 | 2012-07-11 | 株式会社理光 | Low-voltage-difference linear voltage stabilizer |
| CN102591393A (en) * | 2012-02-24 | 2012-07-18 | 电子科技大学 | A low dropout linear regulator |
| CN102609025A (en) * | 2012-03-16 | 2012-07-25 | 电子科技大学 | Dynamic current doubling circuit and linear voltage regulator integrated with the circuit |
| CN102650893A (en) * | 2011-02-25 | 2012-08-29 | 株式会社理光 | Low dropout linear regulator |
| CN102929319A (en) * | 2012-10-10 | 2013-02-13 | 清华大学 | Low dropout linear voltage regulator |
| CN103268134A (en) * | 2013-06-03 | 2013-08-28 | 上海宏力半导体制造有限公司 | Low-dropout voltage adjuster capable of improving transient response |
| CN101739050B (en) * | 2008-11-11 | 2013-10-09 | 半导体元件工业有限责任公司 | Low dropout (ldo) voltage regulator and method therefor |
| TWI411903B (en) * | 2010-10-29 | 2013-10-11 | Winbond Electronics Corp | Low drop out voltage regulator |
| CN103677038A (en) * | 2012-09-18 | 2014-03-26 | 株式会社理光 | Low-dropout regulator |
| CN103795240A (en) * | 2009-10-02 | 2014-05-14 | 电力集成公司 | Method and apparatus for implementing slew rate control using bypass capacitor |
| CN104063003A (en) * | 2014-06-27 | 2014-09-24 | 合肥工业大学 | Low-power dissipation off-chip-capacitor-free LDO (Low Dropout Regulator) integrating slew rate enhancing circuit |
| CN104076854A (en) * | 2014-06-27 | 2014-10-01 | 电子科技大学 | Capless LDO (Low Dropout Regulator) |
| TWI468894B (en) * | 2012-06-13 | 2015-01-11 | Elite Semiconductor Esmt | Low dropout regulator with improved transient response |
| CN104756032A (en) * | 2012-11-06 | 2015-07-01 | 高通股份有限公司 | Method and apparatus reduced switch-on rate low dropout regulator (LDO) bias and compensation |
| CN105334900A (en) * | 2015-11-19 | 2016-02-17 | 成都华微电子科技有限公司 | Fast transient response low-dropout linear voltage regulator |
| CN105446403A (en) * | 2014-08-14 | 2016-03-30 | 登丰微电子股份有限公司 | Low dropout linear regulator |
| CN105474118A (en) * | 2013-08-21 | 2016-04-06 | 桑迪士克科技股份有限公司 | Active regulator wake-up time improvement by capacitive regulation |
| CN106155162A (en) * | 2016-08-09 | 2016-11-23 | 电子科技大学 | A kind of low pressure difference linear voltage regulator |
| CN106325344A (en) * | 2015-06-29 | 2017-01-11 | 展讯通信（上海）有限公司 | A low voltage difference voltage stabilizer circuit with an auxiliary circuit |
| CN106896857A (en) * | 2017-03-16 | 2017-06-27 | 西安电子科技大学 | It is applied to the load transient response enhancing circuit of linear voltage regulator |
| CN107092296A (en) * | 2017-04-28 | 2017-08-25 | 成都华微电子科技有限公司 | A kind of fast transient response low-voltage difference adjustor |
| CN107102667A (en) * | 2016-02-22 | 2017-08-29 | 联发科技（新加坡）私人有限公司 | Low pressure difference linear voltage regulator |
| CN108491020A (en) * | 2018-06-08 | 2018-09-04 | 长江存储科技有限责任公司 | Low-dropout regulator and flash memory |
| CN108664065A (en) * | 2017-03-31 | 2018-10-16 | 中芯国际集成电路制造(上海)有限公司 | Bias current generating circuit |
| CN108776506A (en) * | 2018-06-25 | 2018-11-09 | 电子科技大学 | A kind of low pressure difference linear voltage regulator of high stability |
| CN109710017A (en) * | 2019-02-12 | 2019-05-03 | 麦堆微电子技术（上海）有限公司 | A kind of low pressure difference linear voltage regulator system |
| CN111522383A (en) * | 2020-05-20 | 2020-08-11 | 上海维安半导体有限公司 | Dynamic bias current boosting method applied to ultra-low power LDO (low dropout regulator) |
| CN112833921A (en) * | 2020-12-31 | 2021-05-25 | 广州导远电子科技有限公司 | A single-axis gyroscope circuit |
| CN113285670A (en) * | 2020-12-30 | 2021-08-20 | 思瑞浦微电子科技（苏州）股份有限公司 | Low frequency offset RC oscillator |
| CN113867466A (en) * | 2021-10-14 | 2021-12-31 | 上海安路信息科技股份有限公司 | Voltage stabilizer circuit |
| CN113992165A (en) * | 2021-10-29 | 2022-01-28 | 上海华虹宏力半导体制造有限公司 | A radio frequency amplifier with adjustable offset and gain |
| CN114035646A (en) * | 2021-10-26 | 2022-02-11 | 北京理工大学 | A Dynamic Bias Circuit of Linear Regulator |
| CN114527820A (en) * | 2022-02-15 | 2022-05-24 | 珠海全志科技股份有限公司 | Voltage stabilizer circuit and electronic equipment |
| CN114879792A (en) * | 2022-05-24 | 2022-08-09 | 中国人民解放军国防科技大学 | A Dual-Loop Low Dropout Linear Regulator with Inverted Voltage Follower Structure |
| CN115756057A (en) * | 2022-11-23 | 2023-03-07 | 西安电子科技大学 | Transient response enhancement type LDO linear voltage regulator |
| CN115840483A (en) * | 2022-11-18 | 2023-03-24 | 西安电子科技大学 | Low dropout regulator with transient enhancement characteristic |
| US20230205245A1 (en) * | 2021-12-29 | 2023-06-29 | Silego Technology Inc. | Integrated user programmable slew-rate controlled soft-start for ldo |
-
2007
- 2007-09-07 CN CNB2007101214826A patent/CN100495281C/en not_active Expired - Fee Related
Cited By (67)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| CN101739050B (en) * | 2008-11-11 | 2013-10-09 | 半导体元件工业有限责任公司 | Low dropout (ldo) voltage regulator and method therefor |
| CN101540542B (en) * | 2009-03-18 | 2010-12-01 | 浙江大学 | A single-inductance switch DC voltage converter and four-mode control method |
| CN103795240A (en) * | 2009-10-02 | 2014-05-14 | 电力集成公司 | Method and apparatus for implementing slew rate control using bypass capacitor |
| CN103795240B (en) * | 2009-10-02 | 2016-09-14 | 电力集成公司 | For using bypass capacitor to realize the method and apparatus that slew rate controls |
| CN102323846A (en) * | 2010-04-05 | 2012-01-18 | 英特赛尔美国股份有限公司 | Accurate current reference with coupling tolerance of high PSRR |
| CN102323846B (en) * | 2010-04-05 | 2014-12-03 | 英特赛尔美国股份有限公司 | Coupling tolerant precise current reference with high PSRR |
| CN101916128A (en) * | 2010-08-18 | 2010-12-15 | 北京大学 | A method for improving the output power supply rejection ratio of a bandgap reference source and a corresponding circuit |
| CN101916128B (en) * | 2010-08-18 | 2012-07-25 | 北京大学 | Method and corresponding circuit for improving output power supply rejection ratio of band-gap reference source |
| TWI411903B (en) * | 2010-10-29 | 2013-10-11 | Winbond Electronics Corp | Low drop out voltage regulator |
| CN102566637B (en) * | 2010-12-31 | 2014-05-07 | 株式会社理光 | Low-voltage-difference linear voltage stabilizer and method for adjusting low-voltage-difference linear voltage stabilizer |
| CN102566637A (en) * | 2010-12-31 | 2012-07-11 | 株式会社理光 | Low-voltage-difference linear voltage stabilizer |
| CN102650893A (en) * | 2011-02-25 | 2012-08-29 | 株式会社理光 | Low dropout linear regulator |
| CN102650893B (en) * | 2011-02-25 | 2014-09-17 | 株式会社理光 | Low dropout linear regulator |
| CN102591393B (en) * | 2012-02-24 | 2013-09-25 | 电子科技大学 | Low-dropout linear regulator |
| CN102591393A (en) * | 2012-02-24 | 2012-07-18 | 电子科技大学 | A low dropout linear regulator |
| CN102609025A (en) * | 2012-03-16 | 2012-07-25 | 电子科技大学 | Dynamic current doubling circuit and linear voltage regulator integrated with the circuit |
| TWI468894B (en) * | 2012-06-13 | 2015-01-11 | Elite Semiconductor Esmt | Low dropout regulator with improved transient response |
| CN103677038A (en) * | 2012-09-18 | 2014-03-26 | 株式会社理光 | Low-dropout regulator |
| CN102929319B (en) * | 2012-10-10 | 2014-09-10 | 清华大学 | Low dropout linear voltage regulator |
| CN102929319A (en) * | 2012-10-10 | 2013-02-13 | 清华大学 | Low dropout linear voltage regulator |
| CN104756032B (en) * | 2012-11-06 | 2016-11-09 | 高通股份有限公司 | Method and apparatus for reducing on-rate low dropout regulator (LDO) bias and compensation |
| CN104756032A (en) * | 2012-11-06 | 2015-07-01 | 高通股份有限公司 | Method and apparatus reduced switch-on rate low dropout regulator (LDO) bias and compensation |
| CN103268134A (en) * | 2013-06-03 | 2013-08-28 | 上海宏力半导体制造有限公司 | Low-dropout voltage adjuster capable of improving transient response |
| CN103268134B (en) * | 2013-06-03 | 2015-08-19 | 上海华虹宏力半导体制造有限公司 | The low difference voltage regulator of transient response can be improved |
| CN105474118A (en) * | 2013-08-21 | 2016-04-06 | 桑迪士克科技股份有限公司 | Active regulator wake-up time improvement by capacitive regulation |
| CN104076854B (en) * | 2014-06-27 | 2016-02-03 | 电子科技大学 | A kind of without electric capacity low pressure difference linear voltage regulator |
| CN104076854A (en) * | 2014-06-27 | 2014-10-01 | 电子科技大学 | Capless LDO (Low Dropout Regulator) |
| CN104063003B (en) * | 2014-06-27 | 2015-10-21 | 合肥工业大学 | A kind of low-power consumption of integrated slew rate enhancing circuit is without the outer electric capacity LDO of sheet |
| CN104063003A (en) * | 2014-06-27 | 2014-09-24 | 合肥工业大学 | Low-power dissipation off-chip-capacitor-free LDO (Low Dropout Regulator) integrating slew rate enhancing circuit |
| CN105446403A (en) * | 2014-08-14 | 2016-03-30 | 登丰微电子股份有限公司 | Low dropout linear regulator |
| US9575499B2 (en) | 2014-08-14 | 2017-02-21 | Green Solution Technology Co., Ltd. | Low-dropout voltage regulator |
| CN106325344B (en) * | 2015-06-29 | 2018-01-26 | 展讯通信（上海）有限公司 | Low-dropout regulator circuit with auxiliary circuit |
| CN106325344A (en) * | 2015-06-29 | 2017-01-11 | 展讯通信（上海）有限公司 | A low voltage difference voltage stabilizer circuit with an auxiliary circuit |
| CN105334900B (en) * | 2015-11-19 | 2016-11-30 | 成都华微电子科技有限公司 | Fast transient response low pressure difference linear voltage regulator |
| CN105334900A (en) * | 2015-11-19 | 2016-02-17 | 成都华微电子科技有限公司 | Fast transient response low-dropout linear voltage regulator |
| CN107102667A (en) * | 2016-02-22 | 2017-08-29 | 联发科技（新加坡）私人有限公司 | Low pressure difference linear voltage regulator |
| CN106155162A (en) * | 2016-08-09 | 2016-11-23 | 电子科技大学 | A kind of low pressure difference linear voltage regulator |
| CN106896857A (en) * | 2017-03-16 | 2017-06-27 | 西安电子科技大学 | It is applied to the load transient response enhancing circuit of linear voltage regulator |
| CN106896857B (en) * | 2017-03-16 | 2018-04-17 | 西安电子科技大学 | Load transient response applied to linear voltage regulator strengthens circuit |
| CN108664065A (en) * | 2017-03-31 | 2018-10-16 | 中芯国际集成电路制造(上海)有限公司 | Bias current generating circuit |
| CN107092296B (en) * | 2017-04-28 | 2019-05-21 | 成都华微电子科技有限公司 | A kind of fast transient response low-voltage difference adjustor |
| CN107092296A (en) * | 2017-04-28 | 2017-08-25 | 成都华微电子科技有限公司 | A kind of fast transient response low-voltage difference adjustor |
| CN108491020A (en) * | 2018-06-08 | 2018-09-04 | 长江存储科技有限责任公司 | Low-dropout regulator and flash memory |
| CN108491020B (en) * | 2018-06-08 | 2024-06-07 | 长江存储科技有限责任公司 | Low dropout voltage regulator and flash memory |
| CN108776506A (en) * | 2018-06-25 | 2018-11-09 | 电子科技大学 | A kind of low pressure difference linear voltage regulator of high stability |
| CN109710017A (en) * | 2019-02-12 | 2019-05-03 | 麦堆微电子技术（上海）有限公司 | A kind of low pressure difference linear voltage regulator system |
| CN109710017B (en) * | 2019-02-12 | 2024-03-19 | 麦堆微电子技术（上海）有限公司 | Low-dropout linear voltage regulator system |
| CN111522383A (en) * | 2020-05-20 | 2020-08-11 | 上海维安半导体有限公司 | Dynamic bias current boosting method applied to ultra-low power LDO (low dropout regulator) |
| CN113285670A (en) * | 2020-12-30 | 2021-08-20 | 思瑞浦微电子科技（苏州）股份有限公司 | Low frequency offset RC oscillator |
| CN113285670B (en) * | 2020-12-30 | 2022-12-30 | 思瑞浦微电子科技（苏州）股份有限公司 | Low frequency offset RC oscillator |
| CN112833921A (en) * | 2020-12-31 | 2021-05-25 | 广州导远电子科技有限公司 | A single-axis gyroscope circuit |
| CN113867466A (en) * | 2021-10-14 | 2021-12-31 | 上海安路信息科技股份有限公司 | Voltage stabilizer circuit |
| CN113867466B (en) * | 2021-10-14 | 2023-03-14 | 上海安路信息科技股份有限公司 | Voltage stabilizer circuit |
| CN114035646A (en) * | 2021-10-26 | 2022-02-11 | 北京理工大学 | A Dynamic Bias Circuit of Linear Regulator |
| CN114035646B (en) * | 2021-10-26 | 2024-09-27 | 北京理工大学 | Dynamic bias circuit of linear voltage stabilizer |
| CN113992165A (en) * | 2021-10-29 | 2022-01-28 | 上海华虹宏力半导体制造有限公司 | A radio frequency amplifier with adjustable offset and gain |
| CN113992165B (en) * | 2021-10-29 | 2025-04-04 | 上海华虹宏力半导体制造有限公司 | A radio frequency amplifier with adjustable bias and gain |
| US20230205245A1 (en) * | 2021-12-29 | 2023-06-29 | Silego Technology Inc. | Integrated user programmable slew-rate controlled soft-start for ldo |
| US11914409B2 (en) * | 2021-12-29 | 2024-02-27 | Silego Technology Inc. | Integrated user programmable slew-rate controlled soft-start for LDO |
| CN114527820B (en) * | 2022-02-15 | 2024-04-12 | 珠海全志科技股份有限公司 | Voltage stabilizer circuit and electronic equipment |
| CN114527820A (en) * | 2022-02-15 | 2022-05-24 | 珠海全志科技股份有限公司 | Voltage stabilizer circuit and electronic equipment |
| CN114879792B (en) * | 2022-05-24 | 2024-04-19 | 中国人民解放军国防科技大学 | A dual-loop low-dropout linear regulator with an inverting voltage follower structure |
| CN114879792A (en) * | 2022-05-24 | 2022-08-09 | 中国人民解放军国防科技大学 | A Dual-Loop Low Dropout Linear Regulator with Inverted Voltage Follower Structure |
| CN115840483B (en) * | 2022-11-18 | 2024-09-20 | 西安电子科技大学 | Low dropout linear voltage regulator with transient enhancement characteristic |
| CN115840483A (en) * | 2022-11-18 | 2023-03-24 | 西安电子科技大学 | Low dropout regulator with transient enhancement characteristic |
| CN115756057A (en) * | 2022-11-23 | 2023-03-07 | 西安电子科技大学 | Transient response enhancement type LDO linear voltage regulator |
| CN115756057B (en) * | 2022-11-23 | 2024-08-09 | 西安电子科技大学 | Transient response enhanced LDO linear voltage stabilizer |
Also Published As
| Publication number | Publication date |
|---|---|
| CN100495281C (en) | 2009-06-03 |
Similar Documents
| Publication | Publication Date | Title |
|---|---|---|
| CN100495281C (en) | A low dropout linear regulator | |
| US10019023B2 (en) | Low-dropout linear regulator with super transconductance structure | |
| CN101957628B (en) | Self-adaption zero-frequency compensation circuit in low-voltage difference linear voltage regulator | |
| CN100527039C (en) | Low pressure difference linearity voltage stabilizer for enhancing performance by amplifier embedded compensation network | |
| US7323853B2 (en) | Low drop-out voltage regulator with common-mode feedback | |
| CN101667046B (en) | Low-dropout voltage regulator | |
| US8866457B2 (en) | Voltage regulator | |
| KR100991699B1 (en) | Constant voltage circuit and its operation control method | |
| EP1378808A1 (en) | LDO regulator with wide output load range and fast internal loop | |
| US8427122B2 (en) | Enhancement of power supply rejection for operational amplifiers and voltage regulators | |
| CN111176358B (en) | Low-power-consumption low-dropout linear voltage regulator | |
| KR101238173B1 (en) | A Low Dropout Regulator with High Slew Rate Current and High Unity-Gain Bandwidth | |
| JP2009116679A (en) | Linear regulator circuit, linear regulation method, and semiconductor device | |
| US7113044B2 (en) | Precision current mirror and method for voltage to current conversion in low voltage applications | |
| CN107024958B (en) | A kind of linear voltage-stabilizing circuit with fast load transient response | |
| CN113467559B (en) | Adaptive dynamic zero compensation circuit applied to LDO (low dropout regulator) | |
| CN110366713B (en) | Method and circuit system for compensating low dropout linear regulator | |
| CN114356008B (en) | Low-dropout linear voltage regulator | |
| CN201464838U (en) | Self-adaptation zero-frequency compensating circuit of low-dropout linear voltage regulator | |
| CN113342110A (en) | Error amplifier circuit with dynamic zero compensation | |
| US9128505B2 (en) | Voltage regulator circuit | |
| CN215376184U (en) | Quick response low dropout regulator circuit | |
| CN119088162A (en) | A digitally assisted analog LDO circuit without external capacitor | |
| US6812778B1 (en) | Compensating capacitive multiplier | |
| US7786802B2 (en) | Output stage circuit and operational amplifier thereof |
Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| C06 | Publication | ||
| PB01 | Publication | ||
| C10 | Entry into substantive examination | ||
| SE01 | Entry into force of request for substantive examination | ||
| C14 | Grant of patent or utility model | ||
| GR01 | Patent grant | ||
| CF01 | Termination of patent right due to non-payment of annual fee | ||
| CF01 | Termination of patent right due to non-payment of annual fee |
Granted publication date: 20090603 Termination date: 20210907 |