CN101120514B - Wireless communication method, base station and wireless communication device - Google Patents

Abstract

A method and system for interference cancellation (IC). One aspect relates to traffic interference cancellation. Another aspect relates to joint IC for pilot, overhead and data. Another aspect relates to improved channel estimation. Another aspect relates to adaptation of transmit subchannel gains.

Description

Wireless communication method, base station, and wireless communication device

Claiming priority pursuant to 35U.S.C. § 119

This application claims priority to commonly assigned U.S. provisional application No.60/638,666, entitled "TRAFFICINTERFERENCE CANCELLATION AT THE BTS ON A CDMAREVERSE LINK," filed on 23/12/2004, and U.S. provisional application No.60/638,666 is incorporated herein by reference.

Technical Field

The present disclosure relates generally to wireless communication systems, and more particularly to traffic interference cancellation in wireless communication systems.

Technical Field

A communication system may provide communication between base stations and access terminals. The forward link or downlink refers to transmission from the base station to the access terminal. The reverse link or uplink refers to transmission from the access terminal to the base station. Each access terminal may communicate with one or more base stations on the forward and reverse links at a given moment based on whether the access terminal is in an active state and whether the access terminal is in a soft handoff state.

Drawings

The features, nature, and advantages of the present invention will become more apparent from the detailed description set forth below when taken in conjunction with the drawings. The same reference numbers and symbols may be used throughout the drawings to refer to the same or like parts.

Fig. 1 illustrates a wireless communication system with a base station and an access terminal.

Fig. 2 illustrates an example of a transmitter structure and/or process that may be implemented in the access terminal of fig. 1.

Fig. 3 illustrates an example of receiver processing and/or structure that may be implemented in the base station of fig. 1.

Fig. 4 illustrates another embodiment of a base station receiver process or structure.

Fig. 5 shows a general example of power allocation for three users in the system of fig. 1.

Fig. 6 shows an example of a uniform time offset allocation case for frame asynchronous traffic interference cancellation for users with equal transmit power.

Fig. 7 shows an interleaving structure for the reverse link data packet and the forward link automatic repeat request channel.

Fig. 8 shows the memory grouped across all 16 slots.

Fig. 9A shows an exemplary traffic interference cancellation method of Sequential Interference Cancellation (SIC) without delay decoding.

Fig. 9B illustrates an apparatus for performing the method of fig. 9A.

Fig. 10 shows the receiver sample buffer after arrival of successive subpackets of an interlace with interference cancellation of decoded subpackets.

Fig. 11 shows an overhead channel structure.

Fig. 12A illustrates a method of first performing pilot ic (pic) and then performing overhead ic (oic) and traffic ic (tic) together.

Fig. 12B illustrates an apparatus for performing the method of fig. 12A.

Fig. 13A shows a variation of the method in fig. 12A.

Fig. 13B illustrates an apparatus for performing the method of fig. 13A.

Fig. 14A illustrates a method for performing joint PIC, OIC, and TIC.

Fig. 14B illustrates an apparatus for performing the method of fig. 14A.

Fig. 15A shows a variation of the method of fig. 14A.

Fig. 15B illustrates an apparatus for performing the method of fig. 15A.

Fig. 16 shows a model of a transmission system.

Fig. 17 shows an example response of combined transmit and receive filtering.

Fig. 18A and 18B show examples of channel estimates (real and imaginary parts) from an estimated multipath channel at each of three RAKE fingers.

Fig. 19A-19B illustrate an example of improved channel estimation based on RAKE fingers and despreading with data chips.

Fig. 20A illustrates a method of despreading at a RAKE finger using regenerated data chips.

Fig. 20B illustrates an apparatus for performing the method of fig. 20A.

Fig. 21A and 21B show an example of estimating a composite channel at chipX2 resolution using evenly spaced samples.

Fig. 22A illustrates a method of estimating a composite channel with uniform resolution using regenerated data chips.

Fig. 22B illustrates an apparatus for performing the method of fig. 22A.

Fig. 23 illustrates closed loop power control and gain control using fixed overhead subchannel gains.

Fig. 24 illustrates variations of the power control and gain control of fig. 23 using fixed overhead subchannel gains.

Fig. 25 shows an example of power control using fixed overhead subchannel gains.

Fig. 26 is the same as fig. 24 except for overhead gain control.

Fig. 27 shows a variation of fig. 26 for the overhead gain control using DRC only.

Detailed Description

Any embodiment described herein is not necessarily preferred or advantageous over other embodiments. While the various aspects of the present application are illustrated in the accompanying drawings, the drawings are not necessarily drawn to scale or to be exhaustive.

Fig. 1 illustrates a wireless communication system 100 that includes a system controller 102, base stations 104a-104b, and a plurality of access terminals 106a-106 h. System 100 may have any number of controllers 102, base stations 104, and access terminals 106. Aspects and embodiments of the present application described below may be implemented in the system 100.

The access terminals 106 may be mobile or stationary and may be distributed throughout the communication system 100 of fig. 1. The access terminal 106 may be connected to or implemented in a computing device such as a laptop personal computer. Alternatively, the access terminal may be a stand-alone data device such as a Personal Digital Assistant (PDA). Access terminal 106 may refer to various types of devices such as a wired telephone, a wireless telephone, a cellular telephone, a laptop computer, a wireless communication Personal Computer (PC) card, a PDA, an external or internal modem, and so forth. An access terminal may be any device that provides data to a user by communicating through a wireless channel or through a wired channel, for example using fiber optic or coaxial cables. An access terminal may be referred to by various names, such as mobile station, access unit, subscriber unit, mobile device, mobile terminal, mobile unit, mobile phone, mobile station, remote terminal, remote unit, user device, user equipment, handheld device, etc.

System 100 provides communication for a plurality of cells, where each cell is served by one or more base stations 104. The base station 104 may also be referred to as a Base Transceiver System (BTS), an access point, a portion of an access network, a Modem Pool Transceiver (MPT), or a node B. An access network refers to a network device that provides a data connection between a packet-switched data network (e.g., the internet) and the access terminal 106.

The Forward Link (FL) or downlink refers to transmission from the base station 104 to the access terminal 106. Reverse Link (RL) or uplink refers to transmission from an access terminal 106 to a base station 104.

The base station 104 may transmit data to the access terminal 106 using a data rate selected from a set of different data rates. The access terminal 106 may determine a signal-to-noise-plus-interference ratio (SINR) of a pilot signal transmitted by the base station 104 and determine a data rate required for the base station 104 to transmit data to the access terminal 106. The access terminal 106 may send a data request channel or Data Rate Control (DRC) message to the base station 104 to inform the base station 104 of the required data rate.

A system controller 102, also referred to as a Base Station Controller (BSC), may coordinate and control the base stations 104 and may further control call routing to the access terminals 106 via the base stations 104. The system controller 102 may further be coupled to a Public Switched Telephone Network (PSTN) via a Mobile Switching Center (MSC) and to a packet data network via a Packet Data Serving Node (PDSN).

Communication system 100 may use one or more communication techniques such as Code Division Multiple Access (CDMA), IS-95, High Rate Packet Data (HRPD) (also known as High Data Rate (HDR), as defined in "CDMA 2000high Rate Packet Data Air interference specification," TIA/EIA/IS-856), CDMA1x evolution Data optimized (EV-DO), 1xEV-DV, Wideband CDMA (WCDMA), Universal Mobile Telecommunications System (UMTS), time division synchronous CDMA (TD-SCDMA), Orthogonal Frequency Division Multiplexing (OFDM), and so forth. The following described examples have been given to illustrate the details for the purpose of understanding. The ideas presented in this application are also applicable to other systems and these examples are not intended to limit this application.

Fig. 2 illustrates an example of a transmitter structure and/or process that may be implemented in the access terminal 106 of fig. 1. The functions and components shown in fig. 2 may be implemented in software, hardware, or a combination of software and hardware. Other functions may be added to fig. 2 in addition to or in place of those shown in fig. 2.

A data source 200 provides data to an encoder 202, and the encoder 202 encodes data bits using one or more coding schemes to provide coded data chips. Each coding scheme may include one or more types of coding, such as Cyclic Redundancy Check (CRC), convolutional, Turbo, block, or other types of coding, or no coding at all. Other coding schemes may include automatic repeat request (ARQ), hybrid repeat request (H-ARQ), and incremental redundancy repeat techniques. Different types of data may be encoded using different encoding schemes. Interleaver 204 interleaves the encoded data bits to overcome fading effects.

The modulator 206 modulates the encoded, interleaved data to generate modulated data. Examples of modulation techniques include Binary Phase Shift Keying (BPSK) and Quadrature Phase Shift Keying (QPSK). The modulator 206 may also repeat the sequence of modulated data or a symbol puncturing unit may puncture the symbol bits. Modulator 206 may also spread the modulated data using Walsh cover (i.e., Walsh codes) to form data chips. Modulator 206 may also time-division multiplex data chips using pilot chips and MAC chips to form a stream of chips. The modulator 206 may also use a pseudo-random noise (PN) spreading code to spread the chip stream using one or more PN codes (e.g., short code, long code).

The baseband-to-Radio Frequency (RF) conversion unit 208 may convert baseband signals to RF signals for transmission over a communication link to one or more base stations 104 via an antenna 210.

Fig. 3 illustrates an example of receiver processing and/or structure that may be implemented in the base station 104 of fig. 1. The functions and components shown in fig. 3 may be implemented in software, hardware, or a combination of software and hardware. Other functions may be added to fig. 3 in addition to or in place of the functions shown in fig. 3.

One or more antennas 300 receive reverse link modulated signals from one or more access terminals 106. Multiple antennas may achieve spatial diversity against adverse path effects, such as fading. Each received signal is provided to a respective receiver or RF-to-baseband conversion unit 302 for conditioning (e.g., filtering, amplifying, downconverting) and digitizing the received signal to generate data samples corresponding to the received signal.

Demodulator 304 may demodulate the received signal to provide recovered symbols. For CDMA2000, demodulation attempts to recover the data transmission by: (1) channelizing the despread samples to separate or channelize the received data and pilot signals onto their respective code channels; (2) the channelized data is coherently demodulated using the recovered pilot signal to provide demodulated data. Demodulator 304 may include a receive sample buffer 312 (also referred to as a common front-end RAM (feram) or sample RAM) for storing received signal samples for all user/access terminals, a Rake receiver 314 for despreading and processing various signal instances, and a demodulation symbol buffer 316 (also referred to as a back-end RAM (beram) or demodulation symbol RAM). There may be multiple demodulated symbol buffers 316 corresponding to multiple users/access terminals.

A deinterleaver 306 deinterleaves the data from the demodulator 304.

Decoder 308 can decode the demodulated data to recover the decoded data bits sent by access terminal 106. The decoded data may be provided to a data sink 310.

Fig. 4 illustrates another embodiment of a base station receiver process or structure. In fig. 4, the data bits of successfully decoded users are input to a reconstruction unit 400 including an encoder 402, an interleaver 404, a modulator 406, and a filter 408. The encoder 402, interleaver 404, and modulator 406 may be the same as the encoder 202, interleaver 204, and modulator 206 of fig. 2. Filter 408 forms samples of the decoded user at FERAM resolution, e.g., from code rate to 2x code rate. Then, the influence of the decoded user on the FERAM is removed or eliminated from the FERAM 312.

Although interference cancellation at the base station 104 is described below, the concepts of the present application are also applicable to the access terminal 106 or any other component of the communication system.

Traffic interference cancellation

The capacity of the CDMA reverse link may be limited by interference between users because the signals transmitted by different users are not orthogonal at the BTS 104. Therefore, techniques to reduce interference between users will improve system performance for the CDMA reverse link. Techniques are described herein for efficiently implementing interference cancellation for advanced CDMA systems, such as CDMA20001xEV-DO RevA.

Each DO RevA user transmits traffic, pilot, and overhead signals, all of which cause interference to other users. As shown in fig. 4, the signal may be reconstructed at the BTS104 and subtracted from the front-end RAM 312. The transmitted pilot signal is known at the BTS104 and can be reconstructed based on knowledge of the channel. However, the overhead signals (e.g., Reverse Rate Indicator (RRI), data request channel or Data Rate Control (DRC), Data Source Channel (DSC), Acknowledgement (ACK)) are first demodulated and detected, and then the transmitted data signals are demodulated, deinterleaved, and decoded at the BTS104 to determine the transmitted overhead and traffic chips. Based on determining the transmit chips for a given signal, the reconstruction unit 400 may then reconstruct the impact on the FERAM312 based on the channel knowledge.

Encoder 202, interleaver 204, and/or modulator 206 may repeat and process data packet bits from data source 200 into a plurality of corresponding "subpackets" for transmission to base station 104. If the base station 104 receives a signal with a high signal-to-noise ratio, the first sub-packet may include enough information so that the base station 104 can decode and derive the original data packet. For example, a data packet from the data source 200 may be repeated and processed into four sub-packets. The user terminal 106 transmits the first subpacket to the base station 104. The probability that the base station 104 correctly decodes and derives the original data packet from the first received sub-packet may be relatively low. However, because the base station 104 receives the second, third and fourth subpackets and combines the information derived from the received subpackets, the probability of decoding and deriving the original data packet is increased. Once the base station 104 correctly decodes the original packet (e.g., using a Cyclic Redundancy Check (CRC) or other error detection technique), the base station 104 sends an acknowledgement signal to the user terminal 106 to stop transmitting the subpackets. The user terminal 106 may then transmit the first subpacket of the new packet.

The reverse link of DO-RevA employs H-ARQ (fig. 7), in which each 16-slot packet is divided into 4 subpackets and transmitted using an interlace with 8 slots between subpackets of the same interlace. In addition, different users/access terminals 106 may start their transmissions on different slot boundaries, so that the 4-slot sub-packets for different users arrive at the BTS asynchronously. The following describes the effects of asynchrony and the efficient design of an interference cancellation receiver for H-ARQ and CDMA.

The benefit of interference cancellation depends on the order in which signals are removed from the FERAM 312. The techniques described herein involve decoding users according to traffic-to-pilot (T2P) ratio, effective SINR, or decoding probability (and, if CRC passes, subtracting). Various methods are disclosed for re-attempting to demodulate and decode the user after other signals have been removed from the FERAM 312. Interference cancellation by BTS FERAM312 can be performed efficiently to account for asynchronous CDMA systems such as EV-DO RevA, where users transmit pilot, control, and traffic signals using hybrid ARQ. The application is also applicable to EV-DV Rel D, W-CDMA EUL and CDMA 2000.

Traffic Interference Cancellation (TIC) may be defined as subtractive interference cancellation that removes the effect of a user's data on the FERAM312 (fig. 4) after the user has successfully decoded. The present application addresses some practical problems related to TIC of practical CDMA systems such as CDMA2000, EV-DO, EV-DV, and WCDMA. Many of these problems are caused by the fact that real systems have both user asynchrony and hybrid ARQ. For example, CDMA2000 intentionally spreads user data frames uniformly in time, thereby preventing excessive delays in the backbone network. RevA for EV-DO, Rel D for EV-DV, and EUL for WCDMA also use hybrid ARQ which introduces more than one possible data length.

Multi-user detection is the dominant algorithm in case of TIC degradation and refers to any algorithm that improves performance by making the detections of two different users interact. TIC methods may involve a mix of successive interference cancellation (also known as sequential interference cancellation or SIC) and parallel interference cancellation. "successive interference cancellation" refers to any algorithm that decodes users in order and uses the previously decoded user's data to improve performance. "parallel interference cancellation" generally refers to decoding users simultaneously and subtracting all decoded users simultaneously.

TIC may be different from Pilot Interference Cancellation (PIC). One difference between TIC and PIC is that the transmitted pilot signal is fully known in advance by the receiver. Therefore, the PIC may subtract the pilot contribution of the received signal using only the channel estimate. The second major difference is that the transmitter and receiver interact tightly on the traffic channel through the H-ARQ mechanism. The receiver does not know the transmitted data order before successfully decoding the user.

Also, a technique called Overhead Interference Cancellation (OIC) needs to be employed to remove the overhead channels from the front-end RAM. The overhead channels cannot be removed until the BTS104 knows the transmitted overhead data, depending on decoding and then reassembling the overhead messages.

Successive interference cancellation defines a class of methods. The chain rule of mutual information shows that, in an ideal situation, successive interference cancellation can achieve the capacity of a multiple access channel. The main condition for this is that all users are frame synchronized and the error estimated by the channel of each user is negligible.

Fig. 5 shows a general example of power allocation for three users (user 1, user 2, user 3), where the users transmit frames synchronously (frames from all users are received simultaneously) and each user transmits at the same data rate. Instructing each user to use a particular transmit power, e.g., user 3 transmits with a power substantially equal to the noise; user 2 transmits with a power substantially equal to the power of user 3 plus noise; and user 1 transmits with a power substantially equal to user 2 plus user 3 plus noise.

The receiver processes the signals in descending order of transmit power. Starting with k-1 (user 1 with the highest power), the receiver attempts to decode user 1. If the decoding is successful, its effect on the received signal is formed and subtracted from the channel estimate for user 1. This may be referred to as synchronous sequential interference cancellation. The receiver continues until decoding has been attempted for all users. Each user has the same SINR after interference cancellation of the successive interference cancellation of the previously decoded user.

However, this method is very sensitive to decoding errors. If a single, higher power user (e.g., user 1) is not decoded correctly, then the signal-to-interference-plus-noise ratio (SINR) of all users may be severely degraded. This makes it impossible for all users after that time to be decoded. Another drawback of this approach is that it requires the user to have a certain relative power at the receiver, which is difficult to ensure in a fading channel.

Frame asynchrony and interference cancellation, e.g. cdma2000

It is assumed that the user frame offsets are intentionally staggered with respect to each other. Overall, such frame asynchronous operation has many benefits to the system. For example, the processing power and network bandwidth at the receiver will then have a more uniform usage over time. In contrast, frame asynchrony between users requires a burst of processing power and network resources at the end of each frame boundary, since all users will complete a packet at the same time. In the case of frame asynchrony, the BTS104 may decode the user with the earliest arrival time instead of the user with the largest power.

Fig. 6 shows an example of a uniform time offset allocation case for frame asynchronous TIC for users with equal transmit power. Fig. 6 depicts a snapshot of the time just before frame 1 of user 1 is to be decoded. Since frame 0 has been decoded and cancelled for all users, its impact on interference is shown crosswise (users 2 and 3). Typically, this approach reduces interference by a factor of 2. Half of the interference has been removed by TIC before decoding user 1's frame 1.

In another embodiment, the users in fig. 6 may refer to user groups, e.g., user group 1, user group 2, user group 3.

The benefit of asynchronous and interference cancellation is relative symmetry between users in terms of power level and error statistics if they want similar data rates. Typically, for sequential interference cancellation with equal user data rates, the last user is received with very low power and also relies heavily on successful decoding of all previous users.

Asynchronous, hybrid ARQ, and interleaving, e.g. EV-DO RevA

FIG. 7 shows an interleaving structure (e.g., in 1xEV-DO RevA) for RL data packets and FL ARQ channels. Each interlace (interlace 1, interlace 2, interlace 3) includes a set of time-interleaved segments. In this example, each segment is 4 slots long. During each segment, the user terminal may transmit a subpacket to the base station. There are three interlaces and each segment is four slots long. Thus, there are eight time slots between the end of a sub-packet for a given interlace and the start of the next packet for the same interlace. This gives the receiver enough time to decode the subpacket and relay an ACK or Negative Acknowledgement (NAK) to the transmitter.

Hybrid ARQ takes advantage of the time-varying nature of fading channels. If the channel conditions are good for the first 1, 2 or 3 subpackets, the data frame may be decoded using only those subpackets and the receiver sends an ACK to the transmitter. The ACK causes the transmitter not to send the remaining subpackets, but to start a new packet, if needed.

Receiver architecture for interference cancellation

By using TIC, the data of the decoded user is reconstructed and subtracted (fig. 4), and thus interference of the data of the decoded user to other users can be removed. The TIC receiver may be equipped with two circular memories: FERAM312 and BERAM 316.

The FERAM312 stores the received samples (e.g., at a 2x chip rate) and is common to all users. A receiver without TIC will only use FERAM for approximately 1-2 slots (to accommodate delays in the demodulation process) since no traffic or overhead interference is subtracted. In a TIC receiver for a system with H-ARQ functionality, the FERAM312 may span many slots, e.g., 40, and be updated by TIC by subtracting the interference of the decoding user. In another configuration, the FERAM312 may be of a length that spans less than all of the packet, e.g., a length that spans a time period from a beginning of a sub-packet of the packet to an end of a subsequent sub-packet of the packet.

BERAM316 stores demodulated symbols of received bits generated by Rake receiver 314 of the demodulator. Each user may have a different BERAM because the demodulated symbols are obtained by despreading and combining over the RAKE fingers using the user-specific PN sequence. BERAM316 may be used by both TIC and non-TIC receivers. When the FERAM312 does not span all sub-packets, the FERAM 316 of TIC is employed to store demodulated symbols of previous sub-packets that are no longer stored in the FERAM 312. The BERAM316 may be updated whenever an attempt to decode occurs or whenever a slot exists in the FERAM 312.

Method for selecting FERAM length

The size of the BERAM316 and the FERAM312 may be selected based on various tradeoffs between required processing power, transmission bandwidth from memory to processor, latency, and system performance. Generally, if a shorter FERAM312 is used, the benefits of TIC will be limited because the first subpacket is not updated. On the other hand, a shorter FERAM312 results in a reduced amount of demodulation, puncturing, and a smaller transmission bandwidth.

With RevA interleaving, a 16 slot packet (four subpackets, each transmitted in 4 slots) will span 40 slots. Thus, a FERAM of 40 slots may be used to ensure that users are removed from all affected slots.

FIG. 8 shows the FERAM312 for EV-DO RevA spanning 40 slots of a full 16-slot packet. Whenever a new packet is received, it is attempted to decode the packet using all available subpackets stored in the FERAM 312. If the decoding is successful, the effect of the packet is cancelled from the FERAM312 by reconstructing and subtracting the effect of all sub-packets (1, 2, 3 or 4). For DO-RevA, FERAM lengths of 4, 16, 28 or 40 slots would span 1, 2, 3 or 4 subpackets, respectively. The length of FERAM implemented in the receiver may depend on complexity considerations, the need to support various user arrival times, and the ability to re-demodulate and decode the user on the previous frame offset.

Fig. 9A shows a general method of TIC of an example of Sequential Interference Cancellation (SIC) without delayed decoding. Other modifications will be described below. The process begins at start block 900 and proceeds to a choose delay block 902. In SIC, the choose delay block 902 may be skipped. In block 903, the BTS104 selects a user (or group of users) from those that terminate the subpacket in the current slot.

In block 904, the demodulator 304 decodes the samples of the selected user's subpackets for some or all of the time segments stored in the FERAM312 according to the user's spreading and scrambling sequence and its constellation (constellation) size. In block 906, the decoder 308 attempts to decode the user packet using the previously demodulated symbols and demodulated FERAM samples stored in the BERAM 316.

In block 910, the decoder 308 or another unit may determine whether the user's packet was successfully decoded, i.e., passed the error check, for example, using a Cyclic Redundancy Code (CRC).

If the user packet is not successfully decoded, a NAK is sent back to the access terminal 106 in block 918. If the user packet is decoded correctly, an ACK is sent back to the access terminal 106 in block 908 and Interference Cancellation (IC) is performed in blocks 912 and 914. Block 912 regenerates a user signal based on the decoded signal, the channel impulse response, and the transmit/receive filter. Block 914 subtracts the user's influence from the FERAM312, thereby reducing its interference to users that have not yet been decoded.

Once the decoding fails and succeeds, the receiver continues to decode the next user in block 916. After decoding has been attempted for all users, a new slot is inserted into the FERAM312 and the entire process is repeated on the next slot. The samples may be written into the FERAM312 in real time, i.e., 2x code rate samples may be written every 1/2 chips.

FIG. 9B illustrates an apparatus including modules 930 and 946 for performing the method of FIG. 9A. The modules 930-946 in fig. 9B may be implemented in hardware, software, or a combination of hardware and software.

Method for selecting a decoding order

Block 903 indicates that TIC may be applied to users in order or in parallel. As these groups become larger, implementation complexity may decrease, but the benefits of TIC may also decrease unless TIC is iterated as described below.

The grouping and/or ordering criteria for users may vary depending on the rate at which the channel changes, the type of traffic, and the processing power available. A good decoding order may include users that decoded initially, where it is most beneficial to remove them and where they are most likely to decode. Criteria for achieving maximum gain according to TIC include:

A. payload size and T2P: the BTS104 may group or sort the users according to payload size and decode in order from the user with the highest transmit power, i.e., the highest T2P, to the user with the lowest T2P. Decoding and removing the high T2P users from the FERAM312 is of greatest benefit because they cause the most interference to other users.

B, SINR: the BTS104 may decode users with higher SINR before users with lower SINR because users with higher SINR have higher probability of decoding. Also, users with similar SINR may be grouped together. In the case of a fading channel, the SINR throughout the packet is time-varying, and therefore, an equivalent SINR can be calculated to determine the proper order.

C. Time: the BTS104 may decode "older" packets before "newer" packets (i.e., for which more subpackets have been received at the BTS 104). This choice reflects the assumption that for a given T2P ratio and ARQ termination goal, a packet is more likely to be decoded using each incremental subpacket.

Method for re-attempting decoding

As long as the user is correctly decoded, its interference effect is subtracted from the FERAM312, thus improving the ability to correctly decode all users that will share some slots. It is advantageous to repeat attempts to decode users that were previously decoded without success, since the interference they see may have been significantly reduced. The choose delay block 902 selects a time slot (current or past) as a reference for decoding and IC. The select user block 903 will select the user that terminates the subpacket in the slot of the selected delay. The selection of the delay may be based on the following selections:

A. the current decoding indicates that the selection moves to the next (future) slot when all users have attempted decoding, and the next slot is available in the FERAM 312. In this case, each user is attempted to be decoded once per slot, and this will correspond to successive interference cancellation.

B. Iterative decoding attempts to decode the user more than once for each processing of a slot. The second and subsequent decoding iterations will benefit from the interference of the decoded users being cancelled due to the previous iteration. Iterative decoding yields gains when multiple users are decoded in parallel without intervening IC. In the case of a full iterative decoding of the current slot, the choose delay block 902 will only choose the same slot (i.e., delay) multiple times.

C. Reverse decoding: the receiver demodulates the sub-packets and attempts to decode a packet based on demodulating all available sub-packets corresponding to the packet in the FERAM. After attempting to decode a packet using a subpacket that terminates in the current slot (i.e., a user on the current frame offset), the receiver may attempt to decode a packet that failed to decode in the previous slot (i.e., a user on the previous frame offset). Due to the partial overlap between asynchronous users, the removed interference of the subpacket terminating in the current slot will improve the chance of decoding the previous subpacket. This process can be iterated by tracing back more slots. The maximum delay in the forward link ACK/NAK may limit reverse decoding.

D. Forward decoding: after having attempted to decode all packets using the subpackets that terminate in the current slot, the receiver may also attempt to decode all subpackets of the nearest user before they are written to FERAM. For example, the receiver may attempt to decode the user after 3 of the 4 slots of the most recent subpacket have been received.

Method for updating BERAM

In a non-TIC BTS receiver, the packet is decoded only from the demodulated symbols stored in BERAM, and FERAM is only used to demodulate users from the nearest segment. With TIC, the FERAM312 is still accessed when the receiver attempts to demodulate a new user. However, in the case of TIC, after the user is reconstructed and subtracted to correctly decode the user, the FERAM312 is updated. For complexity reasons, the FERAM buffer length needs to be chosen to be less than the span of a packet (i.e., in EV-DO RevA, 40 slots need to span 16 slot packets). When new slots are written to the FERAM312, they will be overwritten on the oldest samples in the circular buffer. Thus, when a new slot is received, the oldest slot is overwritten and the BERAM316 is used by the decoder 308 for these older slots. It should be noted that even if a given sub-packet is located in the FERAM312, the BERAM316 may be used to store the earliest demodulated symbol (determined by the FERAM312) of the demodulator to which the sub-packet corresponds, as an intermediate step in the interleaving and decoding process. There are two main options for updating BERAM 316:

A. user-based update: the user's BERAM316 is updated only while the binding attempts to decode the user. In this case, the update of the earlier FERAM slot penalizes the user's BERAM316 if the given user is not decoding at the appropriate time (i.e., the updated FERAM slot may move out of the FERAM312 before attempting to decode the user).

B. Time slot based updating: to fully exploit the benefits of TIC, the BERAM316 of all affected users may be updated when the slots exit the FERAM 312. In this case, the contents of BERAM include all interference cancellation done on FERAM 312.

Method for canceling interference of sub-packets arriving due to missed ACK deadlines

In general, the additional processing used by TIC introduces a delay in the decoding process, especially when an iterative or inverse scheme is used. Such a delay may exceed the maximum delay for sending an ACK to the transmitter, thereby preventing transmission of subpackets associated with the same packet. In this case, the receiver can still utilize successful decoding by using the decoded data, thereby subtracting not only the previous subpackets, but also subpackets that will be received soon due to the missed ACK.

With TIC, the data of the decoded user is reconstructed and subtracted, so the base station 104 can remove the interference it causes to the packets of other users. For H-ARQ, an attempt is made to decode the original packet whenever a new subpacket is received. If the decoding is successful, for H-ARQ with TIC, the effect of the packet can be cancelled from the received samples by reconstructing and subtracting part of the sub-packet. Depending on complexity considerations, interference from 1, 2, 3, or 4 subpackets may be cancelled by storing a longer history of samples. In general, the IC may be applied in order to each user or group of users.

Fig. 10 shows the receiver sample buffer 312 at three time instances (slot n, n +12 slots, and n +24 slots). For ease of description, fig. 10 shows a single interlace with subpackets from three users at the same frame offset to highlight the interference cancellation operation in the case of H-ARQ. The receiver sample buffer 312 in fig. 10 spans all 4 subpackets (which for EV-DO RevA is implemented by a 40 slot buffer, since there are 8 slots between each 4 slot subpacket). The undecoded sub-packets are shown as shaded. The decoded subpackets are represented as unshaded portions in the 40 slot buffer and eliminated. Each time instance corresponds to an arrival time of another subpacket on the interlace. In slot n, when the most recent subpackets from users 2 and 3 fail to decode, the four stored subpackets for user 1 are decoded correctly.

In time instance n +12 slots, successive subpackets of the interlace arrive with interference cancellation of decoded (unshaded) subpackets 2, 3, and 4 of user 1. During time instance n +12 slots, the packets for users 2 and 3 are successfully decoded. Fig. 10 applies IC to groups of users on the same frame offset, but does not perform successive interference cancellation within the groups. In a typical group IC, users in the same group do not see common interference cancellation. Thus, as the number of users in a group increases, the execution complexity decreases, but there is a loss due to the lack of cancellation between users of the same group that make the same decoding attempt. However, with H-ARQ, the receiver will try all users in a block before each new subpacket arrives to achieve common interference cancellation for users in the same group. For example, when user 1's packet is decoded at time n, this facilitates user 2's and 3's packets being decoded at time n +12, which also facilitates user 1's decoding at time n + 24. All subpackets of previously decoded packets may be eliminated before re-attempting to decode other users' subsequent subpackets when they arrive. The key is that while a particular user may always be in the same group, their sub-packets observe IC gain when other group members decode.

Joint interference cancellation for pilot, overhead and traffic channels

This section addresses the problems associated with increasing the system capacity of a CDMA RL by efficiently estimating and canceling multi-user interference at the base station receiver. Typically, the RL user's signal includes pilot, overhead and traffic channels. This section describes a joint pilot, overhead and traffic IC scheme for all users.

Two aspects are described. First, overhead ic (oic) is introduced. On the other hand, the overhead of each user is interference to the signals of all other users. For each user, the total interference due to the overhead of all other users may be a large percentage of the total interference experienced by that user. Removing this overhead interference can further improve system performance (e.g., for CDMA20001xEV-DO RevA systems) and increase reverse link capacity beyond that achieved by PIC and TIC.

Second, important interactions between PIC, OIC, and TIC are shown by system performance and Hardware (HW) design tradeoffs. There are fewer schemes describing how to optimally combine all three cancellation procedures. Some may have higher performance gains and some may have greater complexity advantages. For example, one of the described schemes removes all pilot signals prior to decoding any overhead and traffic channels, and then decodes and removes the user's overhead and traffic channels in an orderly fashion.

This section is based on the CDMA20001x EV-DO RevA system and is equally applicable to other CDMA systems, such as W-CDMA, CDMA20001x and CDMA20001 xEV-DV.

Overhead channel elimination method

FIG. 11 shows an RL overhead channel structure such as EV-DO RevA. There are two types of overhead channels: one class, which facilitates RL demodulation/decoding, includes RRI (reverse rate indicator) channels and auxiliary pilot channels (used when payload size is 3072 or higher); another class facilitates Forward Link (FL) operation, which includes DRC (data rate control) channels, DSC (data source control) channels, and ACK (acknowledgement) channels. As shown in fig. 11, ACK and DSC are time multiplexed on a slot basis. When acknowledging a packet sent on the FL to the same user, only the ACK channel is sent.

Between overhead channels, the data of the secondary pilot channel is known a priori at the receiver. Thus, similar to the primary pilot channel, no demodulation and decoding is required for this channel, and the secondary pilot channel can be reconstructed from the channel-related knowledge. The auxiliary pilot signal can be reconstructed at a resolution of 2x chip rate and can be expressed as (on one chip):

<math><mrow> <msub> <mi>p</mi> <mi>f</mi> </msub> <mrow> <mo>[</mo> <mn>2</mn> <mi>n</mi> <mo>+</mo> <msub> <mi>&delta;</mi> <mi>f</mi> </msub> <mo>]</mo> </mrow> <mo>=</mo> <munderover> <mi>&Sigma;</mi> <mrow> <mi>&mu;</mi> <mo>=</mo> <mo>-</mo> <mi>M</mi> </mrow> <mi>M</mi> </munderover> <msub> <mi>c</mi> <mi>f</mi> </msub> <mrow> <mo>[</mo> <mi>n</mi> <mo>-</mo> <mi>&mu;</mi> <mo>]</mo> </mrow> <msub> <mi>w</mi> <mrow> <mi>f</mi> <mo>,</mo> <mi>aux</mi> </mrow> </msub> <mrow> <mo>[</mo> <mi>n</mi> <mo>-</mo> <mi>&mu;</mi> <mo>]</mo> </mrow> <mo>&CenterDot;</mo> <msub> <mi>G</mi> <mi>aux</mi> </msub> <mo>&CenterDot;</mo> <mrow> <mo>(</mo> <msub> <mi>h</mi> <mi>f</mi> </msub> <mi>&phi;</mi> <mrow> <mo>[</mo> <mn>8</mn> <mi>&mu;</mi> <mo>-</mo> <msub> <mi>&alpha;</mi> <mi>f</mi> </msub> <mo>]</mo> </mrow> <mo>)</mo> </mrow> <mo>,</mo> <mi>n</mi> <mo>=</mo> <mn>0</mn> <mo>,</mo> <mo>.</mo> <mo>.</mo> <mo>.</mo> <mo>,</mo> <mn>511</mn> </mrow></math>

<math><mrow> <msub> <mi>p</mi> <mi>f</mi> </msub> <mrow> <mo>[</mo> <mn>2</mn> <mi>n</mi> <mo>+</mo> <msub> <mi>&delta;</mi> <mi>f</mi> </msub> <mo>+</mo> <mn>1</mn> <mo>]</mo> </mrow> <mo>=</mo> <munderover> <mi>&Sigma;</mi> <mrow> <mi>&mu;</mi> <mo>=</mo> <mo>-</mo> <mi>M</mi> </mrow> <mi>M</mi> </munderover> <msub> <mi>c</mi> <mi>f</mi> </msub> <mrow> <mo>[</mo> <mi>n</mi> <mo>-</mo> <mi>&mu;</mi> <mo>]</mo> </mrow> <msub> <mi>w</mi> <mrow> <mi>f</mi> <mo>,</mo> <mi>aux</mi> </mrow> </msub> <mrow> <mo>[</mo> <mi>n</mi> <mo>-</mo> <mi>&mu;</mi> <mo>]</mo> </mrow> <mo>&CenterDot;</mo> <msub> <mi>G</mi> <mi>aux</mi> </msub> <mo>&CenterDot;</mo> <mrow> <mo>(</mo> <msub> <mi>h</mi> <mi>f</mi> </msub> <mi>&phi;</mi> <mrow> <mo>[</mo> <mn>8</mn> <mi>&mu;</mi> <mo>+</mo> <mn>4</mn> <mo>-</mo> <msub> <mi>&alpha;</mi> <mi>f</mi> </msub> <mo>]</mo> </mrow> <mo>)</mo> </mrow> <mo>,</mo> <mi>n</mi> <mo>=</mo> <mn>0</mn> <mo>,</mo> <mo>.</mo> <mo>.</mo> <mo>.</mo> <mo>,</mo> <mn>511</mn> </mrow></math>

auxiliary pilot signal reconstructed by equation 1

Where n corresponds to chipx1 sample rate, f is the number of branches, c_fIs a PN sequence, w_f，auxIs the Walsh code assigned to the auxiliary pilot channel, G_auxIs the relative gain, h, of the channel to the primary pilot_fIs the estimated channel coefficient (or channel response) assumed to be constant over a segment, phi is the filter function or convolution of the transmit pulse at chipx8 resolution and the receiver low pass filter (assuming phi is at-MT_c，MT_c]Non-negligible of middle), γ_fIs provided with alpha_f＝γ_fmod4 and δ_f＝[γ_f/4]Chip x8 of this branch is time-shifted.

The second set of overhead channels, including DRC, DSC, and RRI channels, are encoded by bi-orthogonal codes or simplex codes. At the receiver, the demodulated output is first compared to a threshold for each channel. If the output is below the threshold, an erasure is declared and no reconstruction is attempted for the signal. Otherwise, it is decoded by a symbol-based Maximum Likelihood (ML) detector, which may be within decoder 308 of fig. 4. The decoded output bits are used to reconstruct the corresponding channel as shown in fig. 4. The reconstructed signals for these channels can be expressed as:

<math><mrow> <msub> <mi>o</mi> <mi>f</mi> </msub> <mrow> <mo>[</mo> <mn>2</mn> <mi>n</mi> <mo>+</mo> <msub> <mi>&delta;</mi> <mi>f</mi> </msub> <mo>]</mo> </mrow> <mo>=</mo> <munderover> <mi>&Sigma;</mi> <mrow> <mi>&mu;</mi> <mo>=</mo> <mo>-</mo> <mi>M</mi> </mrow> <mi>M</mi> </munderover> <msub> <mi>c</mi> <mi>f</mi> </msub> <mrow> <mo>[</mo> <mi>n</mi> <mo>-</mo> <mi>&mu;</mi> <mo>]</mo> </mrow> <msub> <mi>w</mi> <mrow> <mi>f</mi> <mo>,</mo> <mi>o</mi> </mrow> </msub> <mrow> <mo>[</mo> <mi>n</mi> <mo>-</mo> <mi>&mu;</mi> <mo>]</mo> </mrow> <mo>&CenterDot;</mo> <msub> <mrow> <msub> <mi>d</mi> <mi>o</mi> </msub> <mi>G</mi> </mrow> <mi>o</mi> </msub> <mo>&CenterDot;</mo> <mrow> <mo>(</mo> <msub> <mi>h</mi> <mi>f</mi> </msub> <mi>&phi;</mi> <mrow> <mo>[</mo> <mn>8</mn> <mi>&mu;</mi> <mo>-</mo> <msub> <mi>&alpha;</mi> <mi>f</mi> </msub> <mo>]</mo> </mrow> <mo>)</mo> </mrow> <mo>,</mo> <mi>n</mi> <mo>=</mo> <mn>0</mn> <mo>,</mo> <mo>.</mo> <mo>.</mo> <mo>.</mo> <mo>,</mo> <mn>511</mn> </mrow></math>

<math><mrow> <msub> <mi>o</mi> <mi>f</mi> </msub> <mrow> <mo>[</mo> <mn>2</mn> <mi>n</mi> <mo>+</mo> <msub> <mi>&delta;</mi> <mi>f</mi> </msub> <mo>+</mo> <mn>1</mn> <mo>]</mo> </mrow> <mo>=</mo> <munderover> <mi>&Sigma;</mi> <mrow> <mi>&mu;</mi> <mo>=</mo> <mo>-</mo> <mi>M</mi> </mrow> <mi>M</mi> </munderover> <msub> <mi>c</mi> <mi>f</mi> </msub> <mrow> <mo>[</mo> <mi>n</mi> <mo>-</mo> <mi>&mu;</mi> <mo>]</mo> </mrow> <msub> <mi>w</mi> <mrow> <mi>f</mi> <mo>,</mo> <mi>o</mi> </mrow> </msub> <mrow> <mo>[</mo> <mi>n</mi> <mo>-</mo> <mi>&mu;</mi> <mo>]</mo> </mrow> <mo>&CenterDot;</mo> <msub> <mrow> <msub> <mi>d</mi> <mi>o</mi> </msub> <mi>G</mi> </mrow> <mi>o</mi> </msub> <mo>&CenterDot;</mo> <mrow> <mo>(</mo> <msub> <mi>h</mi> <mi>f</mi> </msub> <mi>&phi;</mi> <mrow> <mo>[</mo> <mn>8</mn> <mi>&mu;</mi> <mo>+</mo> <mn>4</mn> <mo>-</mo> <msub> <mi>&alpha;</mi> <mi>f</mi> </msub> <mo>]</mo> </mrow> <mo>)</mo> </mrow> <mo>,</mo> <mi>n</mi> <mo>=</mo> <mn>0</mn> <mo>,</mo> <mo>.</mo> <mo>.</mo> <mo>.</mo> <mo>,</mo> <mn>511</mn> </mrow></math>

equation 2 reconstructed overhead (DRC, DSC and RRI) signals

Compared with formula 1, there is a new term d_oIt is overhead channel data, w_f，oIs Walsh cover, and, G_auxRepresenting the overhead channel gain relative to the primary pilot.

The remaining overhead channels are 1-bit ACK channels. It may be BPSK modulated, uncoded, and repeated over half of the slot. The receiver may demodulate the signal and make a hard decision on the ACK channel data. The reconstructed signal model may be the same as equation 2.

Another method for reconstructing the ACK channel signal assumes that the demodulated and accumulated ACK signal after normalization can be expressed as:

y＝x+z，

where x is the transmit signal and z is the signal having a variance σ²The scaled noise term of (a). Then, the log-likelihood ratio (LLR) of y is expressed as:

<math><mrow> <mi>L</mi> <mo>=</mo> <mi>ln</mi> <mfrac> <mrow> <mi>Pr</mi> <mrow> <mo>(</mo> <mi>x</mi> <mo>=</mo> <mn>1</mn> <mo>|</mo> <mi>y</mi> <mo>)</mo> </mrow> </mrow> <mrow> <mi>Pr</mi> <mrow> <mo>(</mo> <mi>x</mi> <mo>=</mo> <mo>-</mo> <mn>1</mn> <mo>|</mo> <mi>y</mi> <mo>)</mo> </mrow> </mrow> </mfrac> <mo>=</mo> <mfrac> <mn>2</mn> <msup> <mi>&sigma;</mi> <mn>2</mn> </msup> </mfrac> <mi>y</mi> <mo>.</mo> </mrow></math>

then, for reconstruction, the soft estimates of the transmitted bits may be:

<math><mrow> <mover> <mi>x</mi> <mo>^</mo> </mover> <mo>=</mo> <mi>Pr</mi> <mrow> <mo>(</mo> <mi>x</mi> <mo>=</mo> <mn>1</mn> <mo>)</mo> </mrow> <mo>&CenterDot;</mo> <mn>1</mn> <mo>+</mo> <mi>Pr</mi> <mrow> <mo>(</mo> <mi>x</mi> <mo>=</mo> <mo>-</mo> <mn>1</mn> <mo>)</mo> </mrow> <mo>&CenterDot;</mo> <mrow> <mo>(</mo> <mo>-</mo> <mn>1</mn> <mo>)</mo> </mrow> <mo>=</mo> <mfrac> <mrow> <mi>exp</mi> <mrow> <mo>(</mo> <mi>L</mi> <mo>)</mo> </mrow> <mo>-</mo> <mn>1</mn> </mrow> <mrow> <mi>exp</mi> <mrow> <mo>(</mo> <mi>L</mi> <mo>)</mo> </mrow> <mo>+</mo> <mn>1</mn> </mrow> </mfrac> <mo>=</mo> <mi>tanh</mi> <mrow> <mo>(</mo> <mi>L</mi> <mo>)</mo> </mrow> <mo>=</mo> <mi>tanh</mi> <mrow> <mo>(</mo> <mfrac> <mn>2</mn> <msup> <mi>&sigma;</mi> <mn>2</mn> </msup> </mfrac> <mi>y</mi> <mo>)</mo> </mrow> <mo>,</mo> </mrow></math>

where the tanh function can be tabulated. In addition to using

Substitution d₀Otherwise, the reconstructed ACK signal is very similar to equation 2. In general, soft estimation and cancellation methods should give better cancellation performance because the receiver does not necessarily know the data, and such methods represent confidence as a picture. This approach can generally be extended to the overhead channels described above. However, the complexity of a Maximum A Posteriori (MAP) detector for obtaining LLRs for each bit increases exponentially with the number of information bits in one code symbol.

An efficient method for performing overhead channel reconstruction is that a branch can scale each decoded overhead signal with its relative gain, cover it with a Walsh code, add them together, and then burst spread with a PN sequence and filter through a scaled filter h phi. This approach may save computational complexity and memory bandwidth for elimination purposes.

<math><mrow> <munder> <mi>&Sigma;</mi> <mi>f</mi> </munder> <mi></mi> <msub> <mi>c</mi> <mi>f</mi> </msub> <mo>&CenterDot;</mo> <msub> <mi>d</mi> <mi>f</mi> </msub> <mo>&CenterDot;</mo> <msub> <mi>h</mi> <mi>f</mi> </msub> <mi>&phi;</mi> </mrow></math> Become into <math><mrow> <mrow> <mo>(</mo> <munder> <mi>&Sigma;</mi> <mi>f</mi> </munder> <mi></mi> <msub> <mi>c</mi> <mi>f</mi> </msub> <msub> <mi>d</mi> <mi>f</mi> </msub> <mo>&CenterDot;</mo> <msub> <mi>h</mi> <mi>f</mi> </msub> <mo>)</mo> </mrow> <mi>&phi;</mi> </mrow></math>

Combined PIC, OIC and TIC

Joint PIC, OIC, and TIC may be performed, thereby achieving high performance and increasing system capacity. Different decoding and cancellation orders for PIC, OIC, and TIC may result in different system performance and different impact on hardware design complexity.

First PIC and then OIC and TIC together (first scheme)

Fig. 12A shows a method of first performing PIC and then performing OIC and TIC together. After start block 1200, the receiver derives channel estimates for all users and performs power control in block 1202. Since the pilot data for all users is known at the BTS, they can be cancelled when their channels are estimated in PIC block 1204. Thus, all users' traffic channels and certain overhead channels observe less interference and can benefit from prior pilot cancellation.

Block 1206 selects a group of G un-decoded users, e.g., whose packets or subpackets terminate at the current slot boundary. Block 1208 and 1210 perform overhead/traffic channel demodulation and decoding. In block 1212, only successfully decoded channel data is reconstructed and subtracted from the user shared front end ram (feram). Block 1214 checks whether there are more users to decode. Block 1216 terminates the process.

Decoding/reconstruction/cancellation may be in order from one user in a group to the next user in the group, which may be referred to as successive interference cancellation. In this way, users of the same group of recent decoding orders benefit from the elimination of users of previous decoding orders. A simplified approach is to first decode all users in the same group and then quickly cancel their interference effects. The second method or scheme (described below) enables a smaller memory bandwidth and a more efficient pipeline architecture. In both cases, the group of packets that do not terminate at the same slot boundary but overlap with the user's packet benefit from this cancellation. This cancellation can account for most of the cancellation gain in asynchronous CDMA systems.

Fig. 12B illustrates an apparatus including a module 1230 and 1244 for performing the method of fig. 12A. The module 1230-1244 in fig. 12B may be implemented by hardware, software or a combination of hardware and software.

Fig. 13A shows a variation of the method of fig. 12A. Block 1204-1210 removes the signal based on the initial channel estimate in block 1202. Block 1300 derives a data-based channel estimate or an improved channel estimate. Data-based channel estimation may provide better channel estimation, as described below. Block 1302 performs the remaining PIC, i.e., removes the modified signal estimate from the improvement in channel estimation in block 1300.

For example, consider block 1204- > 1210 which results in removing the initial signal estimate (e.g., pilot signal) P1[ n ] from the received samples. The method then forms a modified channel estimate p2[ n ] based on the better channel estimate derived in block 1300. The method then removes the incremental difference between P2[ n ] -P1[ n ] from the sample locations in RAM 312.

Fig. 13B illustrates an apparatus including modules 1230 and 1244, 1310, 1312 for performing the method of fig. 13A. The modules 1230, 1310, 1312 in fig. 13B may be implemented in hardware, software, or a combination of hardware and software.

First PIC, then OIC, then TIC (second scheme)

This second scheme is the same as fig. 12A described above, except that the overhead channels of the same group of users are demodulated and decoded before any traffic channels are demodulated and decoded. This scheme is suitable for non-interleaved systems because no strict ACK deadline is imposed. For interleaved systems, such as DO rev.a, because the ACK/NAK responds to the traffic channel subpacket, the allowable decoding delay for the traffic channel subpacket is limited to a pair of slots (1 slot-1.67 ms). Thus, if a particular overhead channel is spread above this time range, the scheme may become unfeasible. Specifically, on DO RevA, the secondary pilot channel and ACK channel are in a short duration format and may be eliminated before TIC.

Combined pilot/overhead/traffic channel cancellation (third scheme)

Fig. 14A illustrates a method for performing joint PIC, OIC, and TIC. After start block 1400, the receiver derives channel estimates for all users and performs power control in block 1402. Block 1404 selects a group of G un-decoded users. Block 1406 re-estimates the channel from the pilot signal. Block 1408-. Block 1412 performs PIC for all users and OIC and TIC for only users with successfully decoded channel data.

Unlike the first scheme (fig. 12A) described above, the pilot signal is not canceled from the FERAM312 immediately after channel estimation is performed for all users (block 1402), and the channel estimation is used for power control as the non-IC scheme. The method then performs sequential decoding in a given order for a group of users that terminate at the same packet/subpacket boundary (blocks 1408 and 1410).

For a user attempting to decode, the method first re-estimates the channel based on the pilot signal (block 1402). When power control is demodulated due to interference cancellation of previously decoded packets overlapping the traffic packet to be decoded, the pilot signal observes less interference than time (block 1402). Thus, the channel estimation quality is improved, which benefits the traffic channel decoding and cancellation performance. The new channel estimates are used for traffic channel decoding (block 1410) and for specific overhead channel decoding (block 1408) (e.g., RRI channel in EV-DO). Once the decoding process is completed for a user in block 1412, the method will remove the interference impact of the user, including its pilot channel and any decoded overhead/traffic channels, from the FERAM 312.

Block 1414 checks whether there are more users to decode. Block 1416 terminates the process.

Fig. 14B illustrates an apparatus including a module 1420-1436 for performing the method of fig. 14A. Module 1420-1436 of fig. 14B may be implemented in hardware, software, or a combination of hardware and software.

Fig. 15A shows a variation of the method of fig. 14A. Block 1500 derives a data-based channel estimate. Block 1502 performs the optional remaining PIC as shown in fig. 13A.

Fig. 15B illustrates an apparatus including modules 1420-1436, 1510, 1512 for performing the method of fig. 15A. Modules 1420, 1510, 1512 of fig. 15B may be implemented in hardware, software, or a combination of hardware and software.

Compromise between the first and third schemes

It is clear that the first scheme should have superior performance compared to the third scheme, since the pilot signals are known at the BTS and it makes sense to cancel them out in advance. If it is assumed that the two schemes have the same cancellation quality, the first scheme may be better than the third scheme at all data rates. However, for the first scheme, the estimated channel coefficients for reconstruction purposes (for pilot and overhead/traffic) may be more noisy, since the pilot channel estimation observes higher interference than the traffic data demodulation. However, for the third scheme, the interference level observed by the improved channel estimation is the same as traffic data demodulation, since the pilot channel estimation is re-performed just prior to traffic data demodulation/decoding. Thus, on average, the cancellation quality of the third scheme is better than the first scheme.

From a hardware design point of view, the third scheme is somewhat stronger: the method can sum the pilot frequency and the decoded overhead and the traffic channel data and eliminate the pilot frequency and the decoded overhead and the traffic channel data together, so that the method saves the bandwidth of a memory. On the other hand, the re-estimation of the pilot signal can be performed together with overhead channel demodulation or traffic channel demodulation (in terms of reading samples from memory), and therefore the memory bandwidth requirements are not increased.

If it is assumed that the cancellation quality of the first scheme is 80% or 90% of the cancellation quality of the third scheme, a tradeoff is required between the data rate of each user and the increase in the number of users. Generally, the first scheme is preferred if all users are in the low data rate region, and vice versa if all are high data rate users. The method may also re-estimate the channel from the traffic channel when decoding a data packet. The cancellation quality will be improved because the traffic channel operates at a (very) higher SNR than the pilot channel.

Overhead channels can be removed (canceled) once they are successfully demodulated and traffic channels can be removed once they have been successfully demodulated and decoded. The base station can successfully demodulate/decode the overhead and traffic channels of all access terminals at any point in time. If this (PIC, OIC, TIC) occurs, FERAM will only include the remaining interference and noise. The pilot, overhead, and traffic channels may be canceled in various orders and canceled for a subset of the access terminals.

One approach is to perform interference cancellation (any combination of PIC, TIC and OIC) on one user at a time from RAM 312. The other method is as follows: (a) adding up the reconstructed signals (any combination of PIC, TIC and OIC) of a group of users; (b) interference cancellation is then performed simultaneously on the groups. These two methods are applicable to any of the methods, protocols, and processes disclosed herein.

Improving channel estimation for interference cancellation

The ability to correctly reconstruct the received samples can significantly impact the system performance of a CDMA receiver that achieves interference cancellation by reconstructing and removing portions of the transmitted data. In a RAKE receiver, the multipath channel is estimated by PN despreading against a pilot sequence and then pilot filtering (i.e., accumulation) over an appropriate time period. Typically, the length of the pilot filtering is chosen based on a trade-off between increasing the estimated SNR by accumulating more samples, while not accumulating too long to degrade the estimated SNR due to time variations of the channel. The channel estimate output by the pilot filter is then used to perform data demodulation.

As illustrated in fig. 4, one possible approach to performing interference cancellation in a CDMA receiver is to reconstruct the effect of various transmit chipx1 streams on (e.g., chipx2) FERAM samples. This involves determining the transmitted chip stream and an estimate of the total channel between the transmitter chips and the receiver samples. Since the channel estimates of the RAKE fingers represent the multipath channel itself, the overall channel estimate should also account for the presence of transmitter and receiver filtering.

This section discloses several techniques for improving the overall channel estimate for interference cancellation in a CDMA receiver. These techniques may be applicable to CDMA2000, 1xEV-DO, 1xEV-DV, WCDMA.

To perform TIC on the decoded correct packet, the receiver in fig. 4 can take the information bits from the decoder output and reconstruct the transmit chip stream by re-encoding, re-interleaving, re-modulating, reapplying the data channel gain, and re-spreading. To estimate the received samples of TIC using the pilot channel estimate, the transmit chip stream is convolved with the channel estimate of the RAKE receiver despread with the pilot PN sequence and a model of the transmitter and receiver filters.

If pilot channel estimation is not used, improved channel estimation (at each RAKE finger delay) can also be obtained by despreading with the reconstructed data chips themselves. This improved channel estimate is not useful for data demodulation of a packet because the packet has been successfully decoded, but is only used to reconstruct the effect of the packet on the front-end samples. With this technique, for each delay of the RAKE finger (e.g., chipX8 resolution), the method may "despread" the received samples (e.g., insert into chipX8) using the reconstructed stream of data chips and accumulate over the appropriate time period. This will improve channel estimation because the traffic channel is transmitted with higher power than the pilot channel (the traffic to pilot T2P ratio is a function of the data rate). Estimating the channel of TIC using the data chips may result in more accurate channel estimates for higher power users, who are critical to cancellation with high accuracy.

Instead of estimating the multipath channel in each RAKE finger delay, this section also describes a channel estimation procedure that will explicitly estimate the combined impact of the transmitter filter, multipath channel, and receiver filter. The estimation may employ the same resolution as the oversampled front-end samples (e.g., chipx2 FERAM). The reconstructed transmit data chips may be used to despread the samples at the front end to achieve a T2P gain in terms of channel estimation accuracy, thereby achieving channel estimation. The time span of the uniformly spaced channel estimates may be selected based on information about the RAKE finger delays and a priori estimates of the combined response of the transmitter and receiver filters. Furthermore, the information of the RAKE fingers can be used to improve the evenly spaced channel estimation.

Fig. 16 shows a model of a transmission system with a transmit filter p (t), a total/composite channel h (t) (compared to the multipath channel g (t) described below), and a receive filter q (t). A digital baseband representation of a wireless communication channel may be modeled with L discrete multipath components:

<math><mrow> <mi>g</mi> <mrow> <mo>(</mo> <mi>t</mi> <mo>)</mo> </mrow> <mo>=</mo> <munderover> <mi>&Sigma;</mi> <mrow> <mi>l</mi> <mo>=</mo> <mn>1</mn> </mrow> <mi>L</mi> </munderover> <msub> <mi>&alpha;</mi> <mi>l</mi> </msub> <mi>&delta;</mi> <mrow> <mo>(</mo> <mi>t</mi> <mo>-</mo> <msub> <mi>&tau;</mi> <mi>l</mi> </msub> <mo>)</mo> </mrow> </mrow></math> equation 3

Wherein the complex path amplitudes are with corresponding delays τ_lA of (a)_l. The combined effect of the transmitter and receiver filters can be defined as phi (t), where

<math><mrow> <mi>&phi;</mi> <mrow> <mo>(</mo> <mi>t</mi> <mo>)</mo> </mrow> <mo>=</mo> <mi>p</mi> <mrow> <mo>(</mo> <mi>t</mi> <mo>)</mo> </mrow> <mo>&CircleTimes;</mo> <mi>q</mi> <mrow> <mo>(</mo> <mi>t</mi> <mo>)</mo> </mrow> </mrow></math> Equation 4

Wherein,

representing a convolution. The combined phi (t) is typically chosen to resemble an elevated cosine response. For example, in CDMA2000 and its derivatives, the response is similar to the example phi (t) shown in fig. 17. The overall channel estimate is expressed as:

<math><mrow> <mover> <mi>h</mi> <mo>^</mo> </mover> <mrow> <mo>(</mo> <mi>t</mi> <mo>)</mo> </mrow> <mo>=</mo> <mi>g</mi> <mrow> <mo>(</mo> <mi>t</mi> <mo>)</mo> </mrow> <mo>&CircleTimes;</mo> <mi>&phi;</mi> <mrow> <mo>(</mo> <mi>t</mi> <mo>)</mo> </mrow> <mo>=</mo> <munderover> <mi>&Sigma;</mi> <mrow> <mi>l</mi> <mo>=</mo> <mn>1</mn> </mrow> <mi>L</mi> </munderover> <msub> <mi>&alpha;</mi> <mi>l</mi> </msub> <mi>&phi;</mi> <mrow> <mo>(</mo> <mi>t</mi> <mo>-</mo> <msub> <mi>&tau;</mi> <mi>l</mi> </msub> <mo>)</mo> </mrow> </mrow></math> equation 5

Fig. 18A and 18B show examples of channel estimates (real and imaginary parts) from an estimated multipath channel at each of three RAKE fingers. In this example, the actual channel is shown as a solid line and a is shown as a star_l. Reconstruction (dashed line) is based on using a in equation 3 above_l. The RAKE finger channel estimation in fig. 18A and 18B is based on despreading with pilot chips (where the overall pilot SNR is-24 dB).

Despreading using regenerated data chips instead of pilot chips at RAKE finger delays

The quality of the channel estimate directly affects the fidelity of the reconstructed user's impact on the received signal. To improve the performance of CDMA systems implementing interference cancellation, the reconstructed data chips of the user may be used to determine an improved channel estimate. This will improve the accuracy of the interference cancellation. A system for a CDMA system may be described as "despreading associated with a user's transmitted data chips" as opposed to typical "despreading associated with a user's transmitted pilot chips".

The RAKE finger channel estimation in fig. 18A-18B is based on despreading with pilot chips (where the overall pilot SNR is-24 dB). Fig. 19A-19B illustrate an example of improved channel estimation based on RAKE fingers and despreading with data chips transmitted using 10dB more power than pilot chips.

Fig. 20A illustrates a method of despreading at a RAKE finger using regenerated data chips. In block 2000, the Rake receiver 314 (fig. 4) despreads the samples at the front end with pilot PN chips to obtain Rake finger values. In block 2002, the demodulator 304 performs data demodulation. In block 2004, the decoder 308 performs data decoding and checks the CRC. In block 2006, if the CRC passes, unit 400 determines to transmit the data chips by re-encoding, re-interleaving, re-modulating, and re-spreading. In block 2008, unit 400 despreads the front-end samples with the transmit data chips to obtain improved channel estimates at each finger delay. In block 2010, unit 400 reconstructs the traffic and overhead impact of the user on the front-end samples using the improved channel estimates.

Fig. 20B shows an apparatus comprising a module 2020 and 2030 for performing the method of fig. 20A. Module 2020 + 2030 in fig. 20B may be implemented in hardware, software, or a combination of hardware and software.

Estimating composite channel at FERAM resolution using regenerated data chips

A typical CDMA receiver may estimate the complex values of the multipath channel at each RAKE finger delay. The receiver front-end before the RAKE receiver may include a low-pass receiver filter (i.e., q (t)) that is matched to the transmitter filter (i.e., p (t)). Thus, for a receiver that performs a filter that matches the channel output, the RAKE receiver itself attempts to match only the multipath channel (i.e., g (t)). The delays of the RAKE fingers are typically driven by separate time tracking loops within a minimum separation requirement (e.g., the fingers are separated by at least one chip). Thus, one approach estimates the composite channel (i.e., h (t)) at the resolution of the front-end samples (e.g., chipx2 FERAM).

Using transmit power control on the CDMA reverse link, the combined branch SNR from all multipath and receiver antennas is typically controlled to be within a particular range. This range of SNRs may enable a composite channel estimate to be derived from despread pilot chips with a relatively large estimation variance. This is why the RAKE receiver tries to place the fingers only at the "peak end" of the energy delay profile. However, with the advantage of T2P despreading with reconstructed data chips, the composite channel estimate can make the estimate of h (T) better than the direct estimate of g (T) combined with the model of phi (T).

The channel estimation process described herein explicitly estimates the combined effect of the transmitter filter, the multipath channel, and the receiver filter. Such estimation may employ the same resolution as the oversampling front-end samples (e.g., chipx2 FERAM). The reconstructed transmit data chips may be used to despread the samples at the front end to achieve a T2P gain in terms of channel estimation accuracy, thereby achieving channel estimation. The time span of the uniformly spaced channel estimates may be selected based on information about the RAKE finger delays and a priori estimates of the combined response of the transmitter and receiver filters. Furthermore, the information of the RAKE fingers can be used to improve the evenly spaced channel estimation. Note that the technique of estimating the composite channel itself is also useful because it does not require the design to use a priori estimates of phi (t).

Fig. 21A, 21B show an example of estimating a composite channel using evenly spaced samples at chipX2 resolution. In FIGS. 21A, 21B, the data chip SNR is-4 dB, corresponding to a pilot SNR of-24 dB and T2P of 20 dB. A uniform channel estimate gives better quality than despreading with data chips only in the RAKE finger region. At high SNR, the effect of the "fat path" limits the ability to correctly reconstruct the channel using the RAKE finger positions. The uniform sampling method is particularly useful when the estimated SNR is high, corresponding to the case of despreading with data chips of high T2P. When T2P is high for a particular user, channel reconstruction fidelity is critical.

Fig. 22A illustrates a method of estimating a composite channel with uniform resolution using regenerated data chips. Blocks 2000-2006 and 2010 are similar to FIG. 20A described above. In block 2200, RAKE receiver 314 (fig. 4) or another component determines a uniformly constructed time span based on the RAKE finger delays. In block 2202, the demodulator 304 or another component determines an improved channel estimate by despreading front-end samples with transmitted data chips at uniform delays for an appropriate time span.

Fig. 22B shows an apparatus comprising modules 2020, 2030, 2220, 2222 for performing the method of fig. 22A. Module 2020 + 2030 in fig. 22B may be implemented in hardware, software, or a combination of hardware and software.

In the above description, g (t) is the wireless multipath channel itself, while h (t) includes the wireless multipath channel and transmitter and receiver filtering: convolution is performed with phi (t) and h (t) g (t).

In the above description, "samples" may take any rate (e.g., two per chip), but "data chips" are one per chip.

The "regenerated data chips" are formed by re-encoding, re-interleaving, re-demodulating, and re-spreading, as shown in block 2006 of 20A, and as described above. In principle, "regeneration" is the process of mimicking the passage of information bits through a mobile transmitter (access terminal).

"reconstructed samples" means samples (e.g., two per chip) stored in the FERAM312 or in a memory different from the FERAM312 of the receiver. These reconstructed samples are formed by convolving the (regenerated) transmitted data chips with the channel estimate.

The terms "reconstruct" and "regenerate" may be used interchangeably if the context provided is used to reform the transmitted data chips or reform the received samples. The samples or chips may be reformed because the "chips" are reformed by re-encoding or the like, and the "samples" are reformed based on the action of the wireless channel (channel estimate) using the reformed chips in combination with the transmitter and receiver filtering. The terms "reconstitution" and "regeneration" refer essentially to reconstitution or reformation. There is no technical difference. One embodiment uses only "regeneration" for data chips and only "reconstruction" for samples. Therefore, the receiver may have a data chip regeneration unit and a sample reconstruction unit.

Adaptation of transmit subchannel gain on the reverse link of a CDMA system with interference cancellation

Multi-user interference is a limiting factor in CDMA transmission systems and any receiver technique that reduces this interference can lead to significant improvements in achievable throughput. This section describes techniques for varying the transmit subchannel gain for a system having an IC.

In reverse link transmission, each user transmits pilot, overhead, and traffic signals. The pilot signal enables synchronization and estimation of the transmission channel. Overhead subchannels (e.g., RRIs, DRCs, and ACKs) are required for MAC and traffic decoding devices. The pilot, overhead and traffic subchannels have different requirements on the signal to interference plus noise ratio (SINR). In a CDMA system, single power control may vary the transmit power of the pilot, while the power of the overhead and traffic subchannels have fixed gains relative to the pilot. When a BTS is equipped with PIC, OIC, and TIC, various subchannels observe different levels of interference depending on the order and cancellation capability of the IC. In this case, the static relationship between the subchannel gains can compromise system performance.

This section describes new gain control strategies for different logical sub-channels associated with the system implementing the IC. These techniques are based on CDMA systems such as EV-DO RevA and are applicable to EV-DV Rel D, W-CDMAEUL, and CDMA 2000.

The described techniques perform power and gain control on different sub-channels by adaptively changing the gain of each sub-channel according to measured performance in terms of packet error rate, SINR, or interference power. The goal is to improve reliable power and gain control mechanisms, thereby enabling full utilization of the IC potential, while providing robustness for transmission over time-varying dispersive subchannels.

Interference cancellation refers to removing the effect of logical subchannels on the front-end sampling after these subchannels have been decoded, thereby reducing interference to other signals that will be decoded later. In PIC, the transmit pilot signal is known at the BTS and the channel estimate is used to reconstruct the receive pilot signal. In TIC or OIC, the interference is removed by reconstructing the received sub-channel via a decoded version of the received sub-channel at the BTS.

The current BTS (without IC) controls the power E of the pilot subchannel_cpThereby meeting the error rate requirements in the traffic channel. The power of the traffic subchannel is related to the pilot signal by a fixed factor T2P, which depends on the payload type and the target termination purpose. The adaptation of the pilot power is performed by a closed loop power control mechanism comprising an inner loop and an outer loop. The inner loop is used to maintain the SINR (Ecp/Nt) of the pilot signal at a threshold level T, while the outer loop power control changes the threshold level T, for example, according to the Packet Error Rate (PER).

When IC is performed at the receiver (fig. 4), the adaptivity of the subchannel gains is beneficial to the system. In fact, because each subchannel observes a different level of interference, their gains should be changed accordingly with respect to the pilot signal to provide the desired performance. This section may address the problem of gain control for overhead and pilot subchannels, and describes techniques for the adaptivity of T2P to improve system throughput by leveraging IC.

Important parameters in systems with ICs

Two parameters that may be adjusted are the overhead subchannel gain and the traffic and pilot (T2P) gain. When TIC is active, the overhead subchannel gain may be increased (relative to no TIC), thereby achieving a more flexible tradeoff between pilot and overhead channels. By denoting by G the baseline G used in the current system, the new value of the overhead channel will be:

G′＝G·Δ_G。

in the non-TIC scheme, the overhead/pilot subchannel observes the same interference level, since the traffic channel and a specific ratio T2P/G can give satisfactory performance for overhead and traffic channel performance as well as pilot channel estimation. When IC is used, the interference level is different for overhead/pilot and traffic and T2P can be reduced, allowing both types of subchannels to achieve consistent performance. For a given payload, the method passes a factor Δ relative to the tabulated value_T2PBut T2P can be made to decrease to meet the requirements. By representing the baseline T2P for a particular payload in the current system with T2P, the new value of T2P will be:

T2P′＝T2P·Δ_T2P。

the parameter Δ can be set_T2PQuantized into a set of finite or discrete values (e.g., -0.1dB to-1.0 dB), and transmitted to the access terminal 106.

Some of the quantities that may be controlled are traffic PER, pilot SINR, and thermal delta. The pilot SINR should not drop below the minimum level required for good channel estimation. The rise-over-thermal (ROT) is critical to ensure the stability and link budget of the power-controlled CDMA reverse link. In a TIC-free receiver, ROT is defined from the received signal. Generally, the ROT should be kept within a predetermined range to achieve a better capacity/coverage tradeoff.

Thermal increment control

I₀Representing the power of the signal at the input of the receiver. Interference cancellation of the received signal results in a power reduction. I is₀' represents the average power of the signal at the input of demodulator 304 after IC:

I₀′≤I₀。

after having updated I with IC₀After the value of' it can be determined from the front-end sampling. When the IC is to be executed, the IC,ROT is still crucial for the overhead subchannels and should be controlled with respect to the threshold, i.e. to ensure that:

$\mathrm{ROT}=\frac{I_0}{{\mathrm{<figure-callout\; id="0"\; label="N"\; filenames="S05848202220070822E000061.png,S05848202220070822E000211.png"\; state="\{\{state\}\}">N</figure-callout>}}_0}<{\mathrm{ROT}}_{\mathrm{thr}},$

wherein N is₀Is the noise power.

However, traffic and partial overhead subchannels also benefit from IC. The decoding performance of these sub-channels is related to the measured rise-over-thermal after IC. The effective ROT is the ratio of the signal power and the noise power after the IC. The effective ROT may be controlled by a threshold, namely:

<math><mrow> <msub> <mi>ROT</mi> <mi>eff</mi> </msub> <mo>=</mo> <mfrac> <msup> <msub> <mi>I</mi> <mn>0</mn> </msub> <mo>&prime;</mo> </msup> <msub> <mi>N</mi> <mn>0</mn> </msub> </mfrac> <mo><</mo> <msubsup> <mi>ROT</mi> <mi>thr</mi> <mrow> <mo>(</mo> <mi>eff</mi> <mo>)</mo> </mrow> </msubsup> <mo>.</mo> </mrow></math>

to ROT_effCan be equivalently expressed as a pair I₀The constraint of' assumes that the noise level does not change:

<math><mrow> <msup> <msub> <mi>I</mi> <mn>0</mn> </msub> <mo>&prime;</mo> </msup> <mo>&le;</mo> <msubsup> <mi>I</mi> <mn>0</mn> <mrow> <mo>(</mo> <mi>thr</mi> <mo>)</mo> </mrow> </msubsup> <mo>,</mo> </mrow></math> wherein is and

a corresponding signal power threshold.

Fixed overhead gain techniques

As the ROT increases, the SINR of the pilot and overhead channels (which do not benefit from IC) decreases, resulting in a potential increase in erasure rate. To compensate for this effect, the overhead channel gain may be increased by a fixed value or by adapting to specific system conditions.

Techniques are described in which the gain of an overhead subchannel is fixed relative to a pilot. The presented technique changes the level of the pilot subchannel and the delta for each user_T2P。

With a fixed delta _G Closed loop control of T2P at 0dB

FIG. 23 shows the data for E_cpAnd Δ_T2PAnd a fixed delta_GClosed loop Power Control (PC) for 0dB (block 2308). Delta_T2PAnd E_cpThe first adaptation scheme of (1) comprises:

A. the inner and outer rings 2300, 2302 may be paired with each other in a conventional manner_cpPerforms power control adaptively. Outer loop 2300 receives a target PER and a traffic PER. Inner loop 2304 receives threshold T2302 and the measured pilot SINR and outputs E_cp。

B. Closed loop Gain Control (GC)2306 changes Δ based on a measure of removed interference_T2P. Gain control 2306 receives the measured ROT and the measured ROTeff, and outputs Δ_T2P. The receiver determines the interference removed by the IC scheme and adjusts delta_T2P。

C. Can periodically send a in the message_T2PTo all access terminals 106 within the sector.

For Δ_T2PIf interference after IC is from I₀Is reduced to I₀', then the amount of T2P may be subsequently reduced:

<math><mrow> <msub> <mi>&Delta;</mi> <mrow> <mi>T</mi> <mn>2</mn> <mi>P</mi> </mrow> </msub> <mo>=</mo> <mfrac> <msup> <msub> <mi>I</mi> <mn>0</mn> </msub> <mo>&prime;</mo> </msup> <msub> <mi>I</mi> <mn>0</mn> </msub> </mfrac> <mo>&ap;</mo> <mfrac> <msub> <mi>ROT</mi> <mi>eff</mi> </msub> <mi>ROT</mi> </mfrac> <mo>.</mo> </mrow></math>

E_cpwill increase (via PC ring 2304) to:

<math><mrow> <msup> <msub> <mi>E</mi> <mi>cp</mi> </msub> <mo>&prime;</mo> </msup> <mo>=</mo> <mfrac> <msup> <msub> <mi>I</mi> <mn>0</mn> </msub> <mo>&prime;</mo> </msup> <msubsup> <mi>I</mi> <mn>0</mn> <mrow> <mo>(</mo> <mi>thr</mi> <mo>)</mo> </mrow> </msubsup> </mfrac> <msub> <mi>E</mi> <mi>cp</mi> </msub> <mo>.</mo> </mrow></math>

the ratio of the total transmit power of the system with and without IC would be:

<math><mrow> <mi>C</mi> <mo>=</mo> <mfrac> <mrow> <msub> <mi>E</mi> <mi>cp</mi> </msub> <mrow> <mo>(</mo> <mn>1</mn> <mo>+</mo> <mi>G</mi> <mo>+</mo> <mi>T</mi> <mn>2</mn> <mi>P</mi> <mo>)</mo> </mrow> </mrow> <mrow> <msup> <msub> <mi>E</mi> <mi>cp</mi> </msub> <mo>&prime;</mo> </msup> <mrow> <mo>(</mo> <mn>1</mn> <mo>+</mo> <mi>G</mi> <mo>+</mo> <mi>T</mi> <mn>2</mn> <mi>P</mi> <mo>&prime;</mo> <mo>)</mo> </mrow> </mrow> </mfrac> <mo>,</mo> </mrow></math>

where G is the overhead channel gain. For larger values of T2P (relative to G), the ratio C may be approximated as:

<math><mrow> <mi>C</mi> <mo>&ap;</mo> <mfrac> <msubsup> <mi>I</mi> <mn>0</mn> <mrow> <mo>(</mo> <mi>thr</mi> <mo>)</mo> </mrow> </msubsup> <msup> <msub> <mi>I</mi> <mn>0</mn> </msub> <mo>&prime;</mo> </msup> </mfrac> <mo>.</mo> </mrow></math>

to estimate the effective ROT, the effective ROT may change rapidly due to changes in the PC and channel conditions. However,. DELTA._T2PReflecting the ROT_effThe change is slow. Thus, for Δ_T2PAlternatively, the effective ROT is measured by a long averaging window of the signal after the IC. The length of the averaging window may be at least twice the length of the power control update period.

And has a fixed delta _G >Closed loop control of T2P of 0dB

Except that gain control 2306 receives a threshold effective ROT and Δ_G>Fig. 24 is the same as fig. 23 except for 0dB (box 2400). For varying a_T2PThis alternative method of (a) is based on a request for the same cell coverage for both IC and non-IC systems. In both cases, E_cpThe distribution is the same. The IC's effect doubles for a fully loaded system: i) signal power before IC I₀Will increase relative to the signal power of the system without the IC; ii) due to closed loop power control by PER control, I₀' will tend to be the same signal power as a system without an IC. Change of Δ as follows_T2P：

<math><mrow> <msub> <mi>&Delta;</mi> <mrow> <mi>T</mi> <mn>2</mn> <mi>P</mi> </mrow> </msub> <mo>=</mo> <mfrac> <msubsup> <mi>I</mi> <mn>0</mn> <mrow> <mo>(</mo> <mi>thr</mi> <mo>)</mo> </mrow> </msubsup> <msup> <msub> <mi>I</mi> <mn>0</mn> </msub> <mo>&prime;</mo> </msup> </mfrac> <mo>&ap;</mo> <mfrac> <msubsup> <mi>ROT</mi> <mi>thr</mi> <mrow> <mo>(</mo> <mi>eff</mi> <mo>)</mo> </mrow> </msubsup> <msub> <mi>ROT</mi> <mi>eff</mi> </msub> </mfrac> <mo>.</mo> </mrow></math>

Controlling delta based on ACK _T2P

FIG. 25 shows E based on ACK subchannels with fixed overhead subchannel gain_cpAnd Δ_T2PPC (block 2506).

Δ_T2PRequires a feedback signal from the BTS to the ATs, all of which receive a delta from the BTS_T2PThe same broadcast value of (c). An alternative is based on delta_T2POpen-loop GC2510 and closed- loop PCs 2500, 2504 for pilot signals. The closed loop pilot PC includes an inner loop 2504 that is based on a threshold value T ₀2502 Regulation E_cp. The outer loop control 2500 is controlled by the erasure rate of the overhead channel (e.g., the Data Rate Control (DRC) subchannel error probability or DRC erasure rate). Increasing T when DRC erasure rate exceeds a threshold₀However, when the DRC erasure rate is lower than the threshold, it is gradually decreased.

Change of delta by ACK Forward subchannel_T2P. Specifically, by measuring the statistics of ACK and NACK, the AT may estimate the traffic PER AT the BTS (block 2508). The gain control 2510 includes a target traffic PER and a measured PER. When PER is above threshold, increase Δ_T2PUntil T2P' reaches the baseline value T2P for the no-IC system. On the other hand, for lower PERs, Δ is reduced to take full advantage of IC processing_T2P。

Variable overhead gain technique

By making a_T2PAnd the overhead subchannel gain (G overhead) is adapted to the IC processing to allow further optimization of the transceiver. In this case, an additional feedback signal is required. Can convert a_GThe value of (d) is quantized from 0dB to 0.5 dB.

Overhead gain control based on interference power

Fig. 26 is the same as fig. 24 except for the overhead GC 2600. The method of GC2600 for the overhead subchannel is based on the signal power measured after IC. In this case, assume E_cpThe same cell coverage is provided for systems without ICs. The signal before the IC has an increased power I₀And, overhead gain compensates for the increased interference. This embodiment changes the overhead gain by setting:

<math><mrow> <msub> <mi>&Delta;</mi> <mi>G</mi> </msub> <mo>=</mo> <mfrac> <msub> <mi>I</mi> <mn>0</mn> </msub> <msubsup> <mi>I</mi> <mn>0</mn> <mrow> <mo>(</mo> <mi>thr</mi> <mo>)</mo> </mrow> </msubsup> </mfrac> <mo>&ap;</mo> <mfrac> <mi>ROT</mi> <msub> <mi>ROT</mi> <mi>thr</mi> </msub> </mfrac> <mo>.</mo> </mrow></math>

can convert a_GControl is to be not less than 0dB as this will correspond to reducing the overhead subchannel power which is unlikely to be of benefit.

The gain and power control scheme may include for E as in fig. 23_cpInner and outer rings PC2304, 2300, GC ring 2600 for Δ G as described above, GC ring for Δ G_T2POpen loop GC2306 of (1), wherein, when PER is higher than a target value, Δ_T2PIncreases and, when PER is lower than a target value, Δ_T2PAnd decreases. Delta_T2PIs allowed, corresponding to the level of the non-IC receiver.

DRC-only overhead gain control

Fig. 27 shows a variation of fig. 26 using DRC-only overhead gain control 2702.

Performing delta using closed loop even when overhead subchannel gain is changed_T2P Gain control 2700 as described above. In this case, E is controlled according to the scheme of FIG. 23_cpAnd Δ_T2PWhile performing adaptation of the overhead subchannel gains through DRC erasure rate 2702. Specifically, if the DRC erasure rate is above a threshold, the overhead subchannel gain 2702 is increased. When the DRC erasure rate is below a threshold, the overhead gain 2702 is gradually decreased.

Control of T2P in a multi-sector multi-cell network

Since Δ is performed at the cell level_T2PSo that the AT106 can be in a soft handoff state and each sector can generate a differenceTo the adaptive request of (3). In this case, Δ for AT_T2PVarious options may be considered for the selection of the request. At the cell level, one approach may choose to reduce the smallest T2P between T2P requested by a fully loaded sector, namely:

<math><mrow> <msubsup> <mi>&Delta;</mi> <mrow> <mi>T</mi> <mn>2</mn> <mi>P</mi> </mrow> <mrow> <mo>(</mo> <mi>cell</mi> <mo>)</mo> </mrow> </msubsup> <mo>=</mo> <munder> <mi>max</mi> <mrow> <mi>s</mi> <mo>&Element;</mo> <mrow> <mo>{</mo> <mi>loaded sectors</mi> <mo>}</mo> </mrow> </mrow> </munder> <mrow> <mo>{</mo> <msubsup> <mi>&Delta;</mi> <mrow> <mi>T</mi> <mn>2</mn> <mi>P</mi> </mrow> <mrow> <mo>(</mo> <mi>s</mi> <mo>)</mo> </mrow> </msubsup> <mo>}</mo> </mrow> </mrow></math>

wherein,

is the required Δ for sector s_T2P. The AT may receive different requests from various sectors and, also in this case, various criteria may be employed. A method can select a delta corresponding to a serving sector_T2PThereby ensuring maximum reliable communication therewith.

For delta AT cell and AT AT_T2POther choices may be considered, including a minimum, maximum, or average of the requested values.

An important aspect is for the mobile station to use T2P ═ T2Px Δ_T2PWherein, is_T2PIs at the BTS according to Io and Io' (and possibly alsoAnd G '═ Gx Δ), and G' ═ Gx Δ_GWherein, is_GIs also estimated at the BTS. With these delta factors computed at the BTSs, each BTS transmits them to all access terminals that react accordingly.

The concepts described herein are applicable to WCDMA systems that use overhead channels such as a Dedicated Physical Control Channel (DPCCH), an enhanced dedicated physical control channel (E-DPCCH), or a high speed dedicated physical control channel (HS-DPCCH). WCDMA systems may use a Dedicated Physical Data Channel (DPDCH) format and/or an enhanced dedicated physical data channel (E-DPDCH) format.

The disclosed application is applicable to WCDMA systems with two different interleaving structures (e.g., 2 millisecond transmit time interval and 10 millisecond transmit time interval). Thus, the front-end memory, demodulator, and subtractor may span one or more subpackets of a packet having different transmit time intervals.

For TIC, one or more users may transmit traffic data in at least one of EV-DO version 0 format or EV-DO modified version A format.

The particular decoding order described herein may correspond to the order of demodulation and decoding. Re-decoding the packet should be based on re-demodulation because the process of demodulating the packet of FERAM312 translates interference cancellation into a better decoder output.

Those of skill in the art would understand that information and signals may be represented using any of a variety of different technologies and techniques. For example, data, instructions, commands, information, signals, bits, symbols, and chips that may be referenced throughout the above description may be represented by voltages, currents, electromagnetic waves, magnetic fields or particles, optical fields or particles, or any combination thereof.

Those of skill would further appreciate that the various illustrative logical blocks, modules, circuits, and algorithm steps described in connection with the embodiments disclosed herein may be implemented as electronic hardware, computer software, or combinations of both. To clearly illustrate this interchangeability of hardware and software, various illustrative components, blocks, modules, circuits, and steps have been described above generally in terms of their functionality. Whether such functionality is implemented as hardware or software depends upon the particular application and design constraints imposed on the overall system. Skilled artisans may implement the described functionality in varying ways for each particular application, but such implementation decisions should not be interpreted as causing a departure from the scope of the present invention.

The various illustrative logical blocks, modules, and circuits described in connection with the embodiments disclosed herein may be implemented or performed with a general purpose processor, a Digital Signal Processor (DSP), an Application Specific Integrated Circuit (ASIC), a Field Programmable Gate Array (FPGA) or other programmable logic device, discrete gate or transistor logic, discrete hardware components, or any combination thereof designed to perform the functions described herein. A general-purpose processor may be a microprocessor, but in the alternative, the processor may be any conventional processor, controller, microcontroller, or state machine. A processor may also be implemented as a combination of computing devices, e.g., a combination of a DSP and a microprocessor, a plurality of microprocessors, one or more microprocessors in conjunction with a DSP core, or any other such configuration.

The steps of a method or algorithm described in connection with the embodiments disclosed herein may be embodied directly in hardware, in a software module executed by a processor, or in a combination of the two. A software module may reside in RAM memory, flash memory, ROM memory, EPROM memory, EEPROM memory, registers, hard disk, a removable disk, a CD-ROM, or any other form of storage medium. An exemplary storage medium is coupled to the processor such the processor can read information from, and write information to, the storage medium. Of course, the storage medium may also be integral to the processor. The processor and the storage medium may reside in an ASIC. The ASIC may reside in a user terminal. Of course, the processor and the storage medium may reside as discrete components in a user terminal.

The application includes headings for reference and to facilitate locating particular sections. These headings are not intended to limit the scope of the concepts described therein under, which concepts may be applied to other portions of the entire specification.

The previous description of the disclosed embodiments is provided to enable any person skilled in the art to make or use the present invention. Various modifications to these embodiments will be readily apparent to those skilled in the art, and the generic principles defined herein may be applied to other embodiments without departing from the spirit or scope of the invention. Thus, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

Claims (17)

1. A method of wireless communication, comprising:

receiving data samples of signals transmitted from a plurality of access terminals;

despreading the received data samples using pilot chips associated with the first access terminal;

demodulating the despread data samples into data symbols;

decoding the demodulated data symbols;

judging whether the demodulated data symbols are correctly decoded or not;

regenerating data chips transmitted by the first access terminal using the decoded demodulated data symbols if the demodulated data symbols are decoded correctly;

despreading the received data samples using regenerated transmitted data chips in place of the pilot chips to determine a channel estimate;

storing data samples received from the plurality of access terminals in a buffer;

reconstructing data samples transmitted from the first access terminal using the determined channel estimates; and

subtracting the reconstructed data samples transmitted from the first access terminal from the buffer stored data samples.

2. The method of claim 1, wherein the pilot chips are covered with a Pseudorandom Noise (PN) sequence.

3. The method of claim 1, wherein the signal comprises a Code Division Multiple Access (CDMA) signal.

4. The method of claim 1, further comprising:

accumulating the despread data samples over a period of time.

5. The method of claim 1, wherein despreading the received data samples with regenerated transmitted data chips to determine channel estimates occurs at each of a plurality of Rake receiver finger delays.

6. The method of claim 1, wherein regenerating the data chips transmitted by the first access terminal comprises at least one of:

recoding, re-interleaving, re-modulating, reapplying data channel gain, and re-spreading the data.

7. The method of claim 1, further comprising:

despreading the received data samples using pilot Pseudorandom Noise (PN) chips associated with a second access terminal;

demodulating the despread data samples into data symbols;

decoding the demodulated data symbols;

judging whether the demodulated data symbols are correctly decoded or not; and

regenerating data chips transmitted by the second access terminal at respective Rake receiver finger delays using the decoded demodulated data symbols if the demodulated data symbols are decoded correctly.

8. A base station, comprising:

a memory storing data samples of signals received from a plurality of access terminals;

a demodulator that despreads and demodulates the stored data samples using pilot chips corresponding to the first access terminal;

a decoder for decoding data in the demodulated data;

a regeneration unit for regenerating the data chips transmitted by the first access terminal using the correctly decoded data;

a channel estimator for despreading the stored data samples with regenerated data chips in place of the pilot chips to determine a channel estimate;

a reconstruction unit for reconstructing the encoded and modulated data samples using the decoded data and the channel estimate; and

a subtractor that subtracts the reconstructed data samples from the stored samples in the memory to reduce interference, thereby enabling the decoder to subsequently decode data for other access terminals in the stored samples.

9. The base station of claim 8, further comprising:

an accumulator for accumulating the despread data samples over a period of time.

10. The base station of claim 8, wherein the demodulator comprises a Rake receiver having a plurality of finger processing units for processing the multipath signal, each finger processing unit having a unique delay for processing samples from the memory.

11. The base station of claim 8, wherein said regeneration unit regenerates data chips transmitted by said first access terminal by at least one of re-encoding, re-interleaving, re-modulating, re-applying data channel gain, and re-spreading.

12. The base station of claim 8 wherein said pilot chips are Pseudorandom Noise (PN) sequences.

13. A method of wireless communication, comprising:

receiving data samples of signals transmitted from a plurality of access terminals;

despreading the received data samples using pilot chips associated with the first access terminal;

demodulating the despread data samples into data symbols;

decoding the demodulated data symbols;

judging whether the demodulated data symbols are correctly decoded or not;

regenerating data chips transmitted by the first access terminal using the decoded demodulated data symbols if the demodulated data symbols are decoded correctly;

determining a time span for uniform regeneration based on the estimated multipath signal delays; and

the received data samples are despread using regenerated transmitted data chips instead of the pilot chips at uniform delays over the determined time span to determine a channel estimate.

14. The method of claim 13, wherein said pilot chips are covered with a Pseudorandom Noise (PN) sequence.

15. The method of claim 13, further comprising:

storing data samples received from the plurality of access terminals in a buffer;

reconstructing data samples transmitted from the first access terminal using the determined channel estimates; and

subtracting the reconstructed data samples transmitted from the first access terminal from the buffer stored data samples.

16. The method of claim 13, wherein determining the time span comprises:

the time span is selected based on information about the multipath signal delays and a priori estimates of the combined response of the transmitter and receiver filters.

17. A wireless communications apparatus, comprising:

a receiving module that receives data samples of signals transmitted from a plurality of access terminals;

a first despreading module that despreads the received data samples using pilot chips associated with a first access terminal;

a demodulation module that demodulates the despread data samples into data symbols;

a decoding module that decodes the demodulated data symbols;

the judging module is used for judging whether the demodulated data symbol is correctly decoded or not;

a regeneration module that regenerates data chips transmitted by the first access terminal using the decoded demodulated data symbols if the demodulated data symbols are decoded correctly;

a time span determination module that determines a time span for uniform regeneration according to the estimated multipath signal delay; and

a second despreading module despreads the received data samples with regenerated transmitted data chips, at uniform delays over the determined time span, in place of the pilot chips, to determine a channel estimate.

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