CN100536450C - Non-zero complex weighted space-time code for multiple antenna transmission - Google Patents

Abstract

The present invention provides a method and apparatus for transmitting phase-hopping and space-time coded signals over multiple antennas (160, 162, 164, 166). The method and apparatus provide for extending an NxN' spatio-temporal block coding to an MxM' spatio-temporal block coding by using phase jumps within the NxN' spatio-temporal block coding symbols, where M>N, whereby Allows space-time block codes to be transmitted on more than N' diversity antennas. The M transmit antennas can achieve the result of M antenna diversity.

Description

Non-Zero Complex Weighted Space-Time Coding for Multiple Antenna Transmission

field of invention

The present invention relates to a method and device for realizing transmission diversity in a wireless communication system, in particular to a method and device for performing non-zero complex weighting and space-time coding on signals for multi-antenna transmission.

background of the invention

With the development of wireless communication systems, the requirements for wireless system design in terms of equipment and performance requirements are gradually increasing. Compared with the first generation of analog and the second generation digital system currently in use, the future wireless system should be the third or fourth generation system. In addition to providing high-quality voice services, it also needs to provide high-quality Data service with high transfer rate. Consistent with the service performance requirements of the system, there are limitations in equipment design, which greatly affects the design of mobile terminals. Third and fourth generation wireless mobile terminals will require smaller, lighter, multiple energy efficient units capable of providing the inherent voice and data services required by these future wireless systems.

Time-varying multipath fading is an effect in wireless systems in which a transmitted signal propagates along multiple paths to the receiver due to the constructive and destructive summation of the receiver signal, causing reception signal attenuation. Currently known methods for overcoming multipath fading effects include the following, for example, time interleaving with error correction codes, frequency diversity by using spectrum diffusion technology, or transmitter energy control technology. However, each of these technologies has drawbacks for use in third and fourth generation wireless systems. Time interleaving introduces unnecessary delay, spectral diffusion techniques require large bandwidth allocations to overcome large coherence bandwidths, energy control techniques require higher transmitter energy than inherent receiver-transmitter feedback techniques, Feedback technology increases the complexity of mobile terminals. All these disadvantages negatively affect the characteristics required to realize third and fourth generation mobile terminals.

Antenna diversity is another technique used to overcome the effects of multipath fading in wireless systems. In diversity reception, two or more physically separated antennas are used to receive a transmitted signal, which is then combined and switched to produce a received signal. Disadvantages of diversity reception are that the required physical separation between the antennas precludes the use of diversity reception on the forward link in new wireless systems and requires small mobile terminals in new wireless systems. Another method for achieving antenna diversity is transmit diversity. In transmit diversity, a signal is transmitted from two or more antennas and then detected by using e.g. maximum likelihood sequential estimator (MLSE), minimum mean square error (MMSE) receiver, maximum a backward receiver or their The approximation is processed at the receiver. Transmit diversity has more practical applications in the forward link of wireless systems, where multiple antennas can be implemented more easily in base stations than in mobile terminals.

The case of transmit diversity with two antennas has been well studied. Alamouti proposes a method for diversity transmission of two antennas that provides a second level of diversity for complex-valued signals. S. Alamouti, "Simple transmit diversity techniques for wireless communications" (IEEE Journal of Selected Areas for Communications, 1451-1458, October 1998). The Alamouti method involves simultaneous transmission of two signals from two antennas during one symbol period. In one symbol period, the signal transmitted from the first antenna is denoted as S0, and the signal transmitted from the second antenna is denoted as S1. In the next symbol period, the signal transmitted from the first antenna is -S1 ^* , and the signal transmitted from the second antenna is denoted as S0 ^* , where ^* is a complex conjugate indicator. A simple diversity transmission system can also be implemented in the coded domain. For example, two copies of the same symbol can be transmitted in parallel using two orthogonal Walsh codes. The same technique can also be used to construct a frequency domain encoding method.

The Alamouti method does not scale directly to more than two antennas. Tarokh et al. have proposed a method for space-time block coding using complex signal constellations for transmission on three and four antennas using rate=1/2 and 3/4. V. Tarokh, H. Jafarkhani, and A. Calderbank, "Space-Temporal Block Coding for Orthogonal Designs" (IEEE Transactions on Information Theory, 1456-1467, July 1999). This approach has the disadvantage of a loss of transmission ratio, and the multilevel nature of the ST-coded symbols increases the peak-to-average ratio required for the transmitted signal, as well as stringent requirements for linear power amplifier design. Another approach to alleviate these problems is "Complex Spatiotemporal Block Coding for 4Tx Antennas" by O. Tirkkonen and A. Hottinen (Journal of Global Communication Systems 2000, Nov. 2000, San Francisco, USA). Other suggested methods include rate=1, Orthogonal Diversity Transmission (OTD) + 4-antenna Space-Time Diversity Transmission (STTD) method. L.Jalloul, K.Kuchi and J.Chen, "Performance Analysis of CDMA Diversity Transmission Method" (Proceedings of IEEE Media Technology Conference, Fall 1999) and M.Harrison and K.Kuchi "High Data Rate Over 2 and 4 Elements Open and Closed Loop Diversity Emissions" (Motorola Contribution 3GPP-C30-19990817-017). This method requires an outer coding and provides second order diversity due to STTD blocks (Alamouti blocks) and second order interleaving gain using OTD blocks. The performance of this method depends on the strength of the external encoding. Since this method requires external encoding, it is not suitable for unencoded systems. For the case of convolutional encoding of rate=1/3, the performance of the ST block encoding method of the OTD+STTD method and the Tarokh rate=3/4 method is the same. Another rate=1 method is covered in O.Tirkkonen, A.BOARIU and A.Hottinen, "Minimum non-orthogonal rate1 space-time block coding for 3+Tx antennas" (ISSSTA Journal, September 2000) . The method suggested in this article achieves high performance but requires a complex receiver.

Therefore, it would be very beneficial to have a method and apparatus that can provide diversity transmission on more than two antennas without increasing the design complexity of the system.

Summary of the invention

The present invention provides a method and apparatus for transmitting non-zero complex weighted and space-time coded signals over multiple antennas. The method and apparatus provide for extending an NxN' spatio-temporal block coding to an MxM' spatio-temporal block coding by using non-zero complex weights of cyclic and signed within the NxN' spatio-temporal block coding, where N is the number of transmission paths, N' is the number of output symbols on each transmission path, where M>N, thus allowing space-time block coding to be transmitted on M multiple antennas. Diversity transmit paths may include separate antennas or beams. The nonce length of the larger code M' may be equal to the nonce length of the original code N'. In the method and apparatus, a transform is performed on an input symbol stream to produce a transform result comprising a spatio-temporal block encoding. Then the N output streams of the spatio-temporal block coding, each comprising N' output symbols, are repeated, and at least one repeated stream is given a non-zero complex weight in time to produce M streams of N' output symbols, with to be transmitted on M diversity transmit paths. Non-zero complex weighting includes phase shifting.

In one embodiment, N is at least 2 and M is at least 3. At least two of the N streams of the N' output symbols, corresponding to the N original streams of the N' output symbols, each of which is then transmitted on the first at least one antenna, and the N' of the N' symbols At least one of the M-N non-zero complex weighted streams is transmitted on one of the second at least one antennas. The first at least one antenna and the second at least one antenna may include any one of the M antennas.

In another embodiment, the method and apparatus can be implemented in a transmitter with common or dedicated pilot channels that enable efficient channel estimation of the coefficients needed to decode space-time codes. In this embodiment, the common and dedicated pilot channels can be implemented individually or in combination in the transmitter. In an alternative to this embodiment, the training sequence is transmitted on N diversity transmission paths, thus making it possible to evaluate N independent diversity transmission paths. To this end, a dedicated pilot channel coding sequence can be multiplexed onto the N streams of N' output symbols coded per original spatio-temporal block, resulting in N streams of N' output symbols and the pilot channel sequence. Cyclic and non-zero complex weights can then be applied to generate an M-phase shifted stream of N' symbols and a pilot channel sequence. At least two of the N original streams of N' output symbols and the pilot channel sequence are then transmitted on one of the first at least one antenna, and at least one of the M-N complex weighted streams of N' output symbols and pilot channel sequences are then transmitted on a second one of the at least one antennas. Another way to be able to estimate N channels is to transmit common pilot channel signals such that N common pilot channel signals are transmitted on each of the first at least one antenna, some M-N of the N common pilot channels A complex weighted copy is then transmitted on one of the second at least one antennas. The common channels on each second at least one antenna use the same complex weighting factors as are used to form the M-N complex weighted streams of N' output symbols from the N original streams of N' output symbols. In these embodiments, the receiver may or may not know the method used to extend an NxN' spatio-temporal block code to an MxN' spatio-temporal block code, and the temporal weighting sequence used.

In another embodiment, N is at least 2, M is at least 3, and the pilot channel can be configured to be able to evaluate at least N+1 diversity transmission paths. At least one of the N streams of N' output symbols, corresponding to the original N streams of N' output symbols, is then transmitted on a first at least one antenna, of the M-N complex weighted streams of N' symbols At least one is transmitted on a second one of the at least one antennas. A different common pilot channel is transmitted on each of the first at least one antenna and at least one of the second at least one antenna. In these embodiments, the receiver requires at least partial knowledge of the method used to extend an NxN' spatio-temporal block code to an MxN' spatio-temporal block code, and the temporal weighting sequence used.

Complex weighting in various embodiments may be achieved by applying a periodic or random complex weighting pattern to each symbol stream that is complex weighted. The relationship between the complex weights of the symbol streams transmitted on the various antennas can also be predetermined.

Brief description of the drawings

Accompanying drawing 1a has shown the block diagram of the transmitter according to one embodiment of the present invention;

Accompanying drawing 1b has shown the block diagram of the part of common pilot channel STTD transmitter according to one embodiment of the present invention;

Accompanying drawing 3 has shown the block diagram of the part of dedicated pilot channel STTD transmitter according to another embodiment of the present invention;

Accompanying drawing 4 has shown the block diagram of the part of the embodiment of a receiver used with the transmitter of accompanying drawing 1;

Accompanying drawing 5 has shown the block diagram of the part of the embodiment of a receiver used with the transmitter of accompanying drawing 2 or accompanying drawing 3;

FIG. 6 illustrates an embodiment of a rake finder of the STTD demodulator 508 of FIG. 5 .

Figure 7 shows a block diagram of part of an STS transmitter according to one embodiment of the present invention;

Accompanying drawing 8 has shown the block diagram of the part of the OTD transmitter according to an embodiment of the present invention;

Figure 9 shows a block diagram of part of an embodiment of a receiver for use with the transmitter of Figure 7;

Figure 10 shows a block diagram of a portion of an embodiment of a receiver for use with the transmitter of Figure 8;

Figure 11 shows a block diagram of part of a long ST block coded transmitter according to one embodiment of the present invention;

Accompanying drawing 12 has shown the block diagram of the part of a common/dedicated pilot channel STTD transmitter according to another embodiment of the present invention;

Figure 13 shows a block diagram of part of a receiver for use with the transmitter of Figure 12; and

FIG. 14 shows a block diagram of the power control portion of a receiver for use with the transmitter of FIG. 12. FIG.

Figure 15 illustrates a constellation diagram defining a phase shifting pattern used in various embodiments of the present invention.

Detailed description of the invention

Referring now to FIG. 1a, there is shown a block diagram of a transmitter 150 in accordance with one embodiment of the present invention. The transmitter 150 includes an input 152 for receiving an input symbol stream, a block coding processor 154 for performing a transform on an input symbol stream to generate a transform result represented by an orthogonal space-time block code, and outputting two The symbol stream of the transformed result, non-zero complex weighter 156 for applying non-zero complex weight to the first of the two symbol streams, non-zero complex weighter 158 for applying non-zero complex weight to the second of the two symbol streams Non-zero complex weights, an RF transmitter 160 for transmitting the first symbol stream at antenna 1, an RF transmitter 162 for transmitting the non-zero complex weighted stream of symbols at antenna 2, an RF transmitter 164 for transmitting the first symbol stream at antenna 2, and an RF transmitter 164 for An RF transmitter 166 is used to transmit the phase-shifted second symbol stream at antenna 4. Antennas 1-4 may be polarized relative to each other to provide enhanced diversity reception. For example, antenna 1 or 2 may be vertically polarized with horizontally polarized antenna 3 or 4 respectively. The transmitter example in Fig. 1a can be realized in various forms, as long as it is suitable for different technologies and systems that can extend a 2xN' block coding to transmit on 4 diversity transmission paths. In transmitter 150, each of the four diversity transmission paths includes a separate antenna, Antenna1-Antenna4. This can include a Code Division Multiple Access (CDMA) system, a Time Division Multiple Access (TDMA) system, or any other digital communication system that can introduce transmit diversity. In an alternative embodiment of FIG. 1a, non-zero complex weighting may be performed entirely on selected transmission paths, resulting in a phase shift between the transmissions from antenna 1 and antenna 2 or antenna 3 and antenna 4. For example, non-zero complex weighting may also be applied prior to input to RF transmitters 160 and 164, producing a non-zero complex-weighted version of each symbol stream, but maintaining a relative phase shift between transmitted signals. An alternate embodiment 150 of transmitter 150 may use less than 4 antennas to achieve 4 paths for diversity transmission. For example, the signals input to RF transmitter 164 or 166 may be concatenated and transmitted in a single antenna. Another alternative is also possible, where less than 4 diversity transmission paths are used, eg only one of the two data streams can be complex-weighted by a non-zero number and transmitted on two diversity paths. In an alternative embodiment of Figure 1a, the non-zero complex weighting operation may be performed in the RF transmitter blocks 160, 162, 164, 166, i.e. the non-zero complex weighting may be performed after baseband filtering of the modulated and space-time coded symbols as A continuous phase scan.

The non-zero complex weighting of these transmissions on antennas 2 and 4 can be performed according to various alternatives. For example, the phase pattern W1(t) = exp(j ^* pi ^* phase-in-degrees/180) used on antenna 2 can be applied, and the phase pattern -W1(t), which is offset by W1(t) The phase is 180 degrees and can be used on antenna 4. Examples of phase shift patterns of angles are {0, 90, 180, 270} on antenna 2 and {180, 270, 0, 90} on antenna 4 on the 4PSK constellation. Figure 15 illustrates a constellation diagram of another phase shifting pattern that can be used in various embodiments of the present invention. The sequence of degrees of movement {0, 135, 270, 45, 180, 315, 90, 225} can be transmitted on antenna 2 while using the movement angle pattern {180, 315, 90, 225, 0, 135 on antenna 4 , 270, 45}. The phase shift can be periodic or random. Periodic phase shift refers to a predetermined phase pattern, eg, periodically repeating complex weights W1(t). The complex weights can be defined such that the sequence of complex weights defines a path of maximum length for consecutive sampling of the active channels as independently as possible. This can create interleaving redundancy and reduce delayed transmission. The pseudo-random phase shift used may be a sequence of random phases selected from an MPSK constellation. Furthermore, when estimating channel coefficients or metrics related to energy control of non-zero complex-weighted channels, another non-zero complex weighting scheme is advantageous where the phase difference between successive phase states is as large as possible Small. In this case, the phase state can cover 360 degrees during the duration of one encoded block. Channel interleaving may also be used in legacy system embodiments. It is also possible to perform non-zero complex weighted sequences and joint interleaving so that the signs of the interleaver outputs are as independent as possible. Furthermore, by changing the relative phases between antennas 1 and 2 and 3 and 4 respectively, the method can be performed so as to have a phase shift or sweep across all antenna elements but not between antennas 1, 2, 3, 4. The phase shift between is maintained. For example, with a phase sweep, have a 50 Hz phase sweep on antenna 1 and a -50 Hz phase sweep on antenna 2 to have an effective 100 Hz sweep. It is similar for antenna 3 and antenna 4.

It is also possible to change the phase rotation every T seconds. The choice of T depends on the total duration of the data symbols and the method used to estimate the channel coefficients. The phase can be kept constant for the total duration occupied by a data symbol, where at least one space-time coding block and corresponding dedicated or common pilot sequence/training sequence can be used for proper channel estimation. The pilot sequence can be a Walsh code as used in CDMA systems, or a sequence of training symbols with better correction properties for channel estimation in TDMA systems. The pilot symbols may have the same non-zero complex weighting coefficients applied to the data in the spatio-temporal block. In addition, pilots can also be transmitted without phase jumps. In this case, the effective channel of the data may be able to be derived jointly from a previously known hopping pattern and a channel estimate obtained from a non-hopping pattern channel. Where non-zero complex weighting is applied to the common pilots, the same or different phase patterns may be applied for the data and common pilots. Channel estimation using non-hopping pilots or training sequences (transmission on common or dedicated channels) provides better channel estimation due to more stable channels.

Referring now to FIG. 1b, there is shown a partial block diagram of a common pilot channel space-time diversity transmission (STTD) transmitter 100 in accordance with an embodiment of the present invention. Transmitter 100 can be used as a 4-antenna diversity transmission extended to Release 99 of the Wideband CDMA (WCDMA) standard for third generation systems. Transmitter 100 includes input 126, block code processor 124, traffic channel symbol stream processing branch inputs 102a-102d, antenna gain blocks 104a-104d, phase shifters 106a and 106b, phase shifter inputs 112a and 112b, code multiplication 108a-108d, pilot sequence processing branch inputs 114a-114d, antenna gain blocks 116a-116d, code multipliers 118a-118d, RF transmitter 128, including RF transmitters 128z-128d, and antennas 1-4.

In Fig. 1b, data to be transmitted is received at input 126, the data comprising a channel code and an interleaved input symbol stream X(t) comprising symbols S1S2. The block coding processor 124 performs coding every two received symbols S1S2 to generate a transform result comprising a 2×2 orthogonal spatio-temporal block coding. In this embodiment, the block encoding processor 124 may perform an Alamouti transform to generate a block encoding, expressed in the following matrix form:

$\left[\begin{array}{cc}{\mathrm{<span\; class="notranslate">\; S</span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}& {\mathrm{<span\; class="notranslate">\; S</span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}\\ <span\; class="notranslate">\; -</span>{\mathrm{<span\; class="notranslate">\; <figure-callout\; id="2"\; label="S"\; filenames="C0280729000221.png,C0280729000231.png"\; state="\{\{state\}\}">S</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 2</span>}}^{<span\; class="notranslate">\; *</span>}& {\mathrm{<span\; class="notranslate">\; <figure-callout\; id="1"\; label="S"\; filenames="C0280729000221.png,C0280729000231.png"\; state="\{\{state\}\}">S</figure-callout></span>}}_{\mathrm{<span\; class="notranslate">\; 1</span>}}^{<span\; class="notranslate">\; *</span>}\end{array}\right]<span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; -</span><span\; class="notranslate">\; (</span>\mathrm{<span\; class="notranslate">\; 1</span>}<span\; class="notranslate">\; )</span>$

The matrix is then divided into 4 streams of two symbols, each of these streams is input to a traffic channel symbol stream processing branch input 102a-102d. As shown in FIG. 1, stream S1S2 is input to 102a, S1S2 is input to 102b, -S2 ^* S1 ^* is input to 102c, and -S2 ^* S1 ^* is input to 102d. Non-zero complex weighting is performed by antenna gain blocks 104a-104d and phase shifters 106a and 106b. The antenna gain of each processing branch is adjusted in antenna gain blocks 104a-104d. When the antenna gain is adjusted, phase shifters 106a and 106b apply a phase shift to the stream S1S2 output by antenna gain block 104b and the stream -S2 ^* S1 ^* output by antenna gain block 104d. Phase shifter control blocks 112a and 112b control phase shifters 106a and 106b by shifting phases using a continuous or discrete phase skip pattern. A CDMA scrambling code is then input to code multipliers 108a-108d, thereby generating stream S1S2 flowing to RF transmitter 128a on antenna 1 and stream S1S2(exp(jΦk1) flowing to RF transmitter 128b on antenna 2 To transmit, the stream -S1 ^* S2 ^* to RF transmitter 128c, transmits on antenna 3, and the stream -S2 ^* S1 ^* (exp(jΦk2), to RF transmitter 128d, transmits on antenna 4. The RF transmitter can Performs baseband pulse shaping, modulation, and carrier upconversion. In some devices, phase hopping or sweeping can optionally be applied after the baseband pulse shaping and modulation steps.

Common pilot channel sequences X1-X4 are input to pilot sequence processing branch inputs 114a-114d. The pilot sequences are then processed through gain blocks 116a-116d and code multipliers 118a-118d, respectively. The encoded outputs of encoded multipliers 118a-118d are then input into RF transmitters 128a-128d of RF transmitter 130, respectively.

Then pilot sequence X1 is transmitted on antenna 1 , pilot sequence X2 is transmitted on antenna 2 , pilot sequence X3 is transmitted on antenna 3 , and pilot sequence X4 is transmitted on antenna 4 .

Reference is now made to FIG. 4, which is a partial block diagram of a receiver for use with transmitter 100 of FIG. 1b. Figure 4 illustrates signal processing in a rake finder in a receiver unit. The received pilot sequences X1-X4 transmitted from transmitter 100 are received and input to channel estimation processors 402a-402d, respectively. Channel estimator 404 then performs a channel estimation function, such as a low pass filtered moving average function for each of channels 1-4. The estimated values of channels 1-4 are then output from output terminals 406a, 406d to adder 410a, phase shifter 408a, adder 410b and phase shifter 408b. Phase shifter 408a receives input from phase shift control block 414a and shifts for channel 2 by the same phase shift estimate used in traffic channel symbols S1S2 transmitted by antenna 2 of transmitter 100 . Phase shifter 408b receives input from phase shift control block 414b and shifts for channel 4 by the same phase shift estimate used in the traffic channel symbols - S2 ^* S1 ^* transmitted by antenna 4 of transmitter 100 . The estimated phase version of channel 2 is combined with the estimate of channel 1 in adder 410a, and the estimated phase version of channel 4 is combined with the estimate of channel 3 in adder 410b. The combined estimate of channels 1 and 2 (412a) and the combined estimate of channels 3 and 4 (412b) are then input to the STTD demodulator 418, which uses the channel estimate to process the received transmission signal. The demodulated signal is then processed in a rake combiner, deinterleaver and channel decoder 420 to produce received symbols S1S2.

In an alternate 4-antenna diversity common pilot channel embodiment, the common pilot channel is phase shifted in the same manner as the traffic channel prior to transmission. Referring now to FIG. 2, which is a partial block diagram of a common pilot channel STTD transmitter 200 according to another embodiment of the present invention. Transmitter 200 includes input 226, block encoding processor 224, traffic channel symbol stream processing branch inputs 202a-202d, antenna gain blocks 204a-204d, phase shifters 206a and 206b, pilot sequence processing branch inputs 241a-214d, Antenna gain blocks 216a-216d, phase shifters 218a and 218b, phase shift control blocks 224a and 224b, code multipliers 220a-220d, code multiplier input 222, RF transmitter 228, including RF transmitters 228a-228d, and Antennas 1-4.

Traffic channel processing and transmission in transmitter 200 is performed in the same manner as traffic channel processing used in transmitter 100 in FIG. 1 . But the transmitter 200 uses the phase-shifted common pilots. The common pilot channel sequence P1 is input to the pilot sequence processing branch input terminals 214a and 214b, and the common pilot channel sequence P2 is input to the pilot sequence processing branch input terminals 214c and 214d. The pilot sequences are then separately processed through antenna gain blocks 216a-216d. The pilot sequence P1 output from the antenna gain block 216a is input to the code multiplier 220a. The pilot sequence P2 output from the antenna gain block 216c is input to the code multiplier 220c. The pilot sequence P2 output from the antenna gain block 216b is input to the phase shifter 218a. The pilot sequence P2 output from the antenna gain block 216d is input to the phase shifter 218b. Phase shifters 218a and 218b perform a phase shift at the controllers of phase shift controllers 224a and 224b, respectively. The phase shift can be the same continuous pattern used in traffic channels or a separate phase-hopping pattern. The phase-shifted pilot sequence P1 output from the phase shifter 218a is then input to the code multiplier 220b, and the phase-shifted pilot sequence P2 output from the phase shifter 218b is then input to the code multiplier 220d. Coded pilot sequence P1 output by coded multiplier 220a is input to RF transmitter 228a for transmission on antenna 1 . The coded phase-shifted pilot sequence P1 output by the coded multiplier 220b is input to the RF transmitter 228b for transmission on antenna 2, and the coded pilot sequence P2 output by the coded multiplier 220c is input to the RF transmitter 228c for transmission on the antenna 3 Transmission, and the coded phase-shifted pilot sequence P output from coded multiplier 220b is input to RF transmitter 228d for transmission on antenna 4.

Phase shifters 218a and 218b may perform phase shifting according to various alternatives, for example, the phase shifting performed in the embodiment of FIG. 1 described above.

Referring now to FIG. 5, which is a partial block diagram of an embodiment 500 of a receiver for use with the transmitter of FIG. Receiver 500 includes channel 1 and channel 2 estimation processing branch input 502a and channel 3 and channel 4 estimation processing branch input 502b, channel estimator 504, STTD demodulator 508, transmit signal input 510 and rake combiner, Interleaver and Channel Decoder 512.

Reception pilot sequences P1 (ch1+ch2Φ) received from antenna 1 and antenna 2 of transmitter 200 on channel 1 and channel 2, respectively, are input to input terminal 502a. The received pilot sequences P2(ch3+ch4Φ) received from the antenna 3 and the antenna 4 of the transmitter 200 on the channel 3 and the channel 4, respectively, are input to the input terminal 502b. The channel estimator 504 performs channel estimation using a function such as a low-pass filtered moving average and outputs a combined estimate for channels 1 and 2 (chest1,2) and a combined estimate for channels 3 and 4 (chest3,4). estimated value. The channel estimate is then input to STTD demodulator 508 which processes the transmission signal received from input 510 using the channel estimate. The demodulated signal is then processed in a rake combiner, deinterleaver and channel decoder 512 to produce received symbols S1S2. FIG. 6 illustrates an embodiment of the rake finder of STTD demodulator 508 of FIG. 5, which uses channels 1,2 and channels 3,4 to demodulate received transmissions.

In another embodiment of 4-antenna diversity, dedicated pilot channels may be employed in a WCDMA version of transmitter 150 in FIG. 1 . Referring now to FIG. 3, which is a partial block diagram of a dedicated pilot channel STTD transmitter 300 according to another embodiment of the present invention. Transmitter 300 includes input 318, block code processor 316, channel symbol stream processing branch inputs 302a-302d, antenna gain blocks 304a-304d, phase shifters 306a and 306b, phase shifter inputs 312a and 312b, code multiplication 308a-308d, code multiplier input 310, and antennas 1-4.

The transmitter 300 in FIG. 3 is a device that uses a dedicated pilot channel, which is transmitted by embedding a pilot sequence into the traffic channel symbol stream. Input 318 and block encoding processor 316 operate in the same manner as input 126 and block encoding processor 124 of FIG. 1 . In transmitter 300, when symbols S1S2 are input to symbol stream processing branch inputs 302a and 302b, pilot channel sequence U1 is input to inputs 302a and 302b, multiplexed between groups of S1S2 symbols. -S2 ^* S1 ^* is input to symbol stream processing branch inputs 302c and 302d, pilot channel sequence U2 is input to inputs 302c and 302d, and multiplexed between symbol groups of -S2 ^* S1 ^* . Another possibility is to define 4 different dedicated pilot sequences, one for each transmit antenna.

The symbol streams multiplexed at inputs 302a-302d are then input to antenna gain blocks 304a-304d, respectively. Channel gains are applied to antenna gain blocks 304a-304d. A stream including S1S2 and pilot sequence U1 is output from antenna gain block 304a to code multiplier 308a. The stream including S1S2 and pilot sequence U1 is output from antenna gain block 304b to phase shifter 306a where the signal stream is phase shifted according to the input of phase shift control block 312a and then input to code multiplier 308b. A stream comprising -S2 ^* S1 ^* and pilot sequence U2 is output from antenna gain block 304c to code multiplier 308c, then the same stream, -S2 ^* S1 ^* and pilot sequence is output from antenna gain block 304d to phase shifter 306b , where the stream is phase shifted according to the input of the phase shift controller 312b, and then input to the encoding multiplier 308d. Code multipliers 308a-308d multiply the appropriate stream by a scrambling code. The coded multiplied stream S1S2 and pilot sequence U1 are then input to RF transmitter 314a for transmission on antenna 1 . The coded, multiplied and phase-shifted stream S1S2 and the pilot sequence U1 are input to the RF transmitter 314b for transmission on the antenna 2. Code multiplied stream - S2 ^* S1 ^* and pilot sequence U2 is input to RF transmitter 314c, transmitted on antenna 3, and code multiplied phase shifted stream - S2 ^* S1 ^* and pilot sequence U2 is input to RF transmitter 314d, transmits on antenna 4. RF transmitters 314a-314d perform modulation and carrier up-conversion prior to transmitting the streams from antennas 1-4. The RF transmitter can perform baseband pulse shaping, modulation, and carrier upconversion. In some devices there is an option to apply non-zero weighting after baseband pulse shaping and modulation.

The receiver in FIG. 5 can be changed to be used in conjunction with the transmitter 300 in FIG. 3 . In this case, receiver 500 may operate similarly, except that inputs 502a and 502b input U1(Ch1+Ch2Φ) and U2(Ch3+Ch4Φ) to channel estimator 504c, respectively.

In another embodiment of 4-antenna diversity, dedicated pilot channels and common pilot channels can be applied in a joint embodiment. Referring now to FIG. 12, which is a partial block diagram of a dedicated/common pilot channel STTD transmitter 1200 according to another embodiment of the present invention.

Transmitter 1200 basically operates in the same manner as transmitter 300 in FIG. 3 , except that common pilot channels are added on antenna 1 and antenna 3 . Common pilot channels P1 and P2 are input to pilot sequence processing branch inputs 1218a and 1218b, respectively. The pilot sequences are then processed through antenna gain blocks 1220a and 1220b and code multipliers 1222a and 1222b, respectively. The codes output from code multipliers 1222a and 1222b are then input to RF transmitters 1214a and 1214c of RF transmitter 1214, respectively. The RF transmitter can perform baseband pulse shaping, modulation, and carrier upconversion. In some devices, non-zero weighting may also optionally be applied after baseband pulse shaping and modulation.

The transmitter in Figure 12 provides non-hopping common pilot channels on antenna 1 and antenna 3 and dedicated pilot channels on antennas 1,2,3,4. The pilot sequences may be multiplexed within a time slot, eg, 15 time slots within a transmission frame in one embodiment. Antenna gain can be set differently for common and private control channels. Antenna gain can also vary over time.

Reference is now made to FIG. 13, which is a partial block diagram of a receiver for use with the transmitter of FIG. 12. FIG. Receiver 1300 includes channel 1 and channel 2 processing branches having inputs 1302a and 1302b and channel 3 and channel 4 processing branches having inputs 1302c and 1302d. Phase shift input 1304 , channel estimator 1306 , STTD demodulator 1310 , transmit signal input 312 and deinterleaver and decoder 1314 .

The received pilot sequences P1, U1, P2 and U2 are input to receiver 1300 at 1302a, 1302b, 1302c and 1302d, respectively. Channel estimator 1306 performs channel estimation, e.g., using a low-pass filter with an averaging function, and outputs a combined estimate for channels 1 and 2 (chest1,2) 1308a, and a combined estimate for channels 3 and 4 (chest3,4) Outputs a pooled estimate. The channel estimate is then input to STTD demodulator 1310, which processes the transmission signal received from input 1312 using the channel estimate. The demodulated signal is then processed in rake combiner, deinterleaver and channel decoder 1314 to produce received symbols S1S2.

One known technique of phase jumping is used for energy control purposes. Referring now to FIG. 14, there is shown a portion of a receiver for estimating energy control in accordance with an embodiment of the present invention. Receiver 1400 includes channel estimator 1402, channel estimation branch inputs 1404a-1404d, phase shifting inputs 1408a and 1408b, phase shifters 1406a and 1406b, channel estimation outputs 1410a and 1410b, square blocks 1412a and 1412b, energy controller 1414.

Channel estimator 1402 calculates channel coefficients for all four antennas based on common or dedicated channels from eg transmitter 1200 during a given time slot "t". It may be a channel prediction for one time slot t+1, alternatively a channel estimate for time implicit t may be used in a slowly varying fading channel. These channel coefficients are denoted as Chanest #1(t), Chanest #2(t), Chanest #3(t), Chanest #4(t) at inputs 1404a-1404d, respectively. For multiple rake finders, for example, Chanest#1(t) is a vector channel estimate corresponding to all rake finders for antenna 1.

Using prior knowledge of the phase jump at the phase shift inputs 1408a and 1408b and knowledge of the channel estimate for the current time slot "t", the channel coefficient for time slot "t+1" can be estimated as:

Chanest#12(t)＝Chanest#1(t)+Chanest#2(t)e ^Φ12(t+1)

Chanest#34(t)＝Chanest#3(t)+Chanest#4(t)e ^Φ34(t+1) (2)

Among them, Φ12 and Φ34 are a known priority.

The received signal energy estimate for time slot (t+1) can be calculated from Chanest#12(t+1) and Chanest#12(t+1):

received_power(t+1)＝||Chanest#12(t+1)|| ² +||Chanest#34(t+1)|| ²

Using the received energy estimate, processor 1414 generates an energy control command.

The method and apparatus of the present invention can also be implemented in diversity in the Walsh code domain. Reference is now made to FIG. 7, which is a partial block diagram of a space-time dispersive (STS) transmitter 700 in accordance with one embodiment of the present invention.

Transmitter 700 is an STS embodiment of transmitter 150 in FIG. 1a in which the spatio-temporal block processor performs the transform in the Walsh code domain. The STS block coding matrix used can be expressed as:

$\left[\begin{array}{cc}S1{\tilde{W}}_1& -\mathrm{<figure-callout\; id="2"\; label="S"\; filenames="C0280729000221.png,C0280729000231.png"\; state="\{\{state\}\}">S</figure-callout>}{2}^{*}{\tilde{W}}_2\\ S2{\tilde{W}}_1& +\mathrm{<figure-callout\; id="1"\; label="S"\; filenames="C0280729000221.png,C0280729000231.png"\; state="\{\{state\}\}">S</figure-callout>}{1}^{*}{\tilde{W}}_2\end{array}\right]$ in ${\tilde{W}}_1=\left[\begin{array}{cc}W1& W1\end{array}\right]$ ${\tilde{W}}_2=\left[\begin{array}{cc}W1& -W1\end{array}\right]---\left(3\right)$

复数加法器710b输出的结果是矩阵行 Each row of the matrix and its phase-shifted version are transmitted on antennas 1-4 respectively, as done in the embodiment of Figure 1a. Symbols S1 and S2 on each row are transmitted synchronously within two symbol periods, rather than sequentially. Data symbols are input to transmitter 700 at input 718 of channel encoder 720 . Channel encoder 720 encodes, punctures, interleaves and forms the input data symbols, and every other encoder outputs symbol S1 as even data and every other encoder outputs symbol S2 as odd data. Even data is then passed through sign receiving blocks 702a, b, e, f, Walsh functional blocks 704b and 704d, Walsh multipliers 706a, b, e, f, adders 708a-708d and complex adders 710a and 710b are deal with. Odd data is processed through sign receiving blocks 702c, d, g, h, Walsh function blocks 704b and 704d, Walsh multipliers 706c, d, g, h, adders 708a-708d and complex adders 710a and 710b . The output of the complex adder 710a is the matrix row

The output of the complex adder 710b is the matrix row

然后被输入到复数乘法器712a，产生

被输入到复数乘法器712b，产生

然后被输入到RF发射机714a，在天线1上传输，

被输入到RF发射机714b，在天线2上传输，

被输入到RF发射机714c，在天线3上传输，以及

被输入到RF发射机714d，在天线4上传输。

is then input to complex multiplier 712a, producing

is input to the complex multiplier 712b, which produces

is then input to RF transmitter 714a for transmission on antenna 1,

is input to RF transmitter 714b, transmitted on antenna 2,

is input to RF transmitter 714c, transmitted on antenna 3, and

is input to RF transmitter 714d for transmission on antenna 4.

Referring now to FIG. 9, which is a partial block diagram illustrating one embodiment of a receiver 900 for use with the transmitter 700 of FIG. The transmitter 700 includes an input 912, Walsh functional blocks 902b and 902d, Walsh multipliers 902a and 902c, channel multipliers 904a-904d, complex adders 906a and 906b, a multiplier (MUX) 908, and an output 910. A received signal is received at input 912 and processed by the STS demodulator. The pilot channel transmission and channel estimation process are the same as in the case of STTD. Channel estimates 904c and 904b are the same as 412a and 412b in Figure 4 for the non-hopping common pilot channel case. For the case of hopping common pilot or dedicated pilot transmissions, the channel estimate can be obtained from the channel estimation module 504 of FIG. 5 . These channel estimates are input to the STS demodulator of FIG. 9 as h1 and h2, h1 corresponding to the combined channel estimate from antenna 1, antenna 2 and h2 corresponding to the channel estimate from antenna 3, antenna 4. After using 902a, b, c, d and 904a, b, c, d and 906a, b for STS demodulation, the output of 908 is the STS demodulated signal, and the signal is sent to the rake combination in Figure 5 , deinterleaver and channel decoder blocks.

The invention suggested here can also be applied in an Orthogonal Diversity (OTD) transmission embodiment of the present invention. Reference is now made to FIG. 8, which is a partial block diagram of an OTD transmitter 800 according to one embodiment of the present invention. Transmitter 800 includes input 822, channel encoder 820, symbol receiving blocks 802a-802d, Walsh functional blocks 804a and 804b, Walsh multipliers 806a-806d, complex adders 808a-808d, complex multiplier 810a and 810b, RF transmitters 812a-812d. The transmitter is an Orthogonal Diversity (OTD) embodiment of transmitter 150 in Figure 1a with a spatio-temporal block coding processor performing the transform in the Walsh code domain. The OTD block coding matrix used can be represented as follows:

$\left[\begin{array}{c}S1{\tilde{W}}_1\\ S2{\tilde{W}}_2\end{array}\right]$ here ${\tilde{W}}_1=\left[\begin{array}{cc}{W}_1& W_1\end{array}\right]$ ${\tilde{W}}_2=\left[\begin{array}{cc}{W}_1& -W_1\end{array}\right]---\left(4\right)$

复数加法器808b输出的结果是

然后被输入到复数乘法器818a，产生

被输入到复数乘法器818b，产生

然后被输入到RF发射机812a，用于在天线1上发射，

被输入到RF发射机812b，用于在天线2上发射，

被输入到发射机812c，用于在天线3上发射，以及

被输入到RF发射机812d，用于在天线4上传输。As done in Figure 1a, each row of the matrix and its phase-shifted version are transmitted to antenna 1-antenna 4 respectively. Data symbols are input to transmitter 800 at input 822 of channel encoder 820 . The channel encoder 820 encodes, punctures, interleaves and forms the input data symbols and every other encoder outputs symbol S1 as even data and every other encoder outputs symbol S2 as odd data. Even data is then processed through sign receive blocks 802a and 802b, Walsh function block 804a, Walsh multipliers 806a and 806b, and complex adder 808a. Odd data is processed through sign receive blocks 802c and 802d, Walsh function block 804b, Walsh multipliers 806c and 806d, and complex adder 808b. The output of the complex adder 808a is

The output of the complex adder 808b is

is then input to complex multiplier 818a, producing

is input to complex multiplier 818b, which produces

is then input to RF transmitter 812a for transmission on antenna 1,

is input to RF transmitter 812b for transmission on antenna 2,

is input to transmitter 812c for transmission on antenna 3, and

is input to RF transmitter 812d for transmission on antenna 4.

Referring now to FIG. 10, a partial block diagram of one embodiment of a receiver 100 for use with the transmitter 800 of FIG. 8 is shown. Transmitter 800 includes input 1010 , Walsh functional blocks 1002 a and 1002 b , Walsh multipliers 1010 a and 1010 b , multipliers 1004 a and 1004 b , multiplier 1006 and output 1008 . The received input signal received at input 912 is demodulated using OTD demodulator 1000 with knowledge of channel coefficients h1 ^* and h2 ^* . The channel coefficients h1 and h2 of this OTD module are derived in the same way as in Fig. 4 and explained in the accompanying drawings. OTD demodulator 1000 is implemented using 1010, 1010a, b and 1012a, b and 1004a, b and 1006. The OTD demodulated output 1008 is sent to the rake combiner, deinterleaver and channel decoding module 512 of FIG. 5 .

The embodiment in Fig. 1 can also be applied in a TDMA transmitter in an EDGE system. Reference is now made to FIG. 11, which is a partial block diagram of a long ST block coded transmitter in accordance with one embodiment of the present invention. Transmitter 1100 includes inputs 1118, 1120, symbol stream processing branch inputs 1116a-1116d, time reversal modules 1102 and 1104, complex conjugate modules 1106a and b, multiplier 1108, phase multipliers 1110a and 1110b, phase multiplication control Modules 1112a and 1112b and antennas 1,2,3,4. Channel encoder 1120 encodes, punctures, interleaves, and forms a symbol stream received at receiver 1118 . Channel encoder 1120 also splits the incoming symbol stream into odd and even data streams. Even data streams are input to branch input 1116a and RF transmitter 1122a for transmission on antenna 1 during the first half of a data pulse, odd data streams are input to branch input 1116c and RF transmitter 1112c, for transmitting on antenna 2 during the first half period of the data pulse. During the second half of a pulse, the even data stream is input to branch input 1116b, time is received at time reversal block 1102, complex conjugated at complex conjugation block 1106a and sent to RF transmitter 1122c , for transmitting on antenna 3. During the second half of the data burst, the odd data stream is input to branch input 1116d, time is received at time reversal block 1104, complex conjugated at complex conjugation block 1106b, and multiplied at multiplier 1108 by A negative number is then sent to RF transmitter 1122d for transmission on antenna 4. A training sequence SEQ1 is embedded in the middle of the transmission pulse of antenna 1, and a training sequence SEQ2 is embedded in the middle of the transmission pulse of antenna 2. Phase multipliers 1112a and 1112d phase shift the input to RF transmitters 1122b and 1122d using multiplication blocks 1110a and 1110b, respectively. The output of phase multiplier 1112a is then input to RF transmitter 1112b for transmission on antenna 2 and the output of phase multiplier 1112b is input to RF transmitter 1122d for transmission on antenna 4. The RF transmitter can perform baseband pulse shaping, modulation, and carrier upconversion. In some devices, phase multiplication may optionally be applied after the baseband shaping and modulation steps.

The phase rotation applied in phase multipliers 1122a and 1122b may remain constant over the length of the pulse, with the phase changing on a pulse-by-pulse basis. The phase can be chosen periodically or randomly from a previously explained MPSK constellation. In a preferred embodiment, the phase rotation on antenna 4 remains the same as the phase rotation on antenna 2 by a phase shift of 180 degrees or multiplied by -1. Phase multiplication can be performed before or after baseband shaping. In an alternative embodiment of FIG. 11, the transmissions from antenna 1 and antenna 3 can be internally charged.

The transmitter shown in Fig. 3 can also be used in EDGE with some modifications. The space-time coding described at 316 is applied blockwise rather than symbolically in an EDGE device. The block length can be chosen to be the first half of the pulse. In EDGE, the length of the first half of the pulse and the length of the second half equals 58 symbols. In this case, S1 and S2 represent a symbol block, and () ^* represents the time reversal and negative conjugate operation of the symbol block. S1 ^* indicates that the symbol block S1 has been time reversed and complex conjugated. -S2 ^* indicates that the sign block S2 has been sign-reversed, complex conjugated and multiplied by -1.0. Pilot sequences U1 and U2 can be chosen as two training sequences, eg known CAZAC sequences. Diffusion codes 308a, b, c, d are not applied to EDGE. Phase multiplication blocks 306a and 306b are retained.

A receiver designed for 2-antenna space-time block coding can also use the embodiment of the receiver in Fig. 1 or Fig. 2 .

Based on the above description and embodiments, those skilled in the art should be able to appreciate that although the method and apparatus of the present invention are described with reference to specific embodiments, it should be understood that various changes can be made to the embodiments described above. With modifications and variations, various embodiments of the invention can be practiced without departing from the spirit and scope of the invention as defined in the following claims.

Claims (22)

1. A method for transmitting signals from a plurality of antennas, comprising: receiving a transform result, the transform result is generated by performing a transform on the input symbol stream, the transform result includes an N×N' orthogonal space-time block coding, and generates N first signals, among the N first signals Each of consists of N' symbols; and In time, at least one of the N first signals of the transformation result is subjected to a non-zero complex weighting to generate at least one second signal, each of which is relative to the one from which it is generated One of the N first signals has a phase shift; said N first signals and at least one second signal together comprise M signals, wherein M is greater than N; wherein said non-zero complex weights are prepared for substantially simultaneously transmitting each of said N first signals of said transformation result on a respective one of said first N antennas, on a respective one of a second at least one antenna Each of said at least one second signal is transmitted on one, thereby transmitting said stream of symbols on M transmit diversity paths. 2. The method of claim 1, wherein said non-zero complex weighting comprises applying a phase shift to at least one of said N first signals according to at least a predetermined hopping sequence, the application comprising: according to an angle of 0, 135, 270, 45, 180, 315, 90, 225 of a first predetermined hopping sequence to apply a phase shift to at least one of said N first signals, and, according to at least an angle of 180, 315, 90, 225, 0, A second predetermined hopping sequence of 135, 270, 45 to apply a phase shift to at least one of said N first signals. 3. The method of claim 2, wherein said N first signals comprise a first first signal and at least a second first signal, and wherein said first first signal is phase shifting, and phase shifting the second first signal according to the second predetermined hopping sequence. 4. The method of claim 3, wherein the phase shift jumps of the first predetermined hopping sequence and the second predetermined hopping sequence are sequenced at successively selected intervals, wherein during said non-zero complex weighting period, the first first signal and the second The two first signals are phase shifted according to the phase shift hopping of the first predetermined hopping sequence and the second predetermined hopping sequence, respectively. 5. The method of claim 4, wherein the successively selected intervals comprise periodic intervals during which the first first signal and the second first signal are sequenced phase-shifted. and the phase shift hop of the second predetermined hop sequence. 6. The method of claim 3, wherein the phase shift hops of the first predetermined hopping sequence and the second predetermined hopping sequence are sequenced at random intervals, wherein during said non-zero complex weighting, the first first signal and the second The first signals are phase shifted according to the phase shifting hops of the first predetermined hopping sequence and the second predetermined hopping sequence, respectively. 7. The method of claim 1, wherein said non-zero complex weighting comprises applying a phase shift to at least one of said N first signals according to at least a predetermined hopping sequence, and wherein the phase shift of the predetermined hopping sequence jumps Sequenced at successively selected intervals, wherein at least one of said N first signals is phase-shifted during said non-zero complex weighting according to a phase-shifting hop of said predetermined hopping sequence. 8. The method of claim 7, wherein the continuously selected interval comprises a periodic interval during which the predetermined hopping sequence of the phase-shifted at least one of the N first signals is sequenced during the continuously selected interval Phase shift jumps. 9. The method of claim 1, wherein said non-zero complex weighting comprises applying a phase shift to at least one of said N first signals according to at least a predetermined hopping sequence, and wherein the phase shift of the predetermined hopping sequence hops by Random interval ordering, wherein during said non-zero complex weighting at least one of said N first signals is phase shifted according to a phase shift hop of the predetermined hopping sequence. 10. The method of claim 1, wherein said input symbol stream comprises symbols S ₁ , S ₂ , and said space-time blocks comprise 2×2 space-time block codes, said N first signals comprising Streams _S1 and _-S2 * emitted, and streams _S2 and _S1 * emitted at t1 and t2, respectively. 11. An apparatus for transmitting a signal, said apparatus comprising: at least one weighter, configured to receive a transformation result, the transformation result is generated by performing transformation on the input symbol stream, the transformation result includes an N×N' orthogonal space-time block coding, and generates N first signals , each of the N first signals includes N' symbols, Wherein, the at least one weighter is further configured to perform non-zero complex weighting on at least one of the N first signals of the transformation result in time, so as to generate at least one second signal, and the at least one Each of the second signals is phase-shifted with respect to one of the N first signals from which it was generated, said N first signals and at least one second signal together comprising M signals, where M is greater than N, wherein said non-zero complex weights are prepared for substantially simultaneously transmitting each of said N first signals of said transformation result on a respective one of said first N antennas, on a second at least one antenna Each of said at least one second signal is transmitted on a corresponding one, thereby transmitting said symbol stream on M transmit diversity paths. 12. The apparatus of claim 11 , wherein said non-zero complex weighting comprises applying a phase shift to at least one of said N first signals according to a predetermined hopping sequence, the application comprising: according to an angle of 0, 135, 270, A first predetermined hopping sequence of 45, 180, 315, 90, 225 to apply a phase shift to at least one of said N first signals, and, according to at least an angle of 180, 315, 90, 225, 0, 135, 270, a second predetermined hopping sequence of 45 to apply a phase shift to at least one of said N first signals. 13. The apparatus of claim 12, wherein said N first signals comprise a first first signal and at least a second first signal, and wherein the first first signal is based on the first predetermined hopping sequence phase shifted, and the second first signal is phase shifted according to the second predetermined hopping sequence. 14. The apparatus of claim 13, wherein the phase shift hops of the first predetermined hopping sequence and the second predetermined hopping sequence are sequenced at successively selected intervals, wherein the first first signal and the second first signal are respectively determined by the The at least one weighter is phase shifted according to the phase shifting hops of the first predetermined hopping sequence and the second predetermined hopping sequence. 15. The apparatus of claim 14 , wherein the successively selected intervals comprise periodic intervals during which the first first signal and the second first signal are sequenced phase-shifted. and the phase shift hop of the second predetermined hop sequence. 16. The apparatus of claim 13, wherein the phase shift hops of the first predetermined hopping sequence and the second predetermined hopping sequence are ordered at random intervals, wherein the at least one weighter is based on the first predetermined hopping sequence and the second predetermined hopping sequence The phase shift jumps of , respectively, phase shift the first first signal and the second first signal. 17. The apparatus of claim 12, wherein said at least one weighter comprises a first weighter for non-zero complex number weighting of a first first signal of said N first signals, and at least one second weighter, The second weighter is used for weighting the second first signal of the N first signals with a non-zero complex number. 18. The apparatus of claim 17, wherein said first weighter phase shifts said first first signal according to a first predetermined hopping sequence, and wherein said second weighter phase shifts said first signal according to a second predetermined hopping sequence. The phases of the two first signals are shifted. 19. The apparatus of claim 11 , wherein said non-zero complex weighting comprises applying a phase shift to at least one of said N first signals according to at least a predetermined hopping sequence, and wherein a phase shift of the predetermined hopping sequence jumps Sequencing at successively selected intervals, wherein at least one of said N first signals is phase-shifted by said at least one weighter according to phase-shifting hops of said predetermined hopping sequence. 20. The apparatus of claim 19 , wherein the continuously selected interval comprises a periodic interval during which the phases of the predetermined hopping sequence of phase-shifting at least one of the N first signals are sequenced Move and jump. 21. The apparatus of claim 11 , wherein said non-zero complex weighting comprises applying a phase shift to at least one of said N first signals according to at least a predetermined hopping sequence, and wherein the phase shift of the predetermined hopping sequence jumps Sequenced at random intervals, wherein at least one of the N first signals is phase-shifted by the at least one weighter according to phase-shift hopping of the predetermined hopping sequence. 22. The apparatus of claim 11, wherein said input symbol stream comprises symbols S ₁ , S ₂ , and said space-time blocks comprise 2×2 space-time block codes, said N first signals comprising Streams _S1 and _-S2 * emitted, and streams _S2 and _S1 * emitted at t1 and t2, respectively.

Images