CA1123225A - Compensated capacitive transducer demodulator circuit - Google Patents
Compensated capacitive transducer demodulator circuitInfo
- Publication number
- CA1123225A CA1123225A CA319,755A CA319755A CA1123225A CA 1123225 A CA1123225 A CA 1123225A CA 319755 A CA319755 A CA 319755A CA 1123225 A CA1123225 A CA 1123225A
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- CA
- Canada
- Prior art keywords
- demodulator
- voltage
- transducer
- capacitive transducer
- carrier
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
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Classifications
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01D—MEASURING NOT SPECIALLY ADAPTED FOR A SPECIFIC VARIABLE; ARRANGEMENTS FOR MEASURING TWO OR MORE VARIABLES NOT COVERED IN A SINGLE OTHER SUBCLASS; TARIFF METERING APPARATUS; MEASURING OR TESTING NOT OTHERWISE PROVIDED FOR
- G01D5/00—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable
- G01D5/12—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means
- G01D5/14—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage
- G01D5/24—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage by varying capacitance
- G01D5/241—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage by varying capacitance by relative movement of capacitor electrodes
- G01D5/2417—Mechanical means for transferring the output of a sensing member; Means for converting the output of a sensing member to another variable where the form or nature of the sensing member does not constrain the means for converting; Transducers not specially adapted for a specific variable using electric or magnetic means influencing the magnitude of a current or voltage by varying capacitance by relative movement of capacitor electrodes by varying separation
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01L—MEASURING FORCE, STRESS, TORQUE, WORK, MECHANICAL POWER, MECHANICAL EFFICIENCY, OR FLUID PRESSURE
- G01L9/00—Measuring steady of quasi-steady pressure of fluid or fluent solid material by electric or magnetic pressure-sensitive elements; Transmitting or indicating the displacement of mechanical pressure-sensitive elements, used to measure the steady or quasi-steady pressure of a fluid or fluent solid material, by electric or magnetic means
- G01L9/0041—Transmitting or indicating the displacement of flexible diaphragms
- G01L9/0072—Transmitting or indicating the displacement of flexible diaphragms using variations in capacitance
- G01L9/0075—Transmitting or indicating the displacement of flexible diaphragms using variations in capacitance using a ceramic diaphragm, e.g. alumina, fused quartz, glass
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01L—MEASURING FORCE, STRESS, TORQUE, WORK, MECHANICAL POWER, MECHANICAL EFFICIENCY, OR FLUID PRESSURE
- G01L9/00—Measuring steady of quasi-steady pressure of fluid or fluent solid material by electric or magnetic pressure-sensitive elements; Transmitting or indicating the displacement of mechanical pressure-sensitive elements, used to measure the steady or quasi-steady pressure of a fluid or fluent solid material, by electric or magnetic means
- G01L9/0082—Transmitting or indicating the displacement of capsules by electric, electromechanical, magnetic, or electromechanical means
- G01L9/0086—Transmitting or indicating the displacement of capsules by electric, electromechanical, magnetic, or electromechanical means using variations in capacitance
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01L—MEASURING FORCE, STRESS, TORQUE, WORK, MECHANICAL POWER, MECHANICAL EFFICIENCY, OR FLUID PRESSURE
- G01L9/00—Measuring steady of quasi-steady pressure of fluid or fluent solid material by electric or magnetic pressure-sensitive elements; Transmitting or indicating the displacement of mechanical pressure-sensitive elements, used to measure the steady or quasi-steady pressure of a fluid or fluent solid material, by electric or magnetic means
- G01L9/12—Measuring steady of quasi-steady pressure of fluid or fluent solid material by electric or magnetic pressure-sensitive elements; Transmitting or indicating the displacement of mechanical pressure-sensitive elements, used to measure the steady or quasi-steady pressure of a fluid or fluent solid material, by electric or magnetic means by making use of variations in capacitance, i.e. electric circuits therefor
- G01L9/125—Measuring steady of quasi-steady pressure of fluid or fluent solid material by electric or magnetic pressure-sensitive elements; Transmitting or indicating the displacement of mechanical pressure-sensitive elements, used to measure the steady or quasi-steady pressure of a fluid or fluent solid material, by electric or magnetic means by making use of variations in capacitance, i.e. electric circuits therefor with temperature compensating means
Landscapes
- Physics & Mathematics (AREA)
- General Physics & Mathematics (AREA)
- Engineering & Computer Science (AREA)
- Power Engineering (AREA)
- Chemical & Material Sciences (AREA)
- Ceramic Engineering (AREA)
- Transmission And Conversion Of Sensor Element Output (AREA)
- Measurement Of Resistance Or Impedance (AREA)
- Indication And Recording Devices For Special Purposes And Tariff Metering Devices (AREA)
- Arrangements For Transmission Of Measured Signals (AREA)
- Investigating Or Analyzing Materials By The Use Of Electric Means (AREA)
- Measuring Fluid Pressure (AREA)
- Measurement Of Length, Angles, Or The Like Using Electric Or Magnetic Means (AREA)
Abstract
ABSTRACT OF THE DISCLOSURE
A compensation technique and network circuitry for a quad-diode demodulator and capacitive transducer combi-nation is disclosed. The compensation method includes varying the amplitude of An alternating carrier frequency oppositely to the changes produced in that amplitude by the compensable errors in the demodulator and transducer.
In one preferred embodiment ratiometric compensation and temperature compensation for the demodulator and any capacitive transducer is produced. In a second embodiment linearization and temperature compensation for the demodulator and a quartz capacitive transducer is provided.
A compensation technique and network circuitry for a quad-diode demodulator and capacitive transducer combi-nation is disclosed. The compensation method includes varying the amplitude of An alternating carrier frequency oppositely to the changes produced in that amplitude by the compensable errors in the demodulator and transducer.
In one preferred embodiment ratiometric compensation and temperature compensation for the demodulator and any capacitive transducer is produced. In a second embodiment linearization and temperature compensation for the demodulator and a quartz capacitive transducer is provided.
Description
~Z3Z2~ii ., BACKGR~UND OF THE INV~NTION
l. Pield of the Invention The invention pert~ins generally to demodulation circuits for transducers and ls more particu-larly directed to compensation circuitry for a quad-diode demodulator and capacitive trsnsducer combination.
l. Pield of the Invention The invention pert~ins generally to demodulation circuits for transducers and ls more particu-larly directed to compensation circuitry for a quad-diode demodulator and capacitive trsnsducer combination.
2. Prior Art Capacitive transducers are useful devices for measuring physical parameters such as pressure, distance, surface roughness, angle change or the like ~nd come in an almost limitless variety of shapes, sizes and configu-rations. A change in the sensed parameter will cause the transducer to vary its capacitance accordingly in a proportional or known functional manner. The change in capacitance of the transducer can thereafter be utilized in a number of ways to generate an electrical signal representative of the change in the physical parameter.
Generally, for capacitive transducers, the generation of the electrical signal is accomplished by the modulation of an alternating carrier frequency where one of the characteristics o the carrier is modified electrically by the vari~ble capacitance of the transducer. The carrier is thereafter detected or demod-ulated to obtain the intelligence contained and thereby generates a useful electrical signal representative of the sensed parameter. The circuit used for modulating the carrier is usually in integral combination with the demodulator cirsuit with the whole being termed herein- ;
af ter a transducer demodulator .
Particularly, such a ~ransducer demodulator circuit using a quad-diode configuration is shown in U.- S. Patents 3,318,153; 3,271, 669 issued to T. Lode. These circuits are particularly useful for capacitive pressure transducers. Another advantageous type of capacitive transducer demodulator 1.
~ ' .
1~L23Z25 that has been recently developed is the quad-diode bridge circuit. An example of which i~ illustrated in U.S.
Patents 3,883,812; 3,869,676 issued to Hnrrison et al.
The desirable characteristics o~ this demodulator include a sufficient magnitude o~ output which is relatively independent of excitation waveform a`nd requency. Addi-tionally, this transducer demodulator pro,vides excellent resolution for the change ln capacitance of the transducer and allows the transducer to be conveniently grounded. These are fehtures that will provide for greater use of this circuit in mul~i-farious transducer applications in the future. These circuits and others of their general type will herein further be termed quad-diode demodulators because of their circuitry utilizing four rectification devices or diodes.
If the capacitive transducer is a pressure transducer, a capa~itive transducer and demodulator combination, as deseribed above, can be utilized for sensing manifold absolute pressure (MAP~ changes in an internal combustion engine. The electrical signal obtained from the combination can then be used as is conventional to regulate functional aspects of the engine operation such as air/fuel ratio, timing, EGR, etc. when sensed with other engine parameters. In the automotive environment the convenient grounding of a capacitive transducer is an important feature which allows a direct conn~ction to the chassis and eliminates the problems of isolating a transducer with reference potential that is above or below chassis ground. Also, the referenced Harrison quad-diode circuit lends itself to remote transducer applications which probably will accompany many new developments in auto-motive elec~ronics.
However, there are still problems with using these quad-diode demodulators in harsh environments such as that found in the engine compartment of an automobile.
~23~ZS
The range o~ temperatures through whch the transducer-demodulator circuitry is subjected to i-9 extreme (-40F.
to 120 C.) ~nd the circuitry must, therefore, be provided with accurate temperature compensatlon. This problem Is complicated by the inclusion of the neeessary but non-linear diodes in the demodulator circuitry. These diodes will pro-duce not only different voltage drops for d~ffere~t temperatures but will also produce different voltage drops at the same temperature when different currents are conducted.
Another problem found in many environments but which is particularly troublesome in the automotive environment is the regulation of the power supply. With constantly changing demands on a limited battery and only a rough regulation from the voltage regulator for alternator voltage changes, surges and voltage drops of significant magnitude are not uncommon. Transducer electronios where the informatio~ is contained within the amplitude of transducer signal and changes with respect to a reference are particularly affected by these changes.
One method developed for overcoming this problem is ratiometry. This method contemplates that the output of a particular circuit will change in accordance with the ehanges in the power supply to always remain a predeter-mined percentage of the power supply for non-signal condit~ons. Thus, when a plurality of these circuits are connected together signal information will not be lost and errors will not be introduced because of the regu-lation problems of the power supply. Therefore, when operated in an automotive or other environments where regulation problems are prevalent, the quad-diode demod-ulator and capacitive transducer combination should be provided with ratiometric compensation for facile connec-tion to other system circuitry. Compensation for ratiometric errors is difficult because of the non-linear nature of the diodes of the demodulator which cause an error.
.
~Z3225 In certQin instances it is ~ust as import~nt to compensate for the transducer itself as it is to compensate the demodulstor circuitry. For ex~mple, quartz CQpaCitor transducers are relatively accurate and S inexpensive but they are temperature sensitive and some have linearity problems for reasonably priced transducers. It would be extremely advantageous to compensate a low cost quartz capaeitive transducer to provide a linear output without temperature dependency 10 while retaining the desirable features of a quad-diode demodulator.
SUMMARY OF THE INVENTION
The invention provides a compensation technique for a quad-diode demodulator snd cspacitive transducer combi-15 nation. The technique includes controllably varying the amplitude of an alterneting carrier frequency oppositely to changes produced in that amplitude by the compensable errors in the demodulator and transducer and thus effecting their cancellation.
The technique produces an extremely flexible compen-sation method where many different types of errors may bs compensated for without drastic modification of the demodulator circuitry. Moreover, compensation may be effected without disturbing the desirable characteristics of the quad-diode demodulator and capacitive transducer combination.
In a first preferred embodiment the technique is implemented by compensation network circuitry comprising means for ratiometric compensation and means for tempera-ture compensaeion of the quad-diode demodulator. A
voltage follower circuit is utilized to ~ompare and maintain the equivalency between a ratiometric voltage and the input voltage of a frequency generator minus a non-linear compensation voltage.
~L232;2 5 In the first embodiment the compensation voltage is generated as the complement of the non-linear non-ratiometric error attenuation introduced by the demodulator circuit because of the diodes that comprise the demodulation bridge. The in-put voltage to the frequency generator is therefore the ratio-metric voltage plus the non-linear non-ratiome~ric compensation voltage which compensates for the attenuation of the diodes in the demodulator. Since the diode voltage attenuation in the bridge is temperature dependent, the compensation voltage being the complement of the attenuation is also temperature dependent and produces a demodulator output that has the temperature term canc-eled. The compensation voltage is developed in this embodi-ment by drawing an exact amount of current through a temperature sensitive means. The temperature sensitive means have a temp-erature characteristic identical with the temperature character-istic of the demodulator diodes and of equivalent voltage mag-nitude. Thus, the voltage magnitude of the temperature sensi-tive means cancels the ratiometric error and changes with temperature to cancel the temperature dependency.
In a second preferred embodiment the technique is implemented by compensation network circuitry comprising means for temperature compensation and means for linearization of a quartz capacitive pressure transducer. A voltage follower cir-cuit is utilized to compare and maintain the equivalency between a ratiometric voltage and the input voltage of a frequency generator mlnus a non-linear compensatlon voltage.
In the second embodiment the compensation voltage is generated as the combination of a temperature dependent com-ponent and a non-linear component~ The temperature dependent component of the compensation voltage is used to compensate for the temperature dependency of the quartz capacitive transducer and the non-linear component for the non-linear response of the transducer.
The temperature dependent component in this embodi ment is developed by drawing an exact amount of current jk/ ~
~.Z3ZZS
~~rough a temperature sensitive means with a substantially linear temperature characteristic. The linear change in voltage with respect to temperature of the device is used as a slope multiplier to cancel the change in the output of the quartz capacitive transducer with respect to temperature. The non-linear component is generated in proportion to the inverted output voltage of the demodulator and transducer combination.
The non-linear component is then provided as negative feedback to reduce the input voltage to the frequency generator and cancel the non-linear increases in the quartz capacitive transducer.
In summary, therefore, the present invention may be seen as providing a capacitive transducer and demodulator com-bination circuit comprising: a capacitive transducer for measuring the variances of a physical parameter, the transducer adapted to transform the variances into changes of capacitance of the transducer; a frequency generator means for suppiying an alternating carrier voltage; a quad-diode demodulator electrically connected to receive fro~. the frequency generator means the alternating carrier voltage and further connected to the trans-ducer such that the changes in capacitance modulate the carrier, the demodulator detecting the carrier and generating an output voltage signal representative of the variances in the measured physical parameter; and compensation network means, electrically connected to the frequency generator, for varying the amplitude of the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodula-tor and transducer.
The present invention furthermore may be seen to provide a transducer and demodulator combination circuit com-prising: a transducer for measuring the variances of a physical parameter, the transducer adapted to transform the variances into changes of impedance of the transducer; a sd ~ -7-~L~Z3~25 Frequency generator means for supplying an alternating carrier voltage which is a ratiometric function of an input supply voltage; a deMoclulator electrically connec-ted to receive from the frequency generator means the alterna-ting carrier voltage and further connected to the transducer such tha1: -the chancJes in impedance modulate the carrier, the demodulator detecting the carrier and generating an output vol~age signal representa-tive of the variances in the measured physical parameter; and compensation network means for varying the amplitude o the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodulator and transducer wherein the compensation network means includes means for providing ratiometric compensation for the ratiometric error of the demodulator.
Thesè and other features, and aspects of the invention will be more fully understood and better appreciated from a reading of the following detailed disclosure taken in conjunction f with the appended drawings wherein:
BRIEF DESCRIPTION OF THE DRAWINGS
_ The prior art figure is a schematic view of a quad-diode demodulator circuit and capacitive transducer combination;
Figure l is a block diagram of a quad-diode demodulator circuit and capacitive transducer combination provided with a compensation network according to the invention;
Figure 2 is a detailed block diagram illustrating the quad-diode demodulator and capacitive transducer combination shown in Figure l, and provided with ratiometric and temperature compensation for the quad-diode demodulator; .:~
Figure 3, appearing on the same sheet as Figure 6, is a detailed block diagram illustrating the quad-diode demodula-tor and capacitive transducer combination shown in Figure l, and provided with temperature compensation and llnearization com--sd/~6 -8-ensation for a quartz capacitive transducer;
Figure 4 is a detailed schematic circuit of the block diagram illustrated in Figure 2;
Figure 5 is a detailed schematic circuit of the block diagram illustrated in Figure 3;
Figure 6 is a cross sectional view of c~ quartz capacitive pressure transducer taken along section line 6 in Figure 7;
Figure-7 is a cross sectional side view of a quartz capacitive pressure transducer in a quiescent state;
Figure 8 is a cross sectional side view of the ~-quartz capacitive pressure transducer in a state of flexure;
Figure 9 is an illustrative graphic diagram of a ratiometric conversion;
Figure 10 is an illustrative graphic diagram of temperature versus the voltage across a signal diode for different current levels; and . 5 sd/~ 8A-. .
- , ~ . . .
~Z3~ZS
g Figures 11 through 16 are graphic illustrations of functional relations at various point~ throughout the circuitry of ~igure 5.
The convention that like reference numeral refer to identical elements throughout the figures has been maintained for facilitating B clear description of the invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
Illustrated in the prior art figure is a quad-diode demodulator ior the detection of capaciti~e changes in a transducer caused by the variance of a physical par~meter. The demodulator comprises generally a frequency generator 10, a quad-diode bridge circuit 12, and an output circuit 14. The fre~uency generator 10 produces an alternating carrier frequency of an amplitude Vp to the junction of two coupling capacitors Ca and C~
of the bridge circuit 12. The other terminals of the coupling capacitors Ca and Cb are connected respectively between first and second diode bridge nodes ~ormed by diodes Dl, D3 and diodes D2, D4, and ground. The diodes ~Dl, D2, D3, and.D4 form.a quad-diode bridge and have parallel capacitors Cr, C connected between a third bridge node formed by the diodes Dl, D2, and ground, A
fourth bridge node formed by the diodes D3, D4, has a capacitor Cp connected between the node and ground.
Conventionally, Cp is representative of the variable capacitance of a capacitive transducer and Cr is a capacitance of a value that is equivalent to the zero point value of the capacitive trflnsducer. The capacitance Cr may be contained within the tr~nsducer for compensation or may be supplied ex~ernally. The capscitor Cs is provided as a small variable c~pacitor in parallel with Cr to enable exact zeroing o~ the initisl output of the capAcitive trarsducer. ~he total ~23~2s --, o-- .
.
capacit~nce Cs and Cr is substantially equivalent to Cr and this value will be used for further description when referring to these in combination.
Consider now one half of the circuit where values of C~7 Cb are much, much greater than Cp, Cr, Assuming Ca is charged, on transitions of the frequency generator 10, the diode Dl cond~cts and C charges to a voltage dependent upon the value of Cr si~ce C >~ Cr. Ca therefore discharges by an amount of charge dependent upon the v~lue of C .
Subsequently, on negatlve transitions, diode D3 conducts and the capacitor Cp charges an amount deter-mined by the value of Cp since C~>>CpCa charges by an equivaIent amount. If the capacitors Cp snd Cr are equal, then there is no net signal voltage left on Ca.
However~ if Cp changes as the result of sensing a change in a physical parameter, then a net DC voltage or charge will remain on capacitor CQ and appear as a signal output at point A. The signal voltage developed is a function of the difference of the values of capacitors Cp and Cr.
In a similar manner, diodes D2, D4 in cooperation with capacitors Cr and Cp will produce a net DC voltage at point B. This voltage will be equivalent to the voltage at point A but of opposite polarity. At a steady state condition, equal amounts of charge are transferred between Cp and Cr and $he voltage output from the demodulator will reduce to C - C
Vdc = p d C~~~~~~~ (1) p r where Vdc is the output volta~e measured across points AB, Vp is the voltage amplitude of the frequency generator, and Vd is the voltage drop across one of the diodes Dl - D4.
The output circuitry 14~ comprising an impedance Ra connected between point A and an output terminal 15 and an impedance Rb connected betwPen point B ~nd a ground terminal 17, provides a single ended output from the ~2~ZZ5 ~, differential voltages presented at points A and B
respectively. A capacitor Cd is provided by connecting one lead to the terminnl 15 and thè other lead to terminal 17 for coacting with the impednaces R~, Rb.
This demodulator circuit is more fully described in the two before referenced U.S. Patents issu0d to Harrison, the disclosure of which is expressly incorporated herein by reference.
With attention now directed to Figure 1, there is shown to advantage in block form a preferred quad-diode demodulator circuit including a frequency generator 10, quad-diode bridge circuit 12, output circuit 14, and capucitive transducer 18, the function and operation of which are identical to that described in the above prior art figure. In accordance with the invention, a compen-sation network 16 is connected to the frequency generator 10 for providing a controllable amplitude for the carrier frequency Vp to cancel a compensable error from the demodulator-transducer combination.
In theory, the output voltage of the demodulator and transducer combination can be written as a transfer function:
Vout/Vp = G~s) ~2) where VOUt is the output ~mplitude of the voltage at a5 terminals 15, 1~, Vp is the voltage amplitude of the frequency generator 10, and G(s) is the impedance transfer function of the demodulator and transducer combination.
The impedance transfer function for the particular circuit illus~rated is a function of a nu~ber of indepen-dent variables including but not limited to temperature, frequency, cnpacitance, and the linear response of the elements at different current levels. Each of these variables, as they change, will vary V(out~ for a constant Vp input according to their particular .: , ~Z322~
characteristics. These changes can be non-linear and either attenu~tions or amplifications. ldeQlly~ however, Vtout) for this combination should only change with the variable capacitance of the transducer and then linearly with the physical variable producing the change. The remaining variables will produce compensable errors in the output. The compensation network 16 will then vary the amplitude Vp, either by increasing or decreasing the level to substantially cancel changes in the V(out) amplitude not due to linear capacitance changes in the transducer.
In one preferred embodiment, as illustrated in ~igure 2, the compensation network 16 includes circuitry comprising a ratiometric compensation network 24 in combination with a temperature compensation network 26 for the quad-diode demodulator. Further included in this particular embodiment is a lineQr amplifier 22 connected to the output circuitry 14. A ratiometric offset circuit 20 provides a level ehange for the output circuitry 14 as will be more fully explained hereinafter.
From examination of equation (1) it is seen that the contribution of the diode drop Vd will cause the need for compensation in the demodulator circuit. As shown in Figure 10, the voltage drop Vd in the equation (l) is not only dependent upon temperature but is further a function of the current drawn through the device. Generally, as seen in the graphical representation~ larger temperatures will cause a decrease in the voltage characteristic and increasing currents will produce a larger voltage drop.
For a normal signal diode as normally used in such a d~modulator circuit such changes will cause a shifting voltage change between .1 - .15V in a voltage drop from a turn on point of approximately .8V to .75V at full current.
This shifting voltage drop of the diode, however, may be still fairly constant with respec$ to the ~Z3~25 substanti~lly larger chsnge in Vp cau~ed by an unregulated automotive power supply. This relatively constant drop of the diode can, therefore, cause a significant ratiometric error in the output ~oltage as it is not ratiometric itself. This non-ratiolmetry cannot be tolerated in certain environments where the analog output of the demodulator and transducer combination is changed to a digital number by a ratiometric conversion as illus-trated in Figure 9.
Normally, this technique envisions a slope generator that provides a constantly rising r~mp whose voltage is compared to the output voltage V(out) of the demodulator transducer combination. The counter begins a count when the ramp is started and when a comparison indicates the equivalency of the ramp voltage and the output voltage at 90 the counter is halted containing a digital represen-tation, N, of the analog voltage. For changes in the power supply Vin, the ramp is changed accordingly, and if the transducer output is ratiometric no error in the output count will appear. For higher voltages at point 92, the number N will still be obtained and at lower voltages at point 94 no error will be introduced.
However, if the output voltage V(out) remains constant or does not change ratiometrically, the error -~e or +Qe will result. The sensitivity of the transducer and demodulator combination will be effectively nullified by the ratiometric error introduced. It is known that in an automotiYe supply of approximately lOV a diode drop of .65V that is not ratiometric will introduce a percentage error in the order of 1-2%.
Figure 4 illustrates circuitry advantageously compensating the demdoulator and transducer combination for the temperature and current characteristics of the diodes in the bridge circuit and further eliminating the ratiometric error of the circuit. The compensation network circuit 16 comprises a voltage follower amplifier ~.23~ZS
ICl eonnected between a positive source of voltage, ~V, and ground. The positive source +V is obtained from regul~ting a ~ource Vin which can be for example ~n auto-motive power supply. The regulation circuit is provided by serially connecting Vin to 8 terminal of a load resistor Rl and thereafter the other terminal to the anode of a steering diode D which is connected at its cathode to the +V terminal. Shunt regulation is produced at the +V terminal by Zener diode Zl and filter capacitor Cl connected in parallel between the +V terminal and ground.
A ratiometric voltage Va is applied to the non-inverting input of the amplifier ICl via the junction of the serial combin~tion of a divider resistor R2 and a divider resistor R3 connected between the supply Vin and ground. For every change in the supply Vin, the divider point voltage Va will change in a ratiometric manner.
A compensation volta~e is developed at the inverting input to the amplifier ICl by the negative feedback of a serial pair of diodes D5, D6 connected between the emitter of a power transistor Tl and the inverting input.
A variable resistor R4 is also connected between the inverting input of the ~mplifier ICl and ground for varying the amount of current drawn throu~h the diode pair. Diodes D5, D6 should be matched to the quad-bridge diodes and R4 adjusted to pull equivalent current through the compensation diodes. The power transistor Tl which is controlled by its connection to the output of amplifier ICl at its base provides a controllable amount of current to a filter capacitor C3 attached between the emitter of the transistor and ground. The collector of the power transistor Tl is connected to the regulated positive supply +V.
In operation, the compensation network 16 produces a voltage Va at the non-inverting input to amplifier ICl which the circui$ will attempt to balance at the 1~3225 inverting input through the action of the power tranQistor Tl and the feedback through diodes DS, D6. An increase In voltage at point Va will increase the voltage at point Yb and the inverting input until it exceeds the voltage at point Va. The transistor Tl will then reduce the voltage at the inverting input until equivalence is reached. The filter capacitor C3 will delay the changes - and damp out any oscillations and produce filtering for the voltage at point Vb. Thus, the voltage at point Vb is two diode voltage drops, 2Vd, above the ratiometric voltQge at point ~a. Thus, Vb = va + 2Vd and if the peak to peak votlage, 2Vp, of the frequency generator 10 is set equivalent to Vb, then Vp = 1/2 Vb or V = 1/2 Va + Vd then substituting equation (4) into equation (1) Vdc Va Cp - Cl (S) p r where Va is ratiometric and the diode effects have been eliminated.
The voltage Vb is used as the power supply or peak to peak voltage of the frequency generator 10 by con-necting the emitter of transistor Tl to the power supply pins of inverters Il-I6 and grounding their reference supply pins. The output of inverter I2 is connected via feedback capacitor C4 and Yariable resistor RS to the input of the inverter Il in a conventional manner to form a free-running astable oscillator whose frequency is determined by the RC time constant of R5, C4. Further, feedback is provided by the connec$ion of a feedback resistor R6 to the input of inverter I2 and to the junction of the resistor R5 and capacitor C4. The 9 ~2~5 frequency should be adjusted such that the eircuit will not be sensitive to the frequency used. Preferably ~or the Harrison demodulQtor circuit this is in the range of 200-300 KHZ.
The output of the oscillator formed by inverters Il, I2 is transmitted from the output of the inverter I~ to the commonly connected inputs of the inverters I3-18.
The output of the invert2rs I3-16 are commonly ~onnected together to generate the output of the frequency generator 10. The inverters I3-I6 act QS buffer amplifiers for the output of the oscillator and do not permit the output voltage to change for differences in the demodulator load.
The ratiometric offset circuit 20 includes a voltage divider comprising the serial combination of a divider resistor R12 and a variable divider resistor Rll connected between the supply Vin and ground. A high frequency filter capacitor C10 is connected at the junction of the resistors to provide decoupling from any power supply transients or ~eedback. In operation, the resistor Rll is adjusted such that any capacitive transducer will produce a zero output for initial condi-tions o~ the physical operating psrameter th~t is to be meausred. The offset preserves the ratiometry developed in the eircuit by voltage Va.
Linear ~mplifications of the demodulator output terminal 15 can be accomplished by ~mplifier IC3 which is connected as a conventional non-inverting voltage amplifier. The power supply pins of IC3 are conneeted to the regulated source +V Qnd ground, and a frequency compensation capacitor C8 is provided normally as is known.
The non-invertinginput, receiving the single ended output of the demodulator via terminal 15, Emplifies the voltage by a variable gain and generates Y(ou$) over resistor R12 from the output of emplifier IC3. The ~L~L232ZS
varlable gain is provided by ad~usting a variable resistor R9 connected between the inverting input and ground in relatlon to a fixed resistor R10 connected between the output snd inverting input of the amplifier IC3. A tilter c~pacitor C9 is provided between ground and the output of amplifier IC3 to attenuate any high frequency noise.
In another pr~ferred embodiment, a quartz capacitive transducer is compenssted for linearization and tempera-ture errors by the compensation network 16 illustrated inthe block diagram of Figure 3. The demodul~tor circuit includes, as described above, a freguency generator 10, quad-diode bridge cirucit 12, and output circuitry 14.
Connected to the reference terminal of the output circuitry is a rQtiometric offset 20 as previously described with reference to Figure 2. A linear amplifier 22 can be used to provide gain for the output voltage of the output circuitry 14 and a 1QW pass filter 28 receives the output of the amplifier 22 to attenuate high fre9uency noise or spikes in the output voltage of the circuit, V(out).
The compensation network 16 includes in combination a transducer temperature compensation circuit 32 and a transducer linearization circuit 34 which provide a controllable voltage Vp to the frequency generator 10 to cancel the errors of the demodulator ~nd transducer combination. A feedbsck circuit 30 is provided to cycle a portion of the output voltage of amplifier 22 into the transducer linearization circuit 34 as a measure of the ¦
amount of linearization needed.
Detailed circuitry for the implementation of the block diagram of Figure 3 is illustrated in Figure 5.
The compensation network 16 is implemented in a similar fashion to the detailed circuit for blo~k lS in Figure 4 with the substitution of resistor R14 for diode D5 in the feedback loop between the emitter of transistor Tl and !
~lZ3~Z5 -18- ;~
, the inverting input of the ~mplifier ICl. Further, a variable resistor Rl3 comprlsing the leedback circuitry of block 30 is connectQd between the output o~ the amplifier IC3 and the inverting input ths amplifier ICl to complete the implementation of the~block 16 of this particular embodiment. Circuit blocks 10, 12, 14 and 20 of Fi~ure 5 comprise identical circuitry as that described in the similar numbered blocks of Figure 4 and will not be further described.
10Further, the block 22 comprising the circuitry for the linear amplifier is similar to that described in Figure 4 but with the addition of a decoupling resistor Rl5 connected between the positive supply +V and the power supply pin of the amplifier IC3. High frequency decoupling capacitors Cl3 and Cll have also been added to the circuit. The capacitor Cl3 is connected between ground and the positive power supply pin of amplifier IC3 while capacitor Cll is connected between the non- i inverting input to amplifier IC3 and ground. I
20The low pass filter 28 can be formëd in a conven- ! `
tional manner by the serial connection of a load resistor Rl2 and 8 capacitor Cl2 between the output of amplifier IC3 and ground. The output terminal voltage Y(out) is then generated from the ~unction formed at the resistor Rl2 and the capacitor Cl2.
Pigures 6, 7, and 8 illustrate in a preferred form for the quart2 capacitive transducer schematically shown ' in block 36 of Figure 5. The transducer 36, shown as Q
pressure transducer, comprises two opposing discs 60, 62 or plates of qusrtz or other vitreous material with similar characteristics. On the face of each disc, for exnmple, dis~ 60, there is formed two capacitor plates 66, 68 of some conductive metal by a screening or vapor deposition process or the like. After formation of the plates 66, 68, 70, 72 on the discs 60, 62 respectively, the discs are joined to form a gap between the plates by ~ .
23~2S
--l 9-- l an ~nnular frit 64 ~nd the interior of the transducer evacuated or set nt a reference pressure Pr.
As ~een in Fi~ure 8, a change in pressure P will cause a deformation of the discs 60, 62 and v~ry the gap distance between the plates of a pressure capacitor Cp which can be detected via the terminals 76J 80.
Normally, a reference CapACitOr, Cr, which c~n be detected via terminal 78, 82 does not change capacitance appreciably and can be used for reference compensation in the demodulator circuit ~or the capacitor Cp. A
capacitor of this type is more fully disclosed in a commonly assigned U.S. Patent 3,858,097 issued to Polye, the diselosure of which is hereby expressly incorpornted by re~erence.
The operation of the circuit illustrated in Figure 5 in combination with the quartz capacitive transducer will now be more fully explained. With respect to thermal sensitivity compensation, the compensation network 16 utilizes the temperature characteristic of the diode D6 to provide the needed correction. For the circuitry shown in Figure 5 and assuming for a moment that R13 - approaches infinity, b = Va + Vd rl4 16) where Vrl4 is the voltage across the resistor R14 and Vd is the diode voltage across D6 now, by OHM's law Vrl4 ~Vb - Vd)R14 (7~
or substituting equations (6) and (7) in equation (1) and reducing V(out) = A(f(x) f(p) ~ Vc) (8) where flx) = Va (1 ~ R14/R4) - Vd ~9) ~(p) = Cp - Cr (10) +__~
z~zs :
Vc = offset voltage, and A = linear amplifier gain If one now inspects equQtion (9) and its graph inFigure 12, the change in f(x) will be due only to temperature because of the -Yd term. For a diode the voltage drop decreases with increasing temperature and hence the subtraction of the Vd term will cause an increasing f(x) from temperature Tl to temperature T2 where T2 is greater than Tl.
Graph 11 illustrates the uncompensated f~p) due to thermally sensitivity of the quartz capacitive transducer. It is seen that the function f(p) has a lower slope for increasing temperatures where T2 is greater than Tl. From equation (8) it is seen that the lS function f(x) acts as a slope multiplier to the function f(p) but in the opposite direction and thus will increase the slope of the higher temperature curve at T2 in Figure 11. The result is illustrated in Figure 13 in which the curves are separated only by the thermal shift in their zero points. The correct multiplication constant can be obtained by adjusting the current drawn through diode D6 via variable resistor R4.
The linearity ccmpensation of the circuit of ~igure 5 will now be more ~ully described. Gensrally, the quartz capacitive transducer illustrated will not generate a linear change in output for a change in physical output variable. Figure 15 shows that the function f(p) based on the capacitanee ahange of the transducer will follow more of a square law with respect to changes in pressure thsn a linear function as repre-sented by the dotted line in the figure. Normally, such a response is caused by the change in pressure causing a relatively representative change in the spacing between the capacitor discs. ~oweYer, capacitance for a parallel plate con~iguration as shown changes inversely with the ~Z3225 .
square of the distance and not linearly. The compen-sation technique causes Vb to change such that V(out) will be compensated as illustrated in Fi~ure 14 and thus cancel the nonlinearity of the transducer. The resultant linear output is illustrated in Figure 16 where V(out) ~s graphically indicated as a first order function of the variable p after combining the compensation and non-linear response of the transducer.
This can be illustrated by writing the circuit transfer function as follows below. Recalling equation (8) and rewriting it for R13 not equal to infinity.
V~out) = A(fl(x) f(p) + Vc) (11) 1 - ~ ~R-14~------ ~ ~
where fl(x) = Va (1 + R14/R4 + R14/R13) - Vd (12 Then for fl(x) much greater than R14/R13, V(out) will be an increasing function that follows the numerator of equation (11) and f(p). The denominator, however, contains a linearizing term A~R14/R15) f(p) which tends to decrease V(out) for increases in f(p) and can be ad~usted controllably by varying R13. Thus, a simple `
adjustment may be made to linearize a quartz capacitive transducer that normally does not have a first order output function.
While preferred embodiments of the invention have been described, it will be obvious to those skilled in the art that various modifications and changes may be -made without departing form the spirit and scope o~ the invention QS iS set forth in the following claims.
WHAT 19 CLAIMED IS:
Generally, for capacitive transducers, the generation of the electrical signal is accomplished by the modulation of an alternating carrier frequency where one of the characteristics o the carrier is modified electrically by the vari~ble capacitance of the transducer. The carrier is thereafter detected or demod-ulated to obtain the intelligence contained and thereby generates a useful electrical signal representative of the sensed parameter. The circuit used for modulating the carrier is usually in integral combination with the demodulator cirsuit with the whole being termed herein- ;
af ter a transducer demodulator .
Particularly, such a ~ransducer demodulator circuit using a quad-diode configuration is shown in U.- S. Patents 3,318,153; 3,271, 669 issued to T. Lode. These circuits are particularly useful for capacitive pressure transducers. Another advantageous type of capacitive transducer demodulator 1.
~ ' .
1~L23Z25 that has been recently developed is the quad-diode bridge circuit. An example of which i~ illustrated in U.S.
Patents 3,883,812; 3,869,676 issued to Hnrrison et al.
The desirable characteristics o~ this demodulator include a sufficient magnitude o~ output which is relatively independent of excitation waveform a`nd requency. Addi-tionally, this transducer demodulator pro,vides excellent resolution for the change ln capacitance of the transducer and allows the transducer to be conveniently grounded. These are fehtures that will provide for greater use of this circuit in mul~i-farious transducer applications in the future. These circuits and others of their general type will herein further be termed quad-diode demodulators because of their circuitry utilizing four rectification devices or diodes.
If the capacitive transducer is a pressure transducer, a capa~itive transducer and demodulator combination, as deseribed above, can be utilized for sensing manifold absolute pressure (MAP~ changes in an internal combustion engine. The electrical signal obtained from the combination can then be used as is conventional to regulate functional aspects of the engine operation such as air/fuel ratio, timing, EGR, etc. when sensed with other engine parameters. In the automotive environment the convenient grounding of a capacitive transducer is an important feature which allows a direct conn~ction to the chassis and eliminates the problems of isolating a transducer with reference potential that is above or below chassis ground. Also, the referenced Harrison quad-diode circuit lends itself to remote transducer applications which probably will accompany many new developments in auto-motive elec~ronics.
However, there are still problems with using these quad-diode demodulators in harsh environments such as that found in the engine compartment of an automobile.
~23~ZS
The range o~ temperatures through whch the transducer-demodulator circuitry is subjected to i-9 extreme (-40F.
to 120 C.) ~nd the circuitry must, therefore, be provided with accurate temperature compensatlon. This problem Is complicated by the inclusion of the neeessary but non-linear diodes in the demodulator circuitry. These diodes will pro-duce not only different voltage drops for d~ffere~t temperatures but will also produce different voltage drops at the same temperature when different currents are conducted.
Another problem found in many environments but which is particularly troublesome in the automotive environment is the regulation of the power supply. With constantly changing demands on a limited battery and only a rough regulation from the voltage regulator for alternator voltage changes, surges and voltage drops of significant magnitude are not uncommon. Transducer electronios where the informatio~ is contained within the amplitude of transducer signal and changes with respect to a reference are particularly affected by these changes.
One method developed for overcoming this problem is ratiometry. This method contemplates that the output of a particular circuit will change in accordance with the ehanges in the power supply to always remain a predeter-mined percentage of the power supply for non-signal condit~ons. Thus, when a plurality of these circuits are connected together signal information will not be lost and errors will not be introduced because of the regu-lation problems of the power supply. Therefore, when operated in an automotive or other environments where regulation problems are prevalent, the quad-diode demod-ulator and capacitive transducer combination should be provided with ratiometric compensation for facile connec-tion to other system circuitry. Compensation for ratiometric errors is difficult because of the non-linear nature of the diodes of the demodulator which cause an error.
.
~Z3225 In certQin instances it is ~ust as import~nt to compensate for the transducer itself as it is to compensate the demodulstor circuitry. For ex~mple, quartz CQpaCitor transducers are relatively accurate and S inexpensive but they are temperature sensitive and some have linearity problems for reasonably priced transducers. It would be extremely advantageous to compensate a low cost quartz capaeitive transducer to provide a linear output without temperature dependency 10 while retaining the desirable features of a quad-diode demodulator.
SUMMARY OF THE INVENTION
The invention provides a compensation technique for a quad-diode demodulator snd cspacitive transducer combi-15 nation. The technique includes controllably varying the amplitude of an alterneting carrier frequency oppositely to changes produced in that amplitude by the compensable errors in the demodulator and transducer and thus effecting their cancellation.
The technique produces an extremely flexible compen-sation method where many different types of errors may bs compensated for without drastic modification of the demodulator circuitry. Moreover, compensation may be effected without disturbing the desirable characteristics of the quad-diode demodulator and capacitive transducer combination.
In a first preferred embodiment the technique is implemented by compensation network circuitry comprising means for ratiometric compensation and means for tempera-ture compensaeion of the quad-diode demodulator. A
voltage follower circuit is utilized to ~ompare and maintain the equivalency between a ratiometric voltage and the input voltage of a frequency generator minus a non-linear compensation voltage.
~L232;2 5 In the first embodiment the compensation voltage is generated as the complement of the non-linear non-ratiometric error attenuation introduced by the demodulator circuit because of the diodes that comprise the demodulation bridge. The in-put voltage to the frequency generator is therefore the ratio-metric voltage plus the non-linear non-ratiome~ric compensation voltage which compensates for the attenuation of the diodes in the demodulator. Since the diode voltage attenuation in the bridge is temperature dependent, the compensation voltage being the complement of the attenuation is also temperature dependent and produces a demodulator output that has the temperature term canc-eled. The compensation voltage is developed in this embodi-ment by drawing an exact amount of current through a temperature sensitive means. The temperature sensitive means have a temp-erature characteristic identical with the temperature character-istic of the demodulator diodes and of equivalent voltage mag-nitude. Thus, the voltage magnitude of the temperature sensi-tive means cancels the ratiometric error and changes with temperature to cancel the temperature dependency.
In a second preferred embodiment the technique is implemented by compensation network circuitry comprising means for temperature compensation and means for linearization of a quartz capacitive pressure transducer. A voltage follower cir-cuit is utilized to compare and maintain the equivalency between a ratiometric voltage and the input voltage of a frequency generator mlnus a non-linear compensatlon voltage.
In the second embodiment the compensation voltage is generated as the combination of a temperature dependent com-ponent and a non-linear component~ The temperature dependent component of the compensation voltage is used to compensate for the temperature dependency of the quartz capacitive transducer and the non-linear component for the non-linear response of the transducer.
The temperature dependent component in this embodi ment is developed by drawing an exact amount of current jk/ ~
~.Z3ZZS
~~rough a temperature sensitive means with a substantially linear temperature characteristic. The linear change in voltage with respect to temperature of the device is used as a slope multiplier to cancel the change in the output of the quartz capacitive transducer with respect to temperature. The non-linear component is generated in proportion to the inverted output voltage of the demodulator and transducer combination.
The non-linear component is then provided as negative feedback to reduce the input voltage to the frequency generator and cancel the non-linear increases in the quartz capacitive transducer.
In summary, therefore, the present invention may be seen as providing a capacitive transducer and demodulator com-bination circuit comprising: a capacitive transducer for measuring the variances of a physical parameter, the transducer adapted to transform the variances into changes of capacitance of the transducer; a frequency generator means for suppiying an alternating carrier voltage; a quad-diode demodulator electrically connected to receive fro~. the frequency generator means the alternating carrier voltage and further connected to the trans-ducer such that the changes in capacitance modulate the carrier, the demodulator detecting the carrier and generating an output voltage signal representative of the variances in the measured physical parameter; and compensation network means, electrically connected to the frequency generator, for varying the amplitude of the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodula-tor and transducer.
The present invention furthermore may be seen to provide a transducer and demodulator combination circuit com-prising: a transducer for measuring the variances of a physical parameter, the transducer adapted to transform the variances into changes of impedance of the transducer; a sd ~ -7-~L~Z3~25 Frequency generator means for supplying an alternating carrier voltage which is a ratiometric function of an input supply voltage; a deMoclulator electrically connec-ted to receive from the frequency generator means the alterna-ting carrier voltage and further connected to the transducer such tha1: -the chancJes in impedance modulate the carrier, the demodulator detecting the carrier and generating an output vol~age signal representa-tive of the variances in the measured physical parameter; and compensation network means for varying the amplitude o the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodulator and transducer wherein the compensation network means includes means for providing ratiometric compensation for the ratiometric error of the demodulator.
Thesè and other features, and aspects of the invention will be more fully understood and better appreciated from a reading of the following detailed disclosure taken in conjunction f with the appended drawings wherein:
BRIEF DESCRIPTION OF THE DRAWINGS
_ The prior art figure is a schematic view of a quad-diode demodulator circuit and capacitive transducer combination;
Figure l is a block diagram of a quad-diode demodulator circuit and capacitive transducer combination provided with a compensation network according to the invention;
Figure 2 is a detailed block diagram illustrating the quad-diode demodulator and capacitive transducer combination shown in Figure l, and provided with ratiometric and temperature compensation for the quad-diode demodulator; .:~
Figure 3, appearing on the same sheet as Figure 6, is a detailed block diagram illustrating the quad-diode demodula-tor and capacitive transducer combination shown in Figure l, and provided with temperature compensation and llnearization com--sd/~6 -8-ensation for a quartz capacitive transducer;
Figure 4 is a detailed schematic circuit of the block diagram illustrated in Figure 2;
Figure 5 is a detailed schematic circuit of the block diagram illustrated in Figure 3;
Figure 6 is a cross sectional view of c~ quartz capacitive pressure transducer taken along section line 6 in Figure 7;
Figure-7 is a cross sectional side view of a quartz capacitive pressure transducer in a quiescent state;
Figure 8 is a cross sectional side view of the ~-quartz capacitive pressure transducer in a state of flexure;
Figure 9 is an illustrative graphic diagram of a ratiometric conversion;
Figure 10 is an illustrative graphic diagram of temperature versus the voltage across a signal diode for different current levels; and . 5 sd/~ 8A-. .
- , ~ . . .
~Z3~ZS
g Figures 11 through 16 are graphic illustrations of functional relations at various point~ throughout the circuitry of ~igure 5.
The convention that like reference numeral refer to identical elements throughout the figures has been maintained for facilitating B clear description of the invention.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
Illustrated in the prior art figure is a quad-diode demodulator ior the detection of capaciti~e changes in a transducer caused by the variance of a physical par~meter. The demodulator comprises generally a frequency generator 10, a quad-diode bridge circuit 12, and an output circuit 14. The fre~uency generator 10 produces an alternating carrier frequency of an amplitude Vp to the junction of two coupling capacitors Ca and C~
of the bridge circuit 12. The other terminals of the coupling capacitors Ca and Cb are connected respectively between first and second diode bridge nodes ~ormed by diodes Dl, D3 and diodes D2, D4, and ground. The diodes ~Dl, D2, D3, and.D4 form.a quad-diode bridge and have parallel capacitors Cr, C connected between a third bridge node formed by the diodes Dl, D2, and ground, A
fourth bridge node formed by the diodes D3, D4, has a capacitor Cp connected between the node and ground.
Conventionally, Cp is representative of the variable capacitance of a capacitive transducer and Cr is a capacitance of a value that is equivalent to the zero point value of the capacitive trflnsducer. The capacitance Cr may be contained within the tr~nsducer for compensation or may be supplied ex~ernally. The capscitor Cs is provided as a small variable c~pacitor in parallel with Cr to enable exact zeroing o~ the initisl output of the capAcitive trarsducer. ~he total ~23~2s --, o-- .
.
capacit~nce Cs and Cr is substantially equivalent to Cr and this value will be used for further description when referring to these in combination.
Consider now one half of the circuit where values of C~7 Cb are much, much greater than Cp, Cr, Assuming Ca is charged, on transitions of the frequency generator 10, the diode Dl cond~cts and C charges to a voltage dependent upon the value of Cr si~ce C >~ Cr. Ca therefore discharges by an amount of charge dependent upon the v~lue of C .
Subsequently, on negatlve transitions, diode D3 conducts and the capacitor Cp charges an amount deter-mined by the value of Cp since C~>>CpCa charges by an equivaIent amount. If the capacitors Cp snd Cr are equal, then there is no net signal voltage left on Ca.
However~ if Cp changes as the result of sensing a change in a physical parameter, then a net DC voltage or charge will remain on capacitor CQ and appear as a signal output at point A. The signal voltage developed is a function of the difference of the values of capacitors Cp and Cr.
In a similar manner, diodes D2, D4 in cooperation with capacitors Cr and Cp will produce a net DC voltage at point B. This voltage will be equivalent to the voltage at point A but of opposite polarity. At a steady state condition, equal amounts of charge are transferred between Cp and Cr and $he voltage output from the demodulator will reduce to C - C
Vdc = p d C~~~~~~~ (1) p r where Vdc is the output volta~e measured across points AB, Vp is the voltage amplitude of the frequency generator, and Vd is the voltage drop across one of the diodes Dl - D4.
The output circuitry 14~ comprising an impedance Ra connected between point A and an output terminal 15 and an impedance Rb connected betwPen point B ~nd a ground terminal 17, provides a single ended output from the ~2~ZZ5 ~, differential voltages presented at points A and B
respectively. A capacitor Cd is provided by connecting one lead to the terminnl 15 and thè other lead to terminal 17 for coacting with the impednaces R~, Rb.
This demodulator circuit is more fully described in the two before referenced U.S. Patents issu0d to Harrison, the disclosure of which is expressly incorporated herein by reference.
With attention now directed to Figure 1, there is shown to advantage in block form a preferred quad-diode demodulator circuit including a frequency generator 10, quad-diode bridge circuit 12, output circuit 14, and capucitive transducer 18, the function and operation of which are identical to that described in the above prior art figure. In accordance with the invention, a compen-sation network 16 is connected to the frequency generator 10 for providing a controllable amplitude for the carrier frequency Vp to cancel a compensable error from the demodulator-transducer combination.
In theory, the output voltage of the demodulator and transducer combination can be written as a transfer function:
Vout/Vp = G~s) ~2) where VOUt is the output ~mplitude of the voltage at a5 terminals 15, 1~, Vp is the voltage amplitude of the frequency generator 10, and G(s) is the impedance transfer function of the demodulator and transducer combination.
The impedance transfer function for the particular circuit illus~rated is a function of a nu~ber of indepen-dent variables including but not limited to temperature, frequency, cnpacitance, and the linear response of the elements at different current levels. Each of these variables, as they change, will vary V(out~ for a constant Vp input according to their particular .: , ~Z322~
characteristics. These changes can be non-linear and either attenu~tions or amplifications. ldeQlly~ however, Vtout) for this combination should only change with the variable capacitance of the transducer and then linearly with the physical variable producing the change. The remaining variables will produce compensable errors in the output. The compensation network 16 will then vary the amplitude Vp, either by increasing or decreasing the level to substantially cancel changes in the V(out) amplitude not due to linear capacitance changes in the transducer.
In one preferred embodiment, as illustrated in ~igure 2, the compensation network 16 includes circuitry comprising a ratiometric compensation network 24 in combination with a temperature compensation network 26 for the quad-diode demodulator. Further included in this particular embodiment is a lineQr amplifier 22 connected to the output circuitry 14. A ratiometric offset circuit 20 provides a level ehange for the output circuitry 14 as will be more fully explained hereinafter.
From examination of equation (1) it is seen that the contribution of the diode drop Vd will cause the need for compensation in the demodulator circuit. As shown in Figure 10, the voltage drop Vd in the equation (l) is not only dependent upon temperature but is further a function of the current drawn through the device. Generally, as seen in the graphical representation~ larger temperatures will cause a decrease in the voltage characteristic and increasing currents will produce a larger voltage drop.
For a normal signal diode as normally used in such a d~modulator circuit such changes will cause a shifting voltage change between .1 - .15V in a voltage drop from a turn on point of approximately .8V to .75V at full current.
This shifting voltage drop of the diode, however, may be still fairly constant with respec$ to the ~Z3~25 substanti~lly larger chsnge in Vp cau~ed by an unregulated automotive power supply. This relatively constant drop of the diode can, therefore, cause a significant ratiometric error in the output ~oltage as it is not ratiometric itself. This non-ratiolmetry cannot be tolerated in certain environments where the analog output of the demodulator and transducer combination is changed to a digital number by a ratiometric conversion as illus-trated in Figure 9.
Normally, this technique envisions a slope generator that provides a constantly rising r~mp whose voltage is compared to the output voltage V(out) of the demodulator transducer combination. The counter begins a count when the ramp is started and when a comparison indicates the equivalency of the ramp voltage and the output voltage at 90 the counter is halted containing a digital represen-tation, N, of the analog voltage. For changes in the power supply Vin, the ramp is changed accordingly, and if the transducer output is ratiometric no error in the output count will appear. For higher voltages at point 92, the number N will still be obtained and at lower voltages at point 94 no error will be introduced.
However, if the output voltage V(out) remains constant or does not change ratiometrically, the error -~e or +Qe will result. The sensitivity of the transducer and demodulator combination will be effectively nullified by the ratiometric error introduced. It is known that in an automotiYe supply of approximately lOV a diode drop of .65V that is not ratiometric will introduce a percentage error in the order of 1-2%.
Figure 4 illustrates circuitry advantageously compensating the demdoulator and transducer combination for the temperature and current characteristics of the diodes in the bridge circuit and further eliminating the ratiometric error of the circuit. The compensation network circuit 16 comprises a voltage follower amplifier ~.23~ZS
ICl eonnected between a positive source of voltage, ~V, and ground. The positive source +V is obtained from regul~ting a ~ource Vin which can be for example ~n auto-motive power supply. The regulation circuit is provided by serially connecting Vin to 8 terminal of a load resistor Rl and thereafter the other terminal to the anode of a steering diode D which is connected at its cathode to the +V terminal. Shunt regulation is produced at the +V terminal by Zener diode Zl and filter capacitor Cl connected in parallel between the +V terminal and ground.
A ratiometric voltage Va is applied to the non-inverting input of the amplifier ICl via the junction of the serial combin~tion of a divider resistor R2 and a divider resistor R3 connected between the supply Vin and ground. For every change in the supply Vin, the divider point voltage Va will change in a ratiometric manner.
A compensation volta~e is developed at the inverting input to the amplifier ICl by the negative feedback of a serial pair of diodes D5, D6 connected between the emitter of a power transistor Tl and the inverting input.
A variable resistor R4 is also connected between the inverting input of the ~mplifier ICl and ground for varying the amount of current drawn throu~h the diode pair. Diodes D5, D6 should be matched to the quad-bridge diodes and R4 adjusted to pull equivalent current through the compensation diodes. The power transistor Tl which is controlled by its connection to the output of amplifier ICl at its base provides a controllable amount of current to a filter capacitor C3 attached between the emitter of the transistor and ground. The collector of the power transistor Tl is connected to the regulated positive supply +V.
In operation, the compensation network 16 produces a voltage Va at the non-inverting input to amplifier ICl which the circui$ will attempt to balance at the 1~3225 inverting input through the action of the power tranQistor Tl and the feedback through diodes DS, D6. An increase In voltage at point Va will increase the voltage at point Yb and the inverting input until it exceeds the voltage at point Va. The transistor Tl will then reduce the voltage at the inverting input until equivalence is reached. The filter capacitor C3 will delay the changes - and damp out any oscillations and produce filtering for the voltage at point Vb. Thus, the voltage at point Vb is two diode voltage drops, 2Vd, above the ratiometric voltQge at point ~a. Thus, Vb = va + 2Vd and if the peak to peak votlage, 2Vp, of the frequency generator 10 is set equivalent to Vb, then Vp = 1/2 Vb or V = 1/2 Va + Vd then substituting equation (4) into equation (1) Vdc Va Cp - Cl (S) p r where Va is ratiometric and the diode effects have been eliminated.
The voltage Vb is used as the power supply or peak to peak voltage of the frequency generator 10 by con-necting the emitter of transistor Tl to the power supply pins of inverters Il-I6 and grounding their reference supply pins. The output of inverter I2 is connected via feedback capacitor C4 and Yariable resistor RS to the input of the inverter Il in a conventional manner to form a free-running astable oscillator whose frequency is determined by the RC time constant of R5, C4. Further, feedback is provided by the connec$ion of a feedback resistor R6 to the input of inverter I2 and to the junction of the resistor R5 and capacitor C4. The 9 ~2~5 frequency should be adjusted such that the eircuit will not be sensitive to the frequency used. Preferably ~or the Harrison demodulQtor circuit this is in the range of 200-300 KHZ.
The output of the oscillator formed by inverters Il, I2 is transmitted from the output of the inverter I~ to the commonly connected inputs of the inverters I3-18.
The output of the invert2rs I3-16 are commonly ~onnected together to generate the output of the frequency generator 10. The inverters I3-I6 act QS buffer amplifiers for the output of the oscillator and do not permit the output voltage to change for differences in the demodulator load.
The ratiometric offset circuit 20 includes a voltage divider comprising the serial combination of a divider resistor R12 and a variable divider resistor Rll connected between the supply Vin and ground. A high frequency filter capacitor C10 is connected at the junction of the resistors to provide decoupling from any power supply transients or ~eedback. In operation, the resistor Rll is adjusted such that any capacitive transducer will produce a zero output for initial condi-tions o~ the physical operating psrameter th~t is to be meausred. The offset preserves the ratiometry developed in the eircuit by voltage Va.
Linear ~mplifications of the demodulator output terminal 15 can be accomplished by ~mplifier IC3 which is connected as a conventional non-inverting voltage amplifier. The power supply pins of IC3 are conneeted to the regulated source +V Qnd ground, and a frequency compensation capacitor C8 is provided normally as is known.
The non-invertinginput, receiving the single ended output of the demodulator via terminal 15, Emplifies the voltage by a variable gain and generates Y(ou$) over resistor R12 from the output of emplifier IC3. The ~L~L232ZS
varlable gain is provided by ad~usting a variable resistor R9 connected between the inverting input and ground in relatlon to a fixed resistor R10 connected between the output snd inverting input of the amplifier IC3. A tilter c~pacitor C9 is provided between ground and the output of amplifier IC3 to attenuate any high frequency noise.
In another pr~ferred embodiment, a quartz capacitive transducer is compenssted for linearization and tempera-ture errors by the compensation network 16 illustrated inthe block diagram of Figure 3. The demodul~tor circuit includes, as described above, a freguency generator 10, quad-diode bridge cirucit 12, and output circuitry 14.
Connected to the reference terminal of the output circuitry is a rQtiometric offset 20 as previously described with reference to Figure 2. A linear amplifier 22 can be used to provide gain for the output voltage of the output circuitry 14 and a 1QW pass filter 28 receives the output of the amplifier 22 to attenuate high fre9uency noise or spikes in the output voltage of the circuit, V(out).
The compensation network 16 includes in combination a transducer temperature compensation circuit 32 and a transducer linearization circuit 34 which provide a controllable voltage Vp to the frequency generator 10 to cancel the errors of the demodulator ~nd transducer combination. A feedbsck circuit 30 is provided to cycle a portion of the output voltage of amplifier 22 into the transducer linearization circuit 34 as a measure of the ¦
amount of linearization needed.
Detailed circuitry for the implementation of the block diagram of Figure 3 is illustrated in Figure 5.
The compensation network 16 is implemented in a similar fashion to the detailed circuit for blo~k lS in Figure 4 with the substitution of resistor R14 for diode D5 in the feedback loop between the emitter of transistor Tl and !
~lZ3~Z5 -18- ;~
, the inverting input of the ~mplifier ICl. Further, a variable resistor Rl3 comprlsing the leedback circuitry of block 30 is connectQd between the output o~ the amplifier IC3 and the inverting input ths amplifier ICl to complete the implementation of the~block 16 of this particular embodiment. Circuit blocks 10, 12, 14 and 20 of Fi~ure 5 comprise identical circuitry as that described in the similar numbered blocks of Figure 4 and will not be further described.
10Further, the block 22 comprising the circuitry for the linear amplifier is similar to that described in Figure 4 but with the addition of a decoupling resistor Rl5 connected between the positive supply +V and the power supply pin of the amplifier IC3. High frequency decoupling capacitors Cl3 and Cll have also been added to the circuit. The capacitor Cl3 is connected between ground and the positive power supply pin of amplifier IC3 while capacitor Cll is connected between the non- i inverting input to amplifier IC3 and ground. I
20The low pass filter 28 can be formëd in a conven- ! `
tional manner by the serial connection of a load resistor Rl2 and 8 capacitor Cl2 between the output of amplifier IC3 and ground. The output terminal voltage Y(out) is then generated from the ~unction formed at the resistor Rl2 and the capacitor Cl2.
Pigures 6, 7, and 8 illustrate in a preferred form for the quart2 capacitive transducer schematically shown ' in block 36 of Figure 5. The transducer 36, shown as Q
pressure transducer, comprises two opposing discs 60, 62 or plates of qusrtz or other vitreous material with similar characteristics. On the face of each disc, for exnmple, dis~ 60, there is formed two capacitor plates 66, 68 of some conductive metal by a screening or vapor deposition process or the like. After formation of the plates 66, 68, 70, 72 on the discs 60, 62 respectively, the discs are joined to form a gap between the plates by ~ .
23~2S
--l 9-- l an ~nnular frit 64 ~nd the interior of the transducer evacuated or set nt a reference pressure Pr.
As ~een in Fi~ure 8, a change in pressure P will cause a deformation of the discs 60, 62 and v~ry the gap distance between the plates of a pressure capacitor Cp which can be detected via the terminals 76J 80.
Normally, a reference CapACitOr, Cr, which c~n be detected via terminal 78, 82 does not change capacitance appreciably and can be used for reference compensation in the demodulator circuit ~or the capacitor Cp. A
capacitor of this type is more fully disclosed in a commonly assigned U.S. Patent 3,858,097 issued to Polye, the diselosure of which is hereby expressly incorpornted by re~erence.
The operation of the circuit illustrated in Figure 5 in combination with the quartz capacitive transducer will now be more fully explained. With respect to thermal sensitivity compensation, the compensation network 16 utilizes the temperature characteristic of the diode D6 to provide the needed correction. For the circuitry shown in Figure 5 and assuming for a moment that R13 - approaches infinity, b = Va + Vd rl4 16) where Vrl4 is the voltage across the resistor R14 and Vd is the diode voltage across D6 now, by OHM's law Vrl4 ~Vb - Vd)R14 (7~
or substituting equations (6) and (7) in equation (1) and reducing V(out) = A(f(x) f(p) ~ Vc) (8) where flx) = Va (1 ~ R14/R4) - Vd ~9) ~(p) = Cp - Cr (10) +__~
z~zs :
Vc = offset voltage, and A = linear amplifier gain If one now inspects equQtion (9) and its graph inFigure 12, the change in f(x) will be due only to temperature because of the -Yd term. For a diode the voltage drop decreases with increasing temperature and hence the subtraction of the Vd term will cause an increasing f(x) from temperature Tl to temperature T2 where T2 is greater than Tl.
Graph 11 illustrates the uncompensated f~p) due to thermally sensitivity of the quartz capacitive transducer. It is seen that the function f(p) has a lower slope for increasing temperatures where T2 is greater than Tl. From equation (8) it is seen that the lS function f(x) acts as a slope multiplier to the function f(p) but in the opposite direction and thus will increase the slope of the higher temperature curve at T2 in Figure 11. The result is illustrated in Figure 13 in which the curves are separated only by the thermal shift in their zero points. The correct multiplication constant can be obtained by adjusting the current drawn through diode D6 via variable resistor R4.
The linearity ccmpensation of the circuit of ~igure 5 will now be more ~ully described. Gensrally, the quartz capacitive transducer illustrated will not generate a linear change in output for a change in physical output variable. Figure 15 shows that the function f(p) based on the capacitanee ahange of the transducer will follow more of a square law with respect to changes in pressure thsn a linear function as repre-sented by the dotted line in the figure. Normally, such a response is caused by the change in pressure causing a relatively representative change in the spacing between the capacitor discs. ~oweYer, capacitance for a parallel plate con~iguration as shown changes inversely with the ~Z3225 .
square of the distance and not linearly. The compen-sation technique causes Vb to change such that V(out) will be compensated as illustrated in Fi~ure 14 and thus cancel the nonlinearity of the transducer. The resultant linear output is illustrated in Figure 16 where V(out) ~s graphically indicated as a first order function of the variable p after combining the compensation and non-linear response of the transducer.
This can be illustrated by writing the circuit transfer function as follows below. Recalling equation (8) and rewriting it for R13 not equal to infinity.
V~out) = A(fl(x) f(p) + Vc) (11) 1 - ~ ~R-14~------ ~ ~
where fl(x) = Va (1 + R14/R4 + R14/R13) - Vd (12 Then for fl(x) much greater than R14/R13, V(out) will be an increasing function that follows the numerator of equation (11) and f(p). The denominator, however, contains a linearizing term A~R14/R15) f(p) which tends to decrease V(out) for increases in f(p) and can be ad~usted controllably by varying R13. Thus, a simple `
adjustment may be made to linearize a quartz capacitive transducer that normally does not have a first order output function.
While preferred embodiments of the invention have been described, it will be obvious to those skilled in the art that various modifications and changes may be -made without departing form the spirit and scope o~ the invention QS iS set forth in the following claims.
WHAT 19 CLAIMED IS:
Claims (21)
1. A capacitive transducer and demodulator combi-nation circuit comprising:
a capacitive transducer for measuring the variances of a physical parameter, said transducer adapted to transform the variances into changes of capacitance of the transducer;
8 frequency generator means for supplying an alternating carrier voltage;
a quad-diode demodulator electrically connected to receive from said frequency generator means the alternating carrier voltage and further connected to said transducer such that the changes in capacitance modulate said carrier, said demodulator detecting said carrier and generating an output voltage signal represen-tative of the variances in the measured physical parameter; and compensation network means, electrically con-nected to said frequency generator, for varying the amplitude of the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodulator and transducer.
a capacitive transducer for measuring the variances of a physical parameter, said transducer adapted to transform the variances into changes of capacitance of the transducer;
8 frequency generator means for supplying an alternating carrier voltage;
a quad-diode demodulator electrically connected to receive from said frequency generator means the alternating carrier voltage and further connected to said transducer such that the changes in capacitance modulate said carrier, said demodulator detecting said carrier and generating an output voltage signal represen-tative of the variances in the measured physical parameter; and compensation network means, electrically con-nected to said frequency generator, for varying the amplitude of the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodulator and transducer.
2. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 1 wherein said compen-sation network means includes:
means for providing ratiometric compensation for the ratiometric error of said quad-diode demodulator.
means for providing ratiometric compensation for the ratiometric error of said quad-diode demodulator.
3. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 2 wherein said compen-sation network means includes:
means for providing temperature compensation for the temperature error of said quad-diode demodulator.
means for providing temperature compensation for the temperature error of said quad-diode demodulator.
4. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 3 which further includes:
ratiometric offset means for adjusting the zero point of said capacitive transducer with a ratiometric voltage.
ratiometric offset means for adjusting the zero point of said capacitive transducer with a ratiometric voltage.
5. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 4 which further includes:
amplification means for the linear amplifi-cation of the output voltage signal of the demodulator.
amplification means for the linear amplifi-cation of the output voltage signal of the demodulator.
6. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 1 wherein said compen-sation network means includes:
voltage follower means for generating the power supply voltage of said frequency generator means where said power supply voltage is generated as the algebraic combination of an input supply voltage and a compensation voltage, said power supply voltage increasing or decreasing in response to changes in said compensation voltage.
voltage follower means for generating the power supply voltage of said frequency generator means where said power supply voltage is generated as the algebraic combination of an input supply voltage and a compensation voltage, said power supply voltage increasing or decreasing in response to changes in said compensation voltage.
7. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 6 wherein said voltage follower means includes:
a differential amplifier with an inverting input and non-inverting input further having an output terminal connected to the control terminal of a variable impedance device, said impedance device connected between a reference supply voltage and a power supply node of said frequency generator; said amplifier having the input supply voltage connected to said non-inverting input and the variable compensation voltage developed by feedback circuitry connected between the power supply node and the inverting input of said amplifier;
said impedance device controlled by the output of said differential amplifier to change the voltage at the power supply node in response to the amplifier balancing the non-inverting and inverting inputs.
a differential amplifier with an inverting input and non-inverting input further having an output terminal connected to the control terminal of a variable impedance device, said impedance device connected between a reference supply voltage and a power supply node of said frequency generator; said amplifier having the input supply voltage connected to said non-inverting input and the variable compensation voltage developed by feedback circuitry connected between the power supply node and the inverting input of said amplifier;
said impedance device controlled by the output of said differential amplifier to change the voltage at the power supply node in response to the amplifier balancing the non-inverting and inverting inputs.
8. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 7 wherein said feed-back circuitry includes:
a pair of first and second signal diodes with characteristic response curves substantially identical to the diodes of said demodulator; the anode of the first diode connected to the power supply node and the cathode of the first diode connected to the anode of the second diode which has its cathode connected to the inverting input of said amplifier; and a variable resistance connected between the inverting input and ground for adjusting the current flow through the diodes.
a pair of first and second signal diodes with characteristic response curves substantially identical to the diodes of said demodulator; the anode of the first diode connected to the power supply node and the cathode of the first diode connected to the anode of the second diode which has its cathode connected to the inverting input of said amplifier; and a variable resistance connected between the inverting input and ground for adjusting the current flow through the diodes.
9. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 8 wherein:
said input supply voltage is developed as a ratiometric voltage in reference to a relatively unregulated supply voltage.
said input supply voltage is developed as a ratiometric voltage in reference to a relatively unregulated supply voltage.
10. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 1 wherein said capacitive transducer is a quartz capacitive pressure transducer.
11. A capacitive transducer and demodulator combi-nation circuit defined in Claim 10 wherein said compen-sation network includes:
means for providing temperature compensation for the temperature error of said quartz capacitive transducer.
means for providing temperature compensation for the temperature error of said quartz capacitive transducer.
12. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 11 wherein said compensation network includes:
means for providing linearization compensation for the linearity error of said quartz capacitive transducer.
means for providing linearization compensation for the linearity error of said quartz capacitive transducer.
13. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 12 which includes:
ratiometric offset means for adjusting the zero point of said quartz capacitive transducer with a ratiometric voltage.
ratiometric offset means for adjusting the zero point of said quartz capacitive transducer with a ratiometric voltage.
14. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 13 which includes:
amplification means for the linear amplifi-cation of the output voltage signal of the demodulator.
amplification means for the linear amplifi-cation of the output voltage signal of the demodulator.
15. A capacitive transducer and demodulator combi-nation circuit as defined in Claim 14 which includes:
filter means electrically connected to the output of said amplification means for providing high frequency and noise attenuation to the amplified output voltage of said demodulator.
filter means electrically connected to the output of said amplification means for providing high frequency and noise attenuation to the amplified output voltage of said demodulator.
16. A capacitive transducer and demodulator combi-nation circuit comprising:
a capacitive transducer for measuring the variances of a physical parameter and for transforming the variances into changes in capacitance of the transducer, wherein said trans-ducer has a variable capacitor Cp which changes capacitance with respect to the physical variable and a reference capaci-tor Cr which is substantially unchanging with respect to the physical parameter;
a quad-diode demodulator electrically connected to the capacitive transducer and electrically connected to a frequency generator means for supplying an alternating carrier voltage such that the carrier voltage is modulated and detected to generate an output voltage signal representative of the variances in the measured physical parameter, said demodulator having a voltage transfer function of:
where Vdc is said output voltage signal, Vp is the voltage amplitude of said alternating carrier voltage, and Vd is the voltage drop across one of the diodes of said demodulator; and compensation network means, electrically connected to said frequency generator, for varying the amplitude of the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodu-lator and transducer.
a capacitive transducer for measuring the variances of a physical parameter and for transforming the variances into changes in capacitance of the transducer, wherein said trans-ducer has a variable capacitor Cp which changes capacitance with respect to the physical variable and a reference capaci-tor Cr which is substantially unchanging with respect to the physical parameter;
a quad-diode demodulator electrically connected to the capacitive transducer and electrically connected to a frequency generator means for supplying an alternating carrier voltage such that the carrier voltage is modulated and detected to generate an output voltage signal representative of the variances in the measured physical parameter, said demodulator having a voltage transfer function of:
where Vdc is said output voltage signal, Vp is the voltage amplitude of said alternating carrier voltage, and Vd is the voltage drop across one of the diodes of said demodulator; and compensation network means, electrically connected to said frequency generator, for varying the amplitude of the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodu-lator and transducer.
17. A capacitive transducer and demodulator combination circuit as set forth in Claim 16 wherein:
said compensation network means is connected to the power supply of said frequency generator means and provides a supply voltage Vb which is given by the function:
Vb = Va + 2Vd where Vb is the peak-to-peak amplitude of the frequency generator carrier voltage and, equal to 2Vp, Va is the voltage of a reference supply, and Vd is equivalent to the voltage drop across one of the diodes of said demodulator; said term 2Vd cancelling an identical term in the voltage transfer function of the demodulator to compensate for the varying character-istics with respect to temperature of the diodes of said demodulator.
said compensation network means is connected to the power supply of said frequency generator means and provides a supply voltage Vb which is given by the function:
Vb = Va + 2Vd where Vb is the peak-to-peak amplitude of the frequency generator carrier voltage and, equal to 2Vp, Va is the voltage of a reference supply, and Vd is equivalent to the voltage drop across one of the diodes of said demodulator; said term 2Vd cancelling an identical term in the voltage transfer function of the demodulator to compensate for the varying character-istics with respect to temperature of the diodes of said demodulator.
18. A capacitive transducer and demodulator com-bination circuit as set forth in Claim 17 wherein:
said reference supply voltage Va is a ratiometric function of an input supply voltage Vin and said term 2Vd cancels an identical term in the voltage transfer function of the demodulator to compensate for the ratiometric error of the combination circuit caused by the diodes of said demodulator.
said reference supply voltage Va is a ratiometric function of an input supply voltage Vin and said term 2Vd cancels an identical term in the voltage transfer function of the demodulator to compensate for the ratiometric error of the combination circuit caused by the diodes of said demodulator.
19. A capacitive transducer and demodulator combi-nation circuit as set forth in Claim 18 further including:
ratiometric offset means for adjusting the zero point of said capacitive transducer with a ratiometric voltage which is a function of Vin.
ratiometric offset means for adjusting the zero point of said capacitive transducer with a ratiometric voltage which is a function of Vin.
20. A transducer and demodulator combination cir-cuit comprising:
a transducer for measuring the variances of a physical parameter, said transducer adapted to transform the variances into changes of impendance of the transducer;
a frequency generator means for supplying an alter-nating carrier voltage which is a ratiometric function of an input supply voltage;
a demodulator electrically connected to receive from said frequency generator means the alternating carrier voltage and further connected to said transducer such that the changes in impedance modulate said carrier, said demodulator detecting said carrier and generating an output voltage signal representa-tive of the variances in the measured physical parameter; and compensation network means for varying the amplitude of the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodulator and transducer wherein said compensation network means includes means for providing ratiometric compensation for the ratiometric error of said demodulator.
a transducer for measuring the variances of a physical parameter, said transducer adapted to transform the variances into changes of impendance of the transducer;
a frequency generator means for supplying an alter-nating carrier voltage which is a ratiometric function of an input supply voltage;
a demodulator electrically connected to receive from said frequency generator means the alternating carrier voltage and further connected to said transducer such that the changes in impedance modulate said carrier, said demodulator detecting said carrier and generating an output voltage signal representa-tive of the variances in the measured physical parameter; and compensation network means for varying the amplitude of the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodulator and transducer wherein said compensation network means includes means for providing ratiometric compensation for the ratiometric error of said demodulator.
21. A transducer and demodulator combination circuit comprising:
a transducer for measuring the variances of a physical parameter, said transducer adapted to transform the variances into changes of impedance of the transducer;
a frequency generator means for supplying an alter-nating carrier voltage;
a quad-diode demodulator having a voltage transfer function relatively independent of carrier waveform and fre-quency electrically connected to receive from said frequency generator means the alternating carrier voltage and further connected to said transducer such that the changes in im-pedance modulate said carrier, said demodulator detecting said carrier and generating an output voltage signal represent-ative of the variances in the measured physical parameter;
and compensation network means, electrically connected to said frequency generator, for varying the amplitude of the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodu-lator and transducer.
a transducer for measuring the variances of a physical parameter, said transducer adapted to transform the variances into changes of impedance of the transducer;
a frequency generator means for supplying an alter-nating carrier voltage;
a quad-diode demodulator having a voltage transfer function relatively independent of carrier waveform and fre-quency electrically connected to receive from said frequency generator means the alternating carrier voltage and further connected to said transducer such that the changes in im-pedance modulate said carrier, said demodulator detecting said carrier and generating an output voltage signal represent-ative of the variances in the measured physical parameter;
and compensation network means, electrically connected to said frequency generator, for varying the amplitude of the carrier voltage oppositely to the amplitude changes in the output signal produced by the compensable errors of the demodu-lator and transducer.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US87805678A | 1978-02-15 | 1978-02-15 | |
| US878,056 | 1978-02-15 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1123225A true CA1123225A (en) | 1982-05-11 |
Family
ID=25371286
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA319,755A Expired CA1123225A (en) | 1978-02-15 | 1979-01-16 | Compensated capacitive transducer demodulator circuit |
Country Status (8)
| Country | Link |
|---|---|
| JP (2) | JPS54115175A (en) |
| AU (1) | AU521105B2 (en) |
| CA (1) | CA1123225A (en) |
| DE (1) | DE2905463A1 (en) |
| ES (1) | ES477752A1 (en) |
| FR (1) | FR2417887B1 (en) |
| GB (1) | GB2015162B (en) |
| IT (1) | IT1111910B (en) |
Families Citing this family (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US4295376A (en) * | 1978-12-01 | 1981-10-20 | Besco Industries, Inc. | Force responsive transducer |
| US4288835A (en) * | 1979-04-16 | 1981-09-08 | The Bendix Corporation | Pressure sensor |
| US4322977A (en) * | 1980-05-27 | 1982-04-06 | The Bendix Corporation | Pressure measuring system |
| US4509007A (en) * | 1982-09-30 | 1985-04-02 | Ibm Corporation | Differential sensor measuring apparatus and method including sensor compensator circuitry |
| FR2632069B1 (en) * | 1988-05-30 | 1990-12-14 | Garcia Manuel | IMPEDANCE VARIATION SPEED DETECTION SENSOR OR IMPEDANCE VARIATION SPEED |
Family Cites Families (10)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US3318153A (en) * | 1962-12-04 | 1967-05-09 | Rosemount Eng Co Ltd | Diode loop capacitor comparative circuit including a pair of transformer windings coupled in phase |
| US3271669A (en) * | 1962-12-04 | 1966-09-06 | Rosemount Eng Co Ltd | Alternating current diode loop capacitance measurement circuits |
| GB1110943A (en) * | 1963-10-01 | 1968-04-24 | Sangamo Weston | Improvements in or relating to electrical bridge networks |
| US3646538A (en) * | 1969-10-27 | 1972-02-29 | Rosemount Eng Co Ltd | Transducer circuitry for converting a capacitance signal to a dc current signal |
| US3648165A (en) * | 1970-09-24 | 1972-03-07 | Sun Oil Co | Capacitance-measuring apparatus including means maintaining the voltage across the unknown capacitance constant |
| US3883812A (en) * | 1971-12-20 | 1975-05-13 | Nasa | Diode-quad bridge circuit means |
| US3869672A (en) * | 1972-05-13 | 1975-03-04 | Int Standard Electric Corp | Method and arrangements for the digital control of operating functions, radio and television receivers |
| DE2314754C2 (en) * | 1973-03-24 | 1979-02-08 | Hottinger Baldwin Messtechnik Gmbh, 6100 Darmstadt | Electrical multi-point measuring device |
| JPS5144662A (en) * | 1974-10-08 | 1976-04-16 | Masakichi Kawahara | |
| JPS5252676A (en) * | 1975-10-24 | 1977-04-27 | Yokogawa Hokushin Electric Corp | Capacity-to-electricity converter |
-
1979
- 1979-01-16 CA CA319,755A patent/CA1123225A/en not_active Expired
- 1979-01-25 AU AU43655/79A patent/AU521105B2/en not_active Ceased
- 1979-01-26 GB GB7902832A patent/GB2015162B/en not_active Expired
- 1979-02-13 DE DE19792905463 patent/DE2905463A1/en active Granted
- 1979-02-14 IT IT20175/79A patent/IT1111910B/en active
- 1979-02-15 JP JP1555279A patent/JPS54115175A/en active Pending
- 1979-02-15 FR FR7903843A patent/FR2417887B1/en not_active Expired
- 1979-02-15 ES ES477752A patent/ES477752A1/en not_active Expired
-
1987
- 1987-11-12 JP JP1987172089U patent/JPS6344731Y2/ja not_active Expired
Also Published As
| Publication number | Publication date |
|---|---|
| GB2015162A (en) | 1979-09-05 |
| GB2015162B (en) | 1982-08-18 |
| IT7920175A0 (en) | 1979-02-14 |
| DE2905463A1 (en) | 1979-08-16 |
| FR2417887A1 (en) | 1979-09-14 |
| JPS6344731Y2 (en) | 1988-11-21 |
| IT1111910B (en) | 1986-01-13 |
| FR2417887B1 (en) | 1985-06-07 |
| AU4365579A (en) | 1979-08-23 |
| AU521105B2 (en) | 1982-03-18 |
| ES477752A1 (en) | 1979-10-16 |
| DE2905463C2 (en) | 1989-12-28 |
| JPS6396415U (en) | 1988-06-22 |
| JPS54115175A (en) | 1979-09-07 |
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