CA1072698A - Electronic telephone network - Google Patents
Electronic telephone networkInfo
- Publication number
- CA1072698A CA1072698A CA324,419A CA324419A CA1072698A CA 1072698 A CA1072698 A CA 1072698A CA 324419 A CA324419 A CA 324419A CA 1072698 A CA1072698 A CA 1072698A
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- amplifier
- telephone
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Abstract
Abstract of the Disclosure An electronic telephone network suitable for use with two-wire telephone line includes a low-output dynamic microphone coupled to a preamplifier. The output signal of the preamplifier is coupled through a first equalization network which is responsive to an equalization signal which is related to the DC line current to equalize the frequency and amplitude spectrum of the transmitted signal irrespective of the telephone line length or loss. The output of the equalization network is applied to a line-driver amplifier, and as a first input to an electronic signal separator or hybrid. The output of the line-driver is coupled to the two-wire line, and as a second input through the separator by way of an attenuator. The output of the separator is coupled to a receive amplifier by way of a second equalization network which is also responsive to the equalization signal to equalize the frequency spectrum of the receive signal as a function of line length.
The loss of the attenuator is selected to be equal to the gain of the line-driver thereby to isolate the transmitted signal and to pass the receive signal to the receive amplifier. In a preferred embodiment, the equalization signal is the DC supply voltage derived from the telephone line itself. The output impedance of the line-driver is lowered during transmit to reduce the sensitivity of the separator isolation and, therefore, the sensitivity of the sidetone signal. The various circuits of the network utilize active loads and other circuitry to provide maximum transmit signal dynamic range even at low line terminal voltages.
The loss of the attenuator is selected to be equal to the gain of the line-driver thereby to isolate the transmitted signal and to pass the receive signal to the receive amplifier. In a preferred embodiment, the equalization signal is the DC supply voltage derived from the telephone line itself. The output impedance of the line-driver is lowered during transmit to reduce the sensitivity of the separator isolation and, therefore, the sensitivity of the sidetone signal. The various circuits of the network utilize active loads and other circuitry to provide maximum transmit signal dynamic range even at low line terminal voltages.
Description
~7Z6~3 Background of the_Invention This invention relates to electronic telephone networks and, more particularly, to a network which adapts itself to a wide range of telephone line lengths, losses, and operating conditions.
Conventional telephone networks include a carbon mi.crophone, a transformer hybrid, a dynamic receiver, and various components such as resistors, capacitors, etc., and are used to interface with a standard two-wire telephone line with the telephone handset. Due to the passive nature of the primary components (e.g., the carbon microphone~ ~he standard telephone networks have several disadvantages such as large physical size, lack of fidelity, compatibility with other electronic telephone devices, and versatility. Elec~ronic telephone networks are also known in the art and obviate many of the above enumerated disadvantages while providing transmit and receive gain and power output sufficient to enable the use of low-eficiency dynamic transducers in the handsets.
However, these electronic telephone networks have several known dis-advantages in that they are expensive, fragile and, more importantly, incompatable with many existing telephone facilities or practices. For example, the incompatibility pro~lem arises due to the relatlvely high DC voltage requirement of the electronic networks as they derive their power from the telephone line. Disadvantageously, this prevents or severclylimits parallel operation with conventional telephone networks.
~urther~ore, ~he high DC resistance of these prîor art networks limits the maximum telephone line loop lengths on exchanges with conventional supervision equipment which require certain minimum current drain levels for satisfactory operation.
These and other disadvan~ages are overcome by the present , ' . - .... . . . :-. . . , . ,. - , , .
~7~
inven~ion wherein an electronic telephone network is provided which provides the normal functions of an electronic telephone network bu~ which further provides: an electronic hybrid to separate the receive signal from the transmit signal while eliminating objectionable sidetone; correct DC line current drain levels for ~elephone central office supervision equipment; relatively cons*ant re~eive and transmit lev~ls over the entire range of conventional telephone line distances between the network and the exchanges; efficient use of available DC power; independence of the AC
impedance characteristics of the telephone line; and, more particularly, while providing operation for DC terminal vol~age between the normal maximum of 8.0 volts down to, and including, 2.2 volts.
Summary of ~he Invention Briefly, an electronic network for receiving and transmitting ; telephone signals over the two-wire line is provided. The network includes a first amplifier and means for applying the transmit portion of ~he telephone signals to the first amplifier. First means are provided which are responsive to the first amplifier for applying the amplified signal output of the amplifier to the line. Second means are provided which are ~ responsive to the first amplifier and the output of the firs~ means for 20 separating the received portion of the telephone signals from the composite telephone signals at an output thereof. The network includes a second amplifier coupled to the ou~put of the second means for increasing the level of the separated received signals; and, third means for deriving operating potential for the network from the telephone lines are also provided. Fourth means couple the first amplifier to the first means and are responsive to the operating potential for altering ~he frequency/amplitude characteristics of the amplified signal output _ z _ : ~ ,.. - ' ' . . .
-:
~0~72~
of the first amplifier in accordance with the level of the operating potential. The preferred circuitry embodiments include means for providing low line terminal voltage operat.ion, and, control of the output impedance of as associated line-driver amplifier to reduce the electronic ~ -hybrid sensitivity and, therefore, the sensitivity of the sidetone signal to the line impedance characteristics.
According to the invention there is provided an amplifier adapted for coupling telephone signals to a ~elephone line and for controlling the DC current flowing through said telephone line to a predetermined value at a given telephone line loop resistance, said amplifier comprising;
at least one output transistor having firs~ and second main electrodes coupled across said tele~hone line, and having a control electrode; means or applying said telephone signals to said control electrode; means coupled between said control electrode and one of said main electrodes for deriving a firs~ signal related to the DC current flowing through said main electrodes; means coupled across said telephone line for deriving a second signal related to the DC voltage appearing across said ~elephone line;
means for comparing said first and second signals to provide a third signal indicative of the difference betw~en s~id first and second signals;
and means coupled between said control electrode and the comparing means, and responsive to said third signal for adjusting the value of said DC
current 10wing through said main electrodes to said predeter~ined value.
According to another aspect of the invention there is provided an amplifier adapted for coupling telephone signals to a telephone line while controlling the DC line current flowing through said telephone line to a predetermined value at a given telephone line loop resis~ance, said amplifier comprising: at least one inpu~ transistor having first and ':
- . . ' . ' !
~0726~
second main electrodes coupled across said telephone line, and having a control electrode; means for applying said telephone signals to said control electrode; first means coupled between said control electrode and one of said main electrodes for deriving a first signal related to the DC line current flowing through said main electrodes; second means for providing a reference signal related to said given tclephone line loop resistance; and means coupled to said first and second means and to said control electrod~, and responsive to said first signal and said reference signal for adjus~ing said DC line current to said predetermined value.
Brief Description of the Drawing The advantages of this invention will become more readily appreciated as the same becomes bet~sr unders~ood by reference to the following detailed description when taken in conjunction with the accompanying drawing wherein:
: Figure 1 is a combined schematic and functional diagram of an electronic telephone network in accordance with the principals of the ~ present invention; and, : Figures2-6 depict preferred circuitry for use in the various blocks of the functional diagram of Figure 1.
20" Detailed Description Referring now to Figure 1 there is shown a block diagram of the electronic network 10 in accordance with the present invention. Network 10 includes a low-output dynamic microphone 12 or any other suitable telephone signal input such as a data input. The output of microphone 12 is coupled to a fixed-gain amplifier 14 the output of which is coupled to a fixed attenuator 16 and to transmit detector 18. The output of detector 18 is coupled as a second input ~o attenuator 16 and as a : .
,, ~.
- -. .
.
. : : . . :
:: . . . ~ :
~7Z69~
first input to output impedance control 20. Th~ ~utput ~f a*tenuator 16 is coupled as a firs~ input to transmit signal equalizer 22 which may take the form of a variable attenuator as described more fully hereinafter.
The output of equalizer 22 is coupled to another fixed-gain amplifier 24 the output of which is coupled to line-driver 26 and as a first input to summing junction 28. The output of line-driver 26 is coupled to a two-wire *elephone line 30 wherein its conductors are depicted as 30a and 30b, by way of polarity guard 32. The network side of guard 32 is connected to the network by way of line conductors 30a' and 30b'.
Conductor 30a' is coupled as a second input to summing junction 28 by way of a fixed attenuator 34. Conductor 30a' is also coupled to ground by way of a filtering network comprising resistor Rl and filtering capacitor Cl. The junction of resis~or Rl and Cl provides a source of operating potantial tVs~ for the electronic network. The other side o the telephone line is connected to ground at conductor 30b' by way of a sensing resistor R2.
Conductor 30b' is also coupled as a second input to output impedance control 20. The ou~put of impedance control 20 is coupled to line-driver 260 The input of line-driver 26 is also coupled, by way o lead 36, as a second input to summing junction 28. The output of summing junction 28 is coupled as a first input to a variable attenuator 38. The output of attenuator 38 is also coupled to an output transducer 40 by ~`
way of a fixed gain receive amplifier 42. Pinally, operating potential Vs ls also coupled as a second input to attenuators 22 and 38 by way of leads 44 and 46, respectively.
The function of electronic ~elephone network 10 of Figure 1 is described as follows. As will be described more fully hereinater with . .
, . , . ~. -, ~: - . .
.. , . . . . . ~ . .. .. . . .
~97Z6~
reference to detailed circuitry Figures 2-S, a primary function of network 10 is to utilize the DC power available at the network end of the telephone line in a con~rolled manner to satisfy the requirements of:
(i) minimum DC current drain for a given loop resistance to ensure proper operation of the central office supervisory equipment; (ii) provide DC bias, or operating potential, to the line-driver amplifier and the receive amplifier to ensure satisfactory dynamic range under all specified operating conditions; and (iii~ to provide a DC voltage signal which accurately reflects the DC line current such ~hat the DC signal can be used for loss equalization over a given range of telephone line losses.
Since line-driver amplifier 26 sinks a majority of the telephone line current, its dominating influence is advantageously utilized ~o provide a stable and repeatable current drain which can be adjusted to set the network current drain level to a given specification.
The DC supply voltage or operating potential Vs of network 10 is preferably provided by a simple RC filter comprising resistor Rl and capacitor Cl. In one constructed embodiment resistor Rl was selected as 360 ohms and capacitor Cl as 220 MFD. For a typical telephone line, potential Vs varies from a minimum 0.8 to 2.2 volts depending on the telephone line loop resistance. Since this derived potential Vs is an accurate indicator of total loop resistance, and therefore DC line current, it îs used, in accordance with a feature of the present invention, to control the gain of attenuators 22 and 38 for cable loss equalization.
Further, the various gain stages of Figure 1 are designed to operate with s~able closed-loop gains over this a~ailable range of operating potential Vs. As will be described more fully hereinafter with reference to the remaining drawing igures, the value of resistor Rl was selected .
.. , ,. . , . , , ~ . .. .
, , ~, . ~ , , ~ . :.
- . ; . ~
- . :
~072G~8 to op~imize the d~lamic range of the transmitted and received signal even at very low ter~inal voltages. Further, the various stages that do not contribute directly to receive or transmit power outputs are designed to ut.ilize minimum DC current flow, thus this design eontributes to low DC voltage operation in accordance with the principals of the present invention.
Referring now more specifically to ~he various stages of the network 10 depicted in Figur~ 1, it can be s~en that the transmit signal flow from the dynamic transducer or microphone 12 is by way of amplifier 14, attenuators 16 and 22, and amplifiers 24 and 26. The overall transmit : -frequency response is determined primarily by microphone 12, amplifier 14, and transmit equalization attenuator 22. Amplifier 14 is selected - to have a fixed gain and in one constructed embodiment it provided 24 db gain at one KHz. Attenuator 16 is under the control of transmit detector 18 to pro~ide either 0 or 10 db of loss. That is, when a transmit signal whi.ch is greate~ than a given threshold level is provided by micro-phone 12, transmit detector 18 senses this transmit level and switches attenuator 16 to provide zero loss. Transmit detector 18 is preferably an A~ level detector having a fast response and slow release, and functions to reduce background noise during the receive operation of network 10 of Figure 1. Transmit detector 18 also functions to control the output of impedance control 20~
Attenuator 22 provid~s variable attenuation and frequency response which varies as a function of the operating potential Vs. Thus, attenuator 22 provides transmit equalization, for amplitude and frequency, for a wide range of different cable losses. Attenuator 22 is also used to limit transmit gain when the DC terminal voltage, or operating potential Vs, . ' 7Z69~
is very low~ thereby ~o prevent clipping at normal voice signal levels.
As previously alluded to, operating potential Vs may vary from a ~aximum of 2.2 volts, which corresponds to an essentially zero loss telephone line, down to 0.8 volts which corresponds to a maximum loss line and the minimum operating voltage of network 10, as utilized in a preferred embodiment of the present invention. The attenuation provided by attenua~or 22 is minimum when Vs is in the range of 1.1 to 1.15 volts.
This range corresponds to a typical total loop resistance of 2650 to 2200 ohms. respectively~ in a typical 48 volt system. As Vs increases from 1.15 to 2.2 volts --2.2 volts being the maximum voltage corresponding to essentially zero line loss-~ the average attenuation provided by attenuator 22 increases to its maxi~um. On the other hand, decreasing Vs from 1.1 volts to the minimum operating voltage of the system which is 0.8 vol~s, also causes the att~nuation provided by attenuator 22 to increase.
This increase in attenuation over the low voltage range is provided in order to correspondillgly decrease transmit gain at approximately the same rate as the transmit dynamic range decreases. Accordingly, this has the desirable effect of xetaining a margin of dynamic range above normal talking levels, there~y ~o prevent distortion as would result from clipping.
Accordingly, the electronic telephone netwoxk, in accordance with the present invention, operates with good quality, ableit with reduced gain, down to the very minimum DC terminal voltage of the network (approximately
Conventional telephone networks include a carbon mi.crophone, a transformer hybrid, a dynamic receiver, and various components such as resistors, capacitors, etc., and are used to interface with a standard two-wire telephone line with the telephone handset. Due to the passive nature of the primary components (e.g., the carbon microphone~ ~he standard telephone networks have several disadvantages such as large physical size, lack of fidelity, compatibility with other electronic telephone devices, and versatility. Elec~ronic telephone networks are also known in the art and obviate many of the above enumerated disadvantages while providing transmit and receive gain and power output sufficient to enable the use of low-eficiency dynamic transducers in the handsets.
However, these electronic telephone networks have several known dis-advantages in that they are expensive, fragile and, more importantly, incompatable with many existing telephone facilities or practices. For example, the incompatibility pro~lem arises due to the relatlvely high DC voltage requirement of the electronic networks as they derive their power from the telephone line. Disadvantageously, this prevents or severclylimits parallel operation with conventional telephone networks.
~urther~ore, ~he high DC resistance of these prîor art networks limits the maximum telephone line loop lengths on exchanges with conventional supervision equipment which require certain minimum current drain levels for satisfactory operation.
These and other disadvan~ages are overcome by the present , ' . - .... . . . :-. . . , . ,. - , , .
~7~
inven~ion wherein an electronic telephone network is provided which provides the normal functions of an electronic telephone network bu~ which further provides: an electronic hybrid to separate the receive signal from the transmit signal while eliminating objectionable sidetone; correct DC line current drain levels for ~elephone central office supervision equipment; relatively cons*ant re~eive and transmit lev~ls over the entire range of conventional telephone line distances between the network and the exchanges; efficient use of available DC power; independence of the AC
impedance characteristics of the telephone line; and, more particularly, while providing operation for DC terminal vol~age between the normal maximum of 8.0 volts down to, and including, 2.2 volts.
Summary of ~he Invention Briefly, an electronic network for receiving and transmitting ; telephone signals over the two-wire line is provided. The network includes a first amplifier and means for applying the transmit portion of ~he telephone signals to the first amplifier. First means are provided which are responsive to the first amplifier for applying the amplified signal output of the amplifier to the line. Second means are provided which are ~ responsive to the first amplifier and the output of the firs~ means for 20 separating the received portion of the telephone signals from the composite telephone signals at an output thereof. The network includes a second amplifier coupled to the ou~put of the second means for increasing the level of the separated received signals; and, third means for deriving operating potential for the network from the telephone lines are also provided. Fourth means couple the first amplifier to the first means and are responsive to the operating potential for altering ~he frequency/amplitude characteristics of the amplified signal output _ z _ : ~ ,.. - ' ' . . .
-:
~0~72~
of the first amplifier in accordance with the level of the operating potential. The preferred circuitry embodiments include means for providing low line terminal voltage operat.ion, and, control of the output impedance of as associated line-driver amplifier to reduce the electronic ~ -hybrid sensitivity and, therefore, the sensitivity of the sidetone signal to the line impedance characteristics.
According to the invention there is provided an amplifier adapted for coupling telephone signals to a ~elephone line and for controlling the DC current flowing through said telephone line to a predetermined value at a given telephone line loop resistance, said amplifier comprising;
at least one output transistor having firs~ and second main electrodes coupled across said tele~hone line, and having a control electrode; means or applying said telephone signals to said control electrode; means coupled between said control electrode and one of said main electrodes for deriving a firs~ signal related to the DC current flowing through said main electrodes; means coupled across said telephone line for deriving a second signal related to the DC voltage appearing across said ~elephone line;
means for comparing said first and second signals to provide a third signal indicative of the difference betw~en s~id first and second signals;
and means coupled between said control electrode and the comparing means, and responsive to said third signal for adjusting the value of said DC
current 10wing through said main electrodes to said predeter~ined value.
According to another aspect of the invention there is provided an amplifier adapted for coupling telephone signals to a telephone line while controlling the DC line current flowing through said telephone line to a predetermined value at a given telephone line loop resis~ance, said amplifier comprising: at least one inpu~ transistor having first and ':
- . . ' . ' !
~0726~
second main electrodes coupled across said telephone line, and having a control electrode; means for applying said telephone signals to said control electrode; first means coupled between said control electrode and one of said main electrodes for deriving a first signal related to the DC line current flowing through said main electrodes; second means for providing a reference signal related to said given tclephone line loop resistance; and means coupled to said first and second means and to said control electrod~, and responsive to said first signal and said reference signal for adjus~ing said DC line current to said predetermined value.
Brief Description of the Drawing The advantages of this invention will become more readily appreciated as the same becomes bet~sr unders~ood by reference to the following detailed description when taken in conjunction with the accompanying drawing wherein:
: Figure 1 is a combined schematic and functional diagram of an electronic telephone network in accordance with the principals of the ~ present invention; and, : Figures2-6 depict preferred circuitry for use in the various blocks of the functional diagram of Figure 1.
20" Detailed Description Referring now to Figure 1 there is shown a block diagram of the electronic network 10 in accordance with the present invention. Network 10 includes a low-output dynamic microphone 12 or any other suitable telephone signal input such as a data input. The output of microphone 12 is coupled to a fixed-gain amplifier 14 the output of which is coupled to a fixed attenuator 16 and to transmit detector 18. The output of detector 18 is coupled as a second input ~o attenuator 16 and as a : .
,, ~.
- -. .
.
. : : . . :
:: . . . ~ :
~7Z69~
first input to output impedance control 20. Th~ ~utput ~f a*tenuator 16 is coupled as a firs~ input to transmit signal equalizer 22 which may take the form of a variable attenuator as described more fully hereinafter.
The output of equalizer 22 is coupled to another fixed-gain amplifier 24 the output of which is coupled to line-driver 26 and as a first input to summing junction 28. The output of line-driver 26 is coupled to a two-wire *elephone line 30 wherein its conductors are depicted as 30a and 30b, by way of polarity guard 32. The network side of guard 32 is connected to the network by way of line conductors 30a' and 30b'.
Conductor 30a' is coupled as a second input to summing junction 28 by way of a fixed attenuator 34. Conductor 30a' is also coupled to ground by way of a filtering network comprising resistor Rl and filtering capacitor Cl. The junction of resis~or Rl and Cl provides a source of operating potantial tVs~ for the electronic network. The other side o the telephone line is connected to ground at conductor 30b' by way of a sensing resistor R2.
Conductor 30b' is also coupled as a second input to output impedance control 20. The ou~put of impedance control 20 is coupled to line-driver 260 The input of line-driver 26 is also coupled, by way o lead 36, as a second input to summing junction 28. The output of summing junction 28 is coupled as a first input to a variable attenuator 38. The output of attenuator 38 is also coupled to an output transducer 40 by ~`
way of a fixed gain receive amplifier 42. Pinally, operating potential Vs ls also coupled as a second input to attenuators 22 and 38 by way of leads 44 and 46, respectively.
The function of electronic ~elephone network 10 of Figure 1 is described as follows. As will be described more fully hereinater with . .
, . , . ~. -, ~: - . .
.. , . . . . . ~ . .. .. . . .
~97Z6~
reference to detailed circuitry Figures 2-S, a primary function of network 10 is to utilize the DC power available at the network end of the telephone line in a con~rolled manner to satisfy the requirements of:
(i) minimum DC current drain for a given loop resistance to ensure proper operation of the central office supervisory equipment; (ii) provide DC bias, or operating potential, to the line-driver amplifier and the receive amplifier to ensure satisfactory dynamic range under all specified operating conditions; and (iii~ to provide a DC voltage signal which accurately reflects the DC line current such ~hat the DC signal can be used for loss equalization over a given range of telephone line losses.
Since line-driver amplifier 26 sinks a majority of the telephone line current, its dominating influence is advantageously utilized ~o provide a stable and repeatable current drain which can be adjusted to set the network current drain level to a given specification.
The DC supply voltage or operating potential Vs of network 10 is preferably provided by a simple RC filter comprising resistor Rl and capacitor Cl. In one constructed embodiment resistor Rl was selected as 360 ohms and capacitor Cl as 220 MFD. For a typical telephone line, potential Vs varies from a minimum 0.8 to 2.2 volts depending on the telephone line loop resistance. Since this derived potential Vs is an accurate indicator of total loop resistance, and therefore DC line current, it îs used, in accordance with a feature of the present invention, to control the gain of attenuators 22 and 38 for cable loss equalization.
Further, the various gain stages of Figure 1 are designed to operate with s~able closed-loop gains over this a~ailable range of operating potential Vs. As will be described more fully hereinafter with reference to the remaining drawing igures, the value of resistor Rl was selected .
.. , ,. . , . , , ~ . .. .
, , ~, . ~ , , ~ . :.
- . ; . ~
- . :
~072G~8 to op~imize the d~lamic range of the transmitted and received signal even at very low ter~inal voltages. Further, the various stages that do not contribute directly to receive or transmit power outputs are designed to ut.ilize minimum DC current flow, thus this design eontributes to low DC voltage operation in accordance with the principals of the present invention.
Referring now more specifically to ~he various stages of the network 10 depicted in Figur~ 1, it can be s~en that the transmit signal flow from the dynamic transducer or microphone 12 is by way of amplifier 14, attenuators 16 and 22, and amplifiers 24 and 26. The overall transmit : -frequency response is determined primarily by microphone 12, amplifier 14, and transmit equalization attenuator 22. Amplifier 14 is selected - to have a fixed gain and in one constructed embodiment it provided 24 db gain at one KHz. Attenuator 16 is under the control of transmit detector 18 to pro~ide either 0 or 10 db of loss. That is, when a transmit signal whi.ch is greate~ than a given threshold level is provided by micro-phone 12, transmit detector 18 senses this transmit level and switches attenuator 16 to provide zero loss. Transmit detector 18 is preferably an A~ level detector having a fast response and slow release, and functions to reduce background noise during the receive operation of network 10 of Figure 1. Transmit detector 18 also functions to control the output of impedance control 20~
Attenuator 22 provid~s variable attenuation and frequency response which varies as a function of the operating potential Vs. Thus, attenuator 22 provides transmit equalization, for amplitude and frequency, for a wide range of different cable losses. Attenuator 22 is also used to limit transmit gain when the DC terminal voltage, or operating potential Vs, . ' 7Z69~
is very low~ thereby ~o prevent clipping at normal voice signal levels.
As previously alluded to, operating potential Vs may vary from a ~aximum of 2.2 volts, which corresponds to an essentially zero loss telephone line, down to 0.8 volts which corresponds to a maximum loss line and the minimum operating voltage of network 10, as utilized in a preferred embodiment of the present invention. The attenuation provided by attenua~or 22 is minimum when Vs is in the range of 1.1 to 1.15 volts.
This range corresponds to a typical total loop resistance of 2650 to 2200 ohms. respectively~ in a typical 48 volt system. As Vs increases from 1.15 to 2.2 volts --2.2 volts being the maximum voltage corresponding to essentially zero line loss-~ the average attenuation provided by attenuator 22 increases to its maxi~um. On the other hand, decreasing Vs from 1.1 volts to the minimum operating voltage of the system which is 0.8 vol~s, also causes the att~nuation provided by attenuator 22 to increase.
This increase in attenuation over the low voltage range is provided in order to correspondillgly decrease transmit gain at approximately the same rate as the transmit dynamic range decreases. Accordingly, this has the desirable effect of xetaining a margin of dynamic range above normal talking levels, there~y ~o prevent distortion as would result from clipping.
Accordingly, the electronic telephone netwoxk, in accordance with the present invention, operates with good quality, ableit with reduced gain, down to the very minimum DC terminal voltage of the network (approximately
2.2 volts) which also corresponds to the minimum 0.8 volts operating potential minimum. Thus, parallel operation with other networks including conventional networks is provided at longer telephone line lengths and at the corresponding low DC terminal voltages.
As just described, increasing the operating potential Ys from . . . . . . . . . .
:: . . . .. : . .. , , ~
~L63 7;~6~
1.15 to 2.2 volts causes the average attenuation to increase to a maximu~. Thc characteristics of a~tenua~or 22 are selected such that the attenuation at the upper end of the frequency spectrum increases at a faster rate (as Vs increases~ than at the lower end of the spectrum or vamp. Stated differently, high frequency roll-off becomes more pronounced for larger values of Vs. This attenuation versus ~requeney characteristic is designed to provide near perfect transmit equalization in both amplitude and frequency response for #26 gauge cable from 0 to 21.5 K feet, as fed from a 48 volt, 400 ohm bridge feed.
Amplifier 24 functions as a fixed gain wide band circuit that increases the relatively low output signal from attenuator 22 su~icient to drive line-driver 26 and to provide a usable signal ~o the input of - separator 28 at input lead 36. However, as previously discussed, since amplifier 24 does not directly contribute to the power output of network 10, it is provided as a low current drain circuit.
Line driver amplifier 26 in addition to establishing and dominating the DC characteristics of the network, also provides the transmit signal power to drive the telephone line and establishes the AC
output impedance of the network. Amplifier 26 is preferably a relatively wide band circuit having a very high open-loop gain. This is done so that the various operating characteristics of amplifier 26 can be determined by passive, external feedback components, such as resistors, and the telephone line itself. l`he output impedance of amplifier 26 is determined by utilizing current eedback from the sensing resistor R2 and in conjunction with a degree of voltage feedback. ~esistor R2 typically has a very low ; value and in one constructed embodiment resistor R2 was three ohms.
The terminal impedance of network 10 ~which impedance determines _ 9 _ .
,.: . . - : . : . . .~ . .- . :.. ; . - : . . -.
~1~7%698 the return loss of the system) is determined by the output impedance Ro of amplifier 26. As pr~viously alluded to and as discuss~d in ~lore detail hereinafterl the output impedance of amplifier 26 is determined by the voltage and current feedback around amplifier 26. Output impedance control 20 functions to switch a portion of the feedback network of amplifier 26 to adjust the output impedance R to either one of two values. That is~
amplifier 26 operates with one value of Ro during transmit ~typically 300 ohms) and at the other value during receive operation ~typically 900 ohms). The transmit or reeeive state is determined by transmit detector 18. As discussed more fully hereinafter, the two level impedance technique is u~ilized to reduce the sensitivity of the sidetone signal to different line impedances. That is, return loss during the higher impedance receive state is near perfect during receive but poor during transmit. Accordingly, line length has very little effect on te~minal impedance.
The electronic hybrid function, which separates the receive signal from the composite transmit and receive signal at the telephone line and routes the recei~e signal to the receive amplifier circuit~y, is provided by amplifier 26, attenuator 34 and summing junction 28. The actual signal separation is provided at the summing junction 28. The signal provided at input lead 36 by the output of amplifier 24 is the transmit signal only due to the low output impedance of amplifier 24 and theiisolation of the receive signal provided by amplifier 26. Amplifier 26 amplifies and inverts this transmit signal by a factor of K. The output ~ of amplifier 26, KTX and the receive signal Rx are attenuated by attenuator ; 34 by a factor of l/k. Thus, attenuator 34 reduces the output of amplifier 26 to substantially the same amplitude as the output signal at the output - 10 - "'"' : ~ ' :: . . . - . . , : - ~ ~
~LC)7;~
of amplifier 2~. AccordinglyJ these t~o signals cancel in the summing junction 28 and only an ~tt~nuated receive signal remalns.
Thus, amplifier 2~ and attenuator 34 are called upon to provide transmit signals at the input of summing junction 28 with equal amplitudes and opposite polarities for all line conditions over the ~requencies of interest. Although this function cannot be performed perfectly, some residual transmit signal is nevertheless desirable so as to provide some ; sidetone. Actually, the ideal network would produce optimum sidetone level irrespective or regardless of line conditions. ~s discussed more fully hereinafter, electronic telephone network lO in accordance with the -teachings of the present invention, achieves a degree of independence from line conditions because of the unique design of attenuator 34 and amplifier 26. In one constructed embodiment, attenuator 34 was a resistive divider and RC phase shift network which would optimize to provide maximum cancellation over the audio frequency band for 1800 ohms of 26 gauge line. This is the worst case condition for sidetone reduction since attenuators 22 and 38 function to provide maximum gains in this range. Whereas, however, the reduction in gain provided by attenuators 22 and 38 at very long, or very short, line lengths tends to reduce the respective gains and to thereby simplify the compensation requirements.
Ideally, amplifier 26 should function as a perfect voltage source;
that is, with zera output impedance. In this case, attenuator 3~ could then be adjusted to exactly offset the constant voltage gain of amplifier 26 and perfect cancellation would therefore occur. However, it is known in the art that such ideal characteristics cannot be obtained as the output of amplifier 26 is across the telephone line and otherwise establishes the network impedance; that is, the received signal must necessarily appear, or , .
,~ ;; -- 1 1 --.
~ . , 72~8 ~e developed, across the output impedance of amplifier 26. However, since the terminal impedance for minimum return loss (i.e. impedance matching) is most critical when a signal is being received, this distinction in criticality is advantageously utilized, in accordance with the principles of the present invention, to reduce sensitivity of the sidetone signal to telephone line impedance variations. This is accomplished by virtue of the switching between two impedance levels in the design of amplifier 26.
In curren~ly preferred practice, the output impedance of amplifier 26 is approximately ~00 ohms ~which is the standard conventional or typical telephone line impedance) during receive and standby or idle periods.
Accordingly~ this receive impedance value permits réceive signals to develop normal voltage levels across the network terminals and provides near optimum re~urn loss.
I~ however, the network impedance, or Ro~ were maintained at ~00 ohms during transmit, the voltage gain of amplifier 26 would be a function of telephone line impedance and sidetone levels would vary greatly with different cable lengths. Therefore9 in accordance with a feature of the presen~ invention, the network impedanc0 Ro is switched to approximately 300 ohms during transmit which brings amplifier 26 significantly nearer to the ideal perfect voltage source. It should now be appreciated that an ~O of 300 ohms reduces the effect of telephone line impedance variations on the voltage gain of amplifier 26, and sidetone levels remain substantially constant for different cable lengths. It has ~een ~ouncl that this techniqu reduces sldetone variations relative ~o line impedance by approximately six db. This is particularly desirable during unfavourable impedance conditions such as parallel operation with conventional telephone networks. Further, sidetone changes between receive and transmit conditions hai been found .. , .. , . ; :, . . .. ,. ... ,~ , . :. -~,, - , . - ..... .. ., . ~. .. ..
, ... . . : . ,: : - . , ~ , . ~ : :
,: : : ..... . ... ::: : ., ~.. , . :
to be negli~ible because of the reduced transmit gain during receive operation as provided ~y a~tenuator 16~ That i5, the reduced gain approximately offsets the degraded cancellation a~ summing junction 28 due to the increase of Ro from 300 to its 900 ofim le~el.
The output of summ;ng junction 28 is t~e received signal plus the sidetone signal. Receive equalization attenuator 38, which is functionally and structurally similar to attenuator 22, equalizes the received signal in both ampli~ude and frequenc~ for differen~ cable lengths.
In one constructed embodiment, attenuator 38 was selected to provide subs~antially identical operation to that of att~nuator 22 between 0 and 1800 ohms of telephone line resistance. The loss of attenuator 38 i5 minimum when the resistance of the t~lephone line is approximately 1800 ohms; however, unlike attenuator 22 the loss of attenuator 38 remains at its minimum level as the telephone line resistance~ or effective telephone line length, increases from 1800 ohms, i.e., reducing the DC terminal voltage.
Finally, amplifier 42 provides a fixed gain and amplifies the relatively low receive signal output from attenuator 38 to a level sufficient to drive output transducer 40. A relatively high gain is required in this stage to overcome the losses introduced by attenuator 34 prior to summing junction 2g. As will be discussed more fully hereinafter with reference to the detailed circuitry figures, the DC characteristics of amplifier 42 provide a maximum voltage swing across output transducer 40 for any DC supply potential Vs.
Referring now to Pigure 2, there are showm schematic circuit diagrams of amplifier 14, attenuator 16 and transmit equalization control 22 suitable for use in the electronic telephone network 10 of Figure 1.
:
~; 13 ': :
, . . . .. .... . ..
~7Z698 Amplifier 14 comprises a conv~ntional common emitter ampli~ier stage ~hich in one constructed embodiment provided 24 db gain, ~owever~ the current supply to transistor Ql of amplifier 14 is provided ~y a constant current load transistor Q2. That is, Ql i5 an active constant curren~ load which provides a substantial DC ~ias current to t~e coll~ctor of Ql ~which is necessary for low noise and sufficient gain at low values of Vs~ without loading the AC signal. That is~ ~ransistor Q2 provides a large AC impedance while supplying a substantially high level of current to transistor Ql.
Transistor Q2 is biased in~o operation as a constant current source by current mirror diode Dl. This curren~ mirror diode potential at the base electrode of transistor Q2 is also made available as a current mirror diode signal ~o the active load or constant current source transistor of other circuits of electronic telephone network 10.
The input signal provided by transducer 12 is also coupled to transmit detector 18 by way of amplifier 14. Detector 18 provides a logic "1" at its ou~put for receive or normal conditions; and a logic "0" for the transmit mode. Detector 18 may take any one of a number of suitable circuit configurations but preferably functions as a fast attack/slow decay switching device thereby ~o provide substantially undetectable control of transmit gain and output impedance. Thè output of detector 18 is coupled to output impedance control 20 and to the input or base elsctrode of transistor Q3 of attenuator 16. It can be seen that when detector 18 provides its logic l output transistor Q3 is biased into conduction thereby shunting a portion of the output signal from amplifier 14 to ground through the main electrode of transistor Q3.
Transmit equalization attenuator 22 comprises a two-pole filter/
attenuator that u~ilizes the dynamic resistance of diodes as gain control . "' ' - :
, . ~ ~ , . :
. , : . .. , ,. ... ': . '.: . ;:
. . : . .
~7Z~98 elements. The transmit audi~ signal flo~ is through resistor R101 and resistor R102. Attenuation frequency response shaping is pro~ided by capacitor C101 in com~ination wit~ d~ode D101, and capacltor ClQ2 ;n combina~ion wi~h diode D102, which function to shunt the transmit signal to ground. Diodes D101 and D102 function to provide variable resistance elements.
The dynamic or small-signal resiskance of diode D101 is determined by the DC current flowing from Vs t~rough diodes D103 and D104~ resistor R103 and diode D101 itself. When Vs is near its maximum value9 the resistance of diode D101 is relatively low and the signal loss through capacitor C101 and diode D101 is relativcly large. However, as Vs decreases toward its minimum value, the resistance o diode D101 increases substantially and signal attenuation accordingly decreases. When Vs is approximately 1.15 volts~ diodes D104 and D101 are at or near cut-off and, therefore, there is substantially no signal loss. Thus, overall transmit gain when diodes D104 and D101 are cut-off is maximum.
Capacitor C102 and diode D102 function in a similar manner. However, due to the relatively small size of capacitor C102 this circuit functions to place a "zero" much higher in the audio spectrum than th0 circuit of capacitor C101 and diode D101. Accordingly, the circuit associated with capacitor C102 and diode D102 functions to provide a majority of the frequency compensation of transmit equalization control 22 in accordance with a feature of the present invention. Stated somewhat differently, high-frequency roll-off becomes more pronounced as Vs increases.
Resistors R104 and R105 of transmit equalization control 22 function to increase the voltage drop of diodes D103 and D104 respectively.
This functions to shape the overall characteristics of transmit .
~7Z6913 equaliza~ion 22 so as to equalize or accept #26 gauge cable. The remaining components associ~ted with ~ransmit equalization control 22 of Figure 2 function to reduce transmit gain as Ys drops f.rom approximately 1.1 volts to the minimum operating potential o 0.8 volts. The current flowing through resistor R105 is larger than the maximum current flowing through resistor R106 throughout a majority of the range of operating potentials Vs.
Accordingly, curren~ me~er transistor Q4 remains saturated as it can't source all of the current ~hat is programmed for it by diode D105.
Accordingly, this functions to keep transistors Q5 and Q6 cut-of~ and essentially out of the circuit of transmit equaliza~ion control 22.
However, as Vs drops ~o approximately 1.1 vol~s --noting that transmit gain is maximum at a Vs of approximately 1.15 volts-- the current flowing through diode D105, D106 and R105 is reduoed to a value equal to the maximum current flowing through diode D107 and resistor R106 due to the impending cut-off of diodes D105 and D106. A further reduction of Vs results in current flow through diode D107 and the mirrored current flow from transistors Q5 and Q6 because transistor Q4 can no longer supply the maximum needed current flow through resistor R106. Accordingly, the current from transistors Q5 and Q6 flows into diode D101 and increases attenuation of ~he transmit signal. As previously discussed, this attenuation begins at approximately 1.1 volts and increases as the operating potential of Vs decreases to the minimum value of 0.8 volts. It should now be appreciated that transmi~ equalization control 22 is preerably designed empirically with respect to its attenuation and frequency response characteris~ics to match a given gauge of cable ~such as #26 gauge cable) for equallzation purposes.
Referring now to ~igure 3, there are shown schematic diagrams . ~ ~ . . . . .
- : . , . : . :: . ~
, : , . ~ . : . . ~ . -: . . . . : . ::
- . .. . :. - . . - - ,- :
7~6~
of line-driver amplifier 26, output impedance control 20 and polarity guard 22. The operation of line~driver ampli~ier 26 will be described in conjunction with Figures 4 and $ whic~ respec~ivel~ provide simplified diagrams of the DC and AC equivalent circuits of amplifîer 26.
As previousl~ alluded to, the ac~ievement of maximum transmit dynamic range for operation at long telep~one line lengthsor parallel operation, particularly when the DC terminal voltage is low, requires that the collector-emitter path of t~e OUtp~lt transistor be coupled directly across the line and capable of saturation operation. That is! any serially coupled element for DC line current stabilization, such as an emitter resistor, for example9 reduces the transmit signal output voltage swing.
In Figures 3 and 4, the paralleled output transistors QO are shown as being directly coupled across the telephone line except for the voltage drop induced by polarity guard 32 which is represented by a diode 32' in Figure 4 and the drop of sensing resistor R2. The telephone line in Figure 4 is schematically represented by the exchange battery Be, the terminal impedance of the central exchange Re and the telephone line resistance Rlo As will be described in detail hereinaf~er, a unique polarity guard is utilized, in accordance with the principals of the present invention, to minimize both DC and signal loss. Sensing resistor R2 is used for AC eedback and typically has a very small value and therefore only a negligible efect on the DC characteristics of the overall electronic telephone network. In one constructed embodiment, sensing resistor R2 had a value of 3 ohms.
0utput transistors QO sink a majority o the DC line current.
Accordingly, it is necessary to accurately control this current for overall control of the electronic networks DC characteristics. The paralleled base electrode connections of transistors QO provide both the information .
'' Z69~3 signal and control necessary for accurately establishing collector current as the base-emitter junction voltage is an accurate indicator of collector current flow. In the presen~ invention9 ~his base-emitter voltage is monitored and used in a feed~ack system to determine the base drive curren~
necessary to establish the required collector current. It will be appreciated by those skilled i~ the art t~at accurat~l~ matched output transistors, such as that resulting from integrated circuit fa~rication, are preferable for this design.
Assuming initially that resistor R120 of Figures 3 and 4 is shorted or 0 o}~ns~ the operation of amplifier 26 is described as follows.
Transis~or Q10 and output transistors QO would therefore ha~e the same base-emitter voltages and since transistors Q10 and QO are matched, the collector current in each transistor would be equal. Accordingly, the collector current o~ transistor Q10 would accurately reflect the DC line current flowing through output transistors QO In actual operation, resistor R120 has a finite value and in one constructed embodiment it had a ~alue of approximately 24~0 ohms. Nevertheless, the accurate current sensing function of transistor Q10 is maintained and the otherwise wasted current of transistor Ql~ is substantially reduced. The collector current of transistor Q10~ which is an accurate ~unction of the collector currents of output transistors QO' flows through diode DllO which, in turn, controls current mirror transistor Qll. Thus, the collector current of Qll is also approximately equal to the collector current o~ Q10. The collector current of transistor Qll is converted to a voltage signal by resistor R121.
Accordingly, the voltage developed across resistor R121 is also an accurate indication of the collector current flo~ing through output ~Tansistors ~O.
The collector of transistor Qll is coupled as a first input : . . ~
... ,- . .. . .
, -.. . - - . - : . , . -; . - , ,: . - : :
. ` :- : : : :- . :, . `: . .
~:: : ~ . : . . - : - . :
6~8 ~"-") to an operational ampli~i~r AII~ Resistors R122 and R123 form a voltage divider and the junction of resistors 122 and 123 is coupled to the other input ("~") of amplifier AII b~ way of a buffer amplifier AI.
Amplifier AII functions to force the voltage across resistor R121 to be equal to the voltage provided ~y the voltage divider ~y adjusting the base drive curren~ to output transistors QO Now, since the voltage across resistor R121 is accurately dependent on DC line current, and ~he voltage provided by the voltage divider resistors R122 and R123 is accurately dependant on DC line voltage, amplifier AII establishes an equilibrium condition controlled by the to~al loop resistance that fixes the DC
characteristics of the electronic telephone network. The voltage divider comprising resis~ors R122 and R123 and operational amplifier AI are actually an integral par~ of fixed gain amplifier 24 of Figure 1. }lowever, since ~ they function to control the DC characteristics of amplifier 26 they are : depicted in Figure 4.
In Figure 3, amplifier AII of Figure 4 is comprised of transistors Q13 through Q20. Transistors Q13 and Q14 provide a balance differential input; and transistors Q12 and Q16-18 are current mirrors which operate under the control of resistor RlZ4 and diode ~111. It can be seen that due 20 to the active or constant-current loads, all gain producing transistors of amplifier 26 of Figure 3 are biased in their linear operation region even when the source of operating potential, Vs, is as low as 0.8 volts.
Referring now to Figure 5, there is shown a simplified AC
~ equivalent circuit diagram of amplifier 26 of Figure 3. Since the AC
:~ characteristics of amplifier 26 are affected b~ output impedance control 20~
circuit 20 is also illustrated in Figure 5. Control 20 includes a switching transistor Q301 which is seriall~ coupled with capacitor C301 and resistor "
19 - :
. ~ : , , : . . ., - .:
. '' : ,: '' ~ - . .
, ~L~7Z69~
R301 which components are coupled across resistor R302. Ac~ordingly, transistor Q301 responds to the output of transmit de~ector 1~ to vary the resistance of resistor R302 which is disposed in t~e eed6ack loop of amplifier 26. That is, the normal or receive state of transmit detector 18 is a logic "1" which keeps transistor 301 turned on. This parallels resistor R301 with resistor R302 and results in the network impedance Ro being relatively high. During transmit, the output of detector 18 goes to its logic "0", or "low" state which ~urns off transistor Q301. This essentially removes resistor R301 from the circuit which results in Ro going to its lower impedance value. In one construc~ed embodiment the component values of amplifier 26 were selected such tha~ the normal or receive impedance, Ro, was 900 ohms whereas the transmit state impedance Ro was approximately 300 ohms.
In Figure 3, amplifier AIII of Figure 5 comprises the differential input transistors Q13 and Q14 of Figure 3, which drive the directly coupled common emitter amplifiers Q15, Ql9 and Q20. Q20 drives the parallel base connection of output transistors QO' i.e., the line-driver. Q20 is biased to supply enough drive signal to saturate output transistors QO even when the DC terminal voltage is near its minimum. Translstors Q16-18 are active loads for transistors Q15 and Ql9 and contribute to the very high open loop gain of amplifier 2~. Transistor Q12 provides a current source bias for th0 emitters of the differential input pair, ~13 and Ql~. Capacitor C112 references the non-inverted input of amplifier AIII to ground and, more importantly, it removes all AC feedback from the DC feedback path provided by transistor Qll as discussed wi~h reference to the DC equivalent circuit of Figure 4. As previously alluded to, sensing resistor R2 provides a sampling point for the AC ~eedback of amplifier 26 as illustrated in Figure 5.
: .
: "' .. ....... ,, :,. .. .. ... . . . . ... . . . . . .
: .. ~. . ... : :':' : . - . - , . , :, : -. :' : : . .. :: - .,... : . ::
: . . . . . ~ : : :.: .. . . . , : . ,: ::
1~17~698 Referring again to Figure 3, the opera~ion of polarity guard 32 will be briefly descrihed~ Assuming that conductor 30a is the positive potential side of the line, this condition ~orward ~iases transistor Q401 and diode D401. Transistor Q401 would then receive its ~ase drive current through resistors R401 and R401~ which complete the circui~ to the negative potential conductor 30b. T~us, t~e transistor Q401 can fully saturate such that i~s Vce is on the order of 0.15 volts. Thus, the total voltage drop across the polarity guard is 0.8 volts which represents the 0.15 volt Vce + 0.65 volts ~or one Vd drop). Similarly, when conductor 30b is the positive potential side of the telephone line, transistor Q402 and diode D402 are forward biased. Thus~ polarity guard 32 supplies operating potential of the correct polarity tothe electronic telephone network of the present invention regardless of the polarity of the telephone line. This is important as a practical consideration as in many telephone systems the respective polarities of the various lines are not consistent with one another. Finally, capacitors C401 and C402 function to maintain the DC
drive to the transistors Q401 and Q402 during large transmit signal excursions.
Referring now to Figure 6, there is shown a schematic diagram of summing junction 28 and receive attenuation control 38 in accordance with the principals of the present invention.
It can be seen that resistors R501 and R502 are disposed across the telephone line in a voltage divider network configuration. The voltage tlwsly provided functions to reduce both transmit and receive voltage signals. The ratio of this voltage divider in conjunction with the ratio of resistor R503 and the shaping network, comprising R504~ C501 and C502, is selected such that the transmit voltage from ampliier 24 of Figure 1 offsets or nulls the inverted amplified transmit voltage provided by line-. ..
- . . .. : .. : . : . . : : . -.,,, .. , ... . .~
.: -: . ; . ' . .: , driver 26 at point E. As previousl~ discussed, the received voltage is attenuated but it i5 not nulled by s~ ~ ing junction 28.
The division ratîo of resistors R501 and R502 and ~he impedance of the shaping network comprising capacitors C501, CS02 and resistor R504 Q
were empirically designed to pro~ide optimu~ null across the audio spectrum for 21.5 K feet of #26 gauge ca~le. ~t has also ~een found that this selection provides good performance not onl~ wit~ 21.5 K feet of #26 gauge cable, but als~ for all practical operation conditions. Accordingly~
summing junction 28, in conjunction with the equalization and output impedance control, in accordance with the present invention, provides optimum sidetone control for all practical operating conditions --including parallel operation with conventional telephone networks.
Referring now to the receive equali~ation attenuator 38 of Figure 6, it can be seen that attenuator 38 is also a two-pole filter attenuator similar in function and structure to attenuator 22 of ~igures 1 and 2.
Accordingly, the operation of attenuator 38 need no~ be described in great detail herein. However, it can be seen that both receive and sidetone signals flow through resistors R601 and R602. ~urther~ capacitor C601 in combination with diode D601, and capacitor C502 in combination with diode D602 provide the control of attenuation and frequency response characteristics, in response to the value of the derived operating potential Vs~
It should be noted however, with reference to attenuator 38 of Figure 6, that the dynamic resistance of a diode is most useful as a variable resistance when the applied AC signal level is kept at a relatively low value so as to prevent excessive distortion. As a practical matter, prevention of load distortion necessitates that the AC signal not ~ ' ;~
.. , . ; ~ , : . . --- ~7~26~3 exceed approximately lO or 12 milliYolts. For this reason, relatively small signal levels are applied to and derived from summing junction 28 and relatively large gains are provided by the receive amplifier 42 of Figure 1, in accordance with anot~er feature of the present invention.
~ hat has been taught, thenJ is an electronic telephone network facilitating, notably, automatic telephone line equalization, substan~ially reduced sensitivit~ of the associated sidetone signal, and operation at very low terminal voltage levels including parallel operation at low voltage operations~ The form of the invention illustrated and described herein is but a preferred embodiment of these teachings, in the form currently preferred for manufacture. It is shown as an illustration of the inventive concepts, however, rather than by way of limitation, and it is pointed out that various modifications, and alterations may be indulged in within the scope of the appended claims.
,: , , . - . . : :
As just described, increasing the operating potential Ys from . . . . . . . . . .
:: . . . .. : . .. , , ~
~L63 7;~6~
1.15 to 2.2 volts causes the average attenuation to increase to a maximu~. Thc characteristics of a~tenua~or 22 are selected such that the attenuation at the upper end of the frequency spectrum increases at a faster rate (as Vs increases~ than at the lower end of the spectrum or vamp. Stated differently, high frequency roll-off becomes more pronounced for larger values of Vs. This attenuation versus ~requeney characteristic is designed to provide near perfect transmit equalization in both amplitude and frequency response for #26 gauge cable from 0 to 21.5 K feet, as fed from a 48 volt, 400 ohm bridge feed.
Amplifier 24 functions as a fixed gain wide band circuit that increases the relatively low output signal from attenuator 22 su~icient to drive line-driver 26 and to provide a usable signal ~o the input of - separator 28 at input lead 36. However, as previously discussed, since amplifier 24 does not directly contribute to the power output of network 10, it is provided as a low current drain circuit.
Line driver amplifier 26 in addition to establishing and dominating the DC characteristics of the network, also provides the transmit signal power to drive the telephone line and establishes the AC
output impedance of the network. Amplifier 26 is preferably a relatively wide band circuit having a very high open-loop gain. This is done so that the various operating characteristics of amplifier 26 can be determined by passive, external feedback components, such as resistors, and the telephone line itself. l`he output impedance of amplifier 26 is determined by utilizing current eedback from the sensing resistor R2 and in conjunction with a degree of voltage feedback. ~esistor R2 typically has a very low ; value and in one constructed embodiment resistor R2 was three ohms.
The terminal impedance of network 10 ~which impedance determines _ 9 _ .
,.: . . - : . : . . .~ . .- . :.. ; . - : . . -.
~1~7%698 the return loss of the system) is determined by the output impedance Ro of amplifier 26. As pr~viously alluded to and as discuss~d in ~lore detail hereinafterl the output impedance of amplifier 26 is determined by the voltage and current feedback around amplifier 26. Output impedance control 20 functions to switch a portion of the feedback network of amplifier 26 to adjust the output impedance R to either one of two values. That is~
amplifier 26 operates with one value of Ro during transmit ~typically 300 ohms) and at the other value during receive operation ~typically 900 ohms). The transmit or reeeive state is determined by transmit detector 18. As discussed more fully hereinafter, the two level impedance technique is u~ilized to reduce the sensitivity of the sidetone signal to different line impedances. That is, return loss during the higher impedance receive state is near perfect during receive but poor during transmit. Accordingly, line length has very little effect on te~minal impedance.
The electronic hybrid function, which separates the receive signal from the composite transmit and receive signal at the telephone line and routes the recei~e signal to the receive amplifier circuit~y, is provided by amplifier 26, attenuator 34 and summing junction 28. The actual signal separation is provided at the summing junction 28. The signal provided at input lead 36 by the output of amplifier 24 is the transmit signal only due to the low output impedance of amplifier 24 and theiisolation of the receive signal provided by amplifier 26. Amplifier 26 amplifies and inverts this transmit signal by a factor of K. The output ~ of amplifier 26, KTX and the receive signal Rx are attenuated by attenuator ; 34 by a factor of l/k. Thus, attenuator 34 reduces the output of amplifier 26 to substantially the same amplitude as the output signal at the output - 10 - "'"' : ~ ' :: . . . - . . , : - ~ ~
~LC)7;~
of amplifier 2~. AccordinglyJ these t~o signals cancel in the summing junction 28 and only an ~tt~nuated receive signal remalns.
Thus, amplifier 2~ and attenuator 34 are called upon to provide transmit signals at the input of summing junction 28 with equal amplitudes and opposite polarities for all line conditions over the ~requencies of interest. Although this function cannot be performed perfectly, some residual transmit signal is nevertheless desirable so as to provide some ; sidetone. Actually, the ideal network would produce optimum sidetone level irrespective or regardless of line conditions. ~s discussed more fully hereinafter, electronic telephone network lO in accordance with the -teachings of the present invention, achieves a degree of independence from line conditions because of the unique design of attenuator 34 and amplifier 26. In one constructed embodiment, attenuator 34 was a resistive divider and RC phase shift network which would optimize to provide maximum cancellation over the audio frequency band for 1800 ohms of 26 gauge line. This is the worst case condition for sidetone reduction since attenuators 22 and 38 function to provide maximum gains in this range. Whereas, however, the reduction in gain provided by attenuators 22 and 38 at very long, or very short, line lengths tends to reduce the respective gains and to thereby simplify the compensation requirements.
Ideally, amplifier 26 should function as a perfect voltage source;
that is, with zera output impedance. In this case, attenuator 3~ could then be adjusted to exactly offset the constant voltage gain of amplifier 26 and perfect cancellation would therefore occur. However, it is known in the art that such ideal characteristics cannot be obtained as the output of amplifier 26 is across the telephone line and otherwise establishes the network impedance; that is, the received signal must necessarily appear, or , .
,~ ;; -- 1 1 --.
~ . , 72~8 ~e developed, across the output impedance of amplifier 26. However, since the terminal impedance for minimum return loss (i.e. impedance matching) is most critical when a signal is being received, this distinction in criticality is advantageously utilized, in accordance with the principles of the present invention, to reduce sensitivity of the sidetone signal to telephone line impedance variations. This is accomplished by virtue of the switching between two impedance levels in the design of amplifier 26.
In curren~ly preferred practice, the output impedance of amplifier 26 is approximately ~00 ohms ~which is the standard conventional or typical telephone line impedance) during receive and standby or idle periods.
Accordingly~ this receive impedance value permits réceive signals to develop normal voltage levels across the network terminals and provides near optimum re~urn loss.
I~ however, the network impedance, or Ro~ were maintained at ~00 ohms during transmit, the voltage gain of amplifier 26 would be a function of telephone line impedance and sidetone levels would vary greatly with different cable lengths. Therefore9 in accordance with a feature of the presen~ invention, the network impedanc0 Ro is switched to approximately 300 ohms during transmit which brings amplifier 26 significantly nearer to the ideal perfect voltage source. It should now be appreciated that an ~O of 300 ohms reduces the effect of telephone line impedance variations on the voltage gain of amplifier 26, and sidetone levels remain substantially constant for different cable lengths. It has ~een ~ouncl that this techniqu reduces sldetone variations relative ~o line impedance by approximately six db. This is particularly desirable during unfavourable impedance conditions such as parallel operation with conventional telephone networks. Further, sidetone changes between receive and transmit conditions hai been found .. , .. , . ; :, . . .. ,. ... ,~ , . :. -~,, - , . - ..... .. ., . ~. .. ..
, ... . . : . ,: : - . , ~ , . ~ : :
,: : : ..... . ... ::: : ., ~.. , . :
to be negli~ible because of the reduced transmit gain during receive operation as provided ~y a~tenuator 16~ That i5, the reduced gain approximately offsets the degraded cancellation a~ summing junction 28 due to the increase of Ro from 300 to its 900 ofim le~el.
The output of summ;ng junction 28 is t~e received signal plus the sidetone signal. Receive equalization attenuator 38, which is functionally and structurally similar to attenuator 22, equalizes the received signal in both ampli~ude and frequenc~ for differen~ cable lengths.
In one constructed embodiment, attenuator 38 was selected to provide subs~antially identical operation to that of att~nuator 22 between 0 and 1800 ohms of telephone line resistance. The loss of attenuator 38 i5 minimum when the resistance of the t~lephone line is approximately 1800 ohms; however, unlike attenuator 22 the loss of attenuator 38 remains at its minimum level as the telephone line resistance~ or effective telephone line length, increases from 1800 ohms, i.e., reducing the DC terminal voltage.
Finally, amplifier 42 provides a fixed gain and amplifies the relatively low receive signal output from attenuator 38 to a level sufficient to drive output transducer 40. A relatively high gain is required in this stage to overcome the losses introduced by attenuator 34 prior to summing junction 2g. As will be discussed more fully hereinafter with reference to the detailed circuitry figures, the DC characteristics of amplifier 42 provide a maximum voltage swing across output transducer 40 for any DC supply potential Vs.
Referring now to Pigure 2, there are showm schematic circuit diagrams of amplifier 14, attenuator 16 and transmit equalization control 22 suitable for use in the electronic telephone network 10 of Figure 1.
:
~; 13 ': :
, . . . .. .... . ..
~7Z698 Amplifier 14 comprises a conv~ntional common emitter ampli~ier stage ~hich in one constructed embodiment provided 24 db gain, ~owever~ the current supply to transistor Ql of amplifier 14 is provided ~y a constant current load transistor Q2. That is, Ql i5 an active constant curren~ load which provides a substantial DC ~ias current to t~e coll~ctor of Ql ~which is necessary for low noise and sufficient gain at low values of Vs~ without loading the AC signal. That is~ ~ransistor Q2 provides a large AC impedance while supplying a substantially high level of current to transistor Ql.
Transistor Q2 is biased in~o operation as a constant current source by current mirror diode Dl. This curren~ mirror diode potential at the base electrode of transistor Q2 is also made available as a current mirror diode signal ~o the active load or constant current source transistor of other circuits of electronic telephone network 10.
The input signal provided by transducer 12 is also coupled to transmit detector 18 by way of amplifier 14. Detector 18 provides a logic "1" at its ou~put for receive or normal conditions; and a logic "0" for the transmit mode. Detector 18 may take any one of a number of suitable circuit configurations but preferably functions as a fast attack/slow decay switching device thereby ~o provide substantially undetectable control of transmit gain and output impedance. Thè output of detector 18 is coupled to output impedance control 20 and to the input or base elsctrode of transistor Q3 of attenuator 16. It can be seen that when detector 18 provides its logic l output transistor Q3 is biased into conduction thereby shunting a portion of the output signal from amplifier 14 to ground through the main electrode of transistor Q3.
Transmit equalization attenuator 22 comprises a two-pole filter/
attenuator that u~ilizes the dynamic resistance of diodes as gain control . "' ' - :
, . ~ ~ , . :
. , : . .. , ,. ... ': . '.: . ;:
. . : . .
~7Z~98 elements. The transmit audi~ signal flo~ is through resistor R101 and resistor R102. Attenuation frequency response shaping is pro~ided by capacitor C101 in com~ination wit~ d~ode D101, and capacltor ClQ2 ;n combina~ion wi~h diode D102, which function to shunt the transmit signal to ground. Diodes D101 and D102 function to provide variable resistance elements.
The dynamic or small-signal resiskance of diode D101 is determined by the DC current flowing from Vs t~rough diodes D103 and D104~ resistor R103 and diode D101 itself. When Vs is near its maximum value9 the resistance of diode D101 is relatively low and the signal loss through capacitor C101 and diode D101 is relativcly large. However, as Vs decreases toward its minimum value, the resistance o diode D101 increases substantially and signal attenuation accordingly decreases. When Vs is approximately 1.15 volts~ diodes D104 and D101 are at or near cut-off and, therefore, there is substantially no signal loss. Thus, overall transmit gain when diodes D104 and D101 are cut-off is maximum.
Capacitor C102 and diode D102 function in a similar manner. However, due to the relatively small size of capacitor C102 this circuit functions to place a "zero" much higher in the audio spectrum than th0 circuit of capacitor C101 and diode D101. Accordingly, the circuit associated with capacitor C102 and diode D102 functions to provide a majority of the frequency compensation of transmit equalization control 22 in accordance with a feature of the present invention. Stated somewhat differently, high-frequency roll-off becomes more pronounced as Vs increases.
Resistors R104 and R105 of transmit equalization control 22 function to increase the voltage drop of diodes D103 and D104 respectively.
This functions to shape the overall characteristics of transmit .
~7Z6913 equaliza~ion 22 so as to equalize or accept #26 gauge cable. The remaining components associ~ted with ~ransmit equalization control 22 of Figure 2 function to reduce transmit gain as Ys drops f.rom approximately 1.1 volts to the minimum operating potential o 0.8 volts. The current flowing through resistor R105 is larger than the maximum current flowing through resistor R106 throughout a majority of the range of operating potentials Vs.
Accordingly, curren~ me~er transistor Q4 remains saturated as it can't source all of the current ~hat is programmed for it by diode D105.
Accordingly, this functions to keep transistors Q5 and Q6 cut-of~ and essentially out of the circuit of transmit equaliza~ion control 22.
However, as Vs drops ~o approximately 1.1 vol~s --noting that transmit gain is maximum at a Vs of approximately 1.15 volts-- the current flowing through diode D105, D106 and R105 is reduoed to a value equal to the maximum current flowing through diode D107 and resistor R106 due to the impending cut-off of diodes D105 and D106. A further reduction of Vs results in current flow through diode D107 and the mirrored current flow from transistors Q5 and Q6 because transistor Q4 can no longer supply the maximum needed current flow through resistor R106. Accordingly, the current from transistors Q5 and Q6 flows into diode D101 and increases attenuation of ~he transmit signal. As previously discussed, this attenuation begins at approximately 1.1 volts and increases as the operating potential of Vs decreases to the minimum value of 0.8 volts. It should now be appreciated that transmi~ equalization control 22 is preerably designed empirically with respect to its attenuation and frequency response characteris~ics to match a given gauge of cable ~such as #26 gauge cable) for equallzation purposes.
Referring now to ~igure 3, there are shown schematic diagrams . ~ ~ . . . . .
- : . , . : . :: . ~
, : , . ~ . : . . ~ . -: . . . . : . ::
- . .. . :. - . . - - ,- :
7~6~
of line-driver amplifier 26, output impedance control 20 and polarity guard 22. The operation of line~driver ampli~ier 26 will be described in conjunction with Figures 4 and $ whic~ respec~ivel~ provide simplified diagrams of the DC and AC equivalent circuits of amplifîer 26.
As previousl~ alluded to, the ac~ievement of maximum transmit dynamic range for operation at long telep~one line lengthsor parallel operation, particularly when the DC terminal voltage is low, requires that the collector-emitter path of t~e OUtp~lt transistor be coupled directly across the line and capable of saturation operation. That is! any serially coupled element for DC line current stabilization, such as an emitter resistor, for example9 reduces the transmit signal output voltage swing.
In Figures 3 and 4, the paralleled output transistors QO are shown as being directly coupled across the telephone line except for the voltage drop induced by polarity guard 32 which is represented by a diode 32' in Figure 4 and the drop of sensing resistor R2. The telephone line in Figure 4 is schematically represented by the exchange battery Be, the terminal impedance of the central exchange Re and the telephone line resistance Rlo As will be described in detail hereinaf~er, a unique polarity guard is utilized, in accordance with the principals of the present invention, to minimize both DC and signal loss. Sensing resistor R2 is used for AC eedback and typically has a very small value and therefore only a negligible efect on the DC characteristics of the overall electronic telephone network. In one constructed embodiment, sensing resistor R2 had a value of 3 ohms.
0utput transistors QO sink a majority o the DC line current.
Accordingly, it is necessary to accurately control this current for overall control of the electronic networks DC characteristics. The paralleled base electrode connections of transistors QO provide both the information .
'' Z69~3 signal and control necessary for accurately establishing collector current as the base-emitter junction voltage is an accurate indicator of collector current flow. In the presen~ invention9 ~his base-emitter voltage is monitored and used in a feed~ack system to determine the base drive curren~
necessary to establish the required collector current. It will be appreciated by those skilled i~ the art t~at accurat~l~ matched output transistors, such as that resulting from integrated circuit fa~rication, are preferable for this design.
Assuming initially that resistor R120 of Figures 3 and 4 is shorted or 0 o}~ns~ the operation of amplifier 26 is described as follows.
Transis~or Q10 and output transistors QO would therefore ha~e the same base-emitter voltages and since transistors Q10 and QO are matched, the collector current in each transistor would be equal. Accordingly, the collector current o~ transistor Q10 would accurately reflect the DC line current flowing through output transistors QO In actual operation, resistor R120 has a finite value and in one constructed embodiment it had a ~alue of approximately 24~0 ohms. Nevertheless, the accurate current sensing function of transistor Q10 is maintained and the otherwise wasted current of transistor Ql~ is substantially reduced. The collector current of transistor Q10~ which is an accurate ~unction of the collector currents of output transistors QO' flows through diode DllO which, in turn, controls current mirror transistor Qll. Thus, the collector current of Qll is also approximately equal to the collector current o~ Q10. The collector current of transistor Qll is converted to a voltage signal by resistor R121.
Accordingly, the voltage developed across resistor R121 is also an accurate indication of the collector current flo~ing through output ~Tansistors ~O.
The collector of transistor Qll is coupled as a first input : . . ~
... ,- . .. . .
, -.. . - - . - : . , . -; . - , ,: . - : :
. ` :- : : : :- . :, . `: . .
~:: : ~ . : . . - : - . :
6~8 ~"-") to an operational ampli~i~r AII~ Resistors R122 and R123 form a voltage divider and the junction of resistors 122 and 123 is coupled to the other input ("~") of amplifier AII b~ way of a buffer amplifier AI.
Amplifier AII functions to force the voltage across resistor R121 to be equal to the voltage provided ~y the voltage divider ~y adjusting the base drive curren~ to output transistors QO Now, since the voltage across resistor R121 is accurately dependent on DC line current, and ~he voltage provided by the voltage divider resistors R122 and R123 is accurately dependant on DC line voltage, amplifier AII establishes an equilibrium condition controlled by the to~al loop resistance that fixes the DC
characteristics of the electronic telephone network. The voltage divider comprising resis~ors R122 and R123 and operational amplifier AI are actually an integral par~ of fixed gain amplifier 24 of Figure 1. }lowever, since ~ they function to control the DC characteristics of amplifier 26 they are : depicted in Figure 4.
In Figure 3, amplifier AII of Figure 4 is comprised of transistors Q13 through Q20. Transistors Q13 and Q14 provide a balance differential input; and transistors Q12 and Q16-18 are current mirrors which operate under the control of resistor RlZ4 and diode ~111. It can be seen that due 20 to the active or constant-current loads, all gain producing transistors of amplifier 26 of Figure 3 are biased in their linear operation region even when the source of operating potential, Vs, is as low as 0.8 volts.
Referring now to Figure 5, there is shown a simplified AC
~ equivalent circuit diagram of amplifier 26 of Figure 3. Since the AC
:~ characteristics of amplifier 26 are affected b~ output impedance control 20~
circuit 20 is also illustrated in Figure 5. Control 20 includes a switching transistor Q301 which is seriall~ coupled with capacitor C301 and resistor "
19 - :
. ~ : , , : . . ., - .:
. '' : ,: '' ~ - . .
, ~L~7Z69~
R301 which components are coupled across resistor R302. Ac~ordingly, transistor Q301 responds to the output of transmit de~ector 1~ to vary the resistance of resistor R302 which is disposed in t~e eed6ack loop of amplifier 26. That is, the normal or receive state of transmit detector 18 is a logic "1" which keeps transistor 301 turned on. This parallels resistor R301 with resistor R302 and results in the network impedance Ro being relatively high. During transmit, the output of detector 18 goes to its logic "0", or "low" state which ~urns off transistor Q301. This essentially removes resistor R301 from the circuit which results in Ro going to its lower impedance value. In one construc~ed embodiment the component values of amplifier 26 were selected such tha~ the normal or receive impedance, Ro, was 900 ohms whereas the transmit state impedance Ro was approximately 300 ohms.
In Figure 3, amplifier AIII of Figure 5 comprises the differential input transistors Q13 and Q14 of Figure 3, which drive the directly coupled common emitter amplifiers Q15, Ql9 and Q20. Q20 drives the parallel base connection of output transistors QO' i.e., the line-driver. Q20 is biased to supply enough drive signal to saturate output transistors QO even when the DC terminal voltage is near its minimum. Translstors Q16-18 are active loads for transistors Q15 and Ql9 and contribute to the very high open loop gain of amplifier 2~. Transistor Q12 provides a current source bias for th0 emitters of the differential input pair, ~13 and Ql~. Capacitor C112 references the non-inverted input of amplifier AIII to ground and, more importantly, it removes all AC feedback from the DC feedback path provided by transistor Qll as discussed wi~h reference to the DC equivalent circuit of Figure 4. As previously alluded to, sensing resistor R2 provides a sampling point for the AC ~eedback of amplifier 26 as illustrated in Figure 5.
: .
: "' .. ....... ,, :,. .. .. ... . . . . ... . . . . . .
: .. ~. . ... : :':' : . - . - , . , :, : -. :' : : . .. :: - .,... : . ::
: . . . . . ~ : : :.: .. . . . , : . ,: ::
1~17~698 Referring again to Figure 3, the opera~ion of polarity guard 32 will be briefly descrihed~ Assuming that conductor 30a is the positive potential side of the line, this condition ~orward ~iases transistor Q401 and diode D401. Transistor Q401 would then receive its ~ase drive current through resistors R401 and R401~ which complete the circui~ to the negative potential conductor 30b. T~us, t~e transistor Q401 can fully saturate such that i~s Vce is on the order of 0.15 volts. Thus, the total voltage drop across the polarity guard is 0.8 volts which represents the 0.15 volt Vce + 0.65 volts ~or one Vd drop). Similarly, when conductor 30b is the positive potential side of the telephone line, transistor Q402 and diode D402 are forward biased. Thus~ polarity guard 32 supplies operating potential of the correct polarity tothe electronic telephone network of the present invention regardless of the polarity of the telephone line. This is important as a practical consideration as in many telephone systems the respective polarities of the various lines are not consistent with one another. Finally, capacitors C401 and C402 function to maintain the DC
drive to the transistors Q401 and Q402 during large transmit signal excursions.
Referring now to Figure 6, there is shown a schematic diagram of summing junction 28 and receive attenuation control 38 in accordance with the principals of the present invention.
It can be seen that resistors R501 and R502 are disposed across the telephone line in a voltage divider network configuration. The voltage tlwsly provided functions to reduce both transmit and receive voltage signals. The ratio of this voltage divider in conjunction with the ratio of resistor R503 and the shaping network, comprising R504~ C501 and C502, is selected such that the transmit voltage from ampliier 24 of Figure 1 offsets or nulls the inverted amplified transmit voltage provided by line-. ..
- . . .. : .. : . : . . : : . -.,,, .. , ... . .~
.: -: . ; . ' . .: , driver 26 at point E. As previousl~ discussed, the received voltage is attenuated but it i5 not nulled by s~ ~ ing junction 28.
The division ratîo of resistors R501 and R502 and ~he impedance of the shaping network comprising capacitors C501, CS02 and resistor R504 Q
were empirically designed to pro~ide optimu~ null across the audio spectrum for 21.5 K feet of #26 gauge ca~le. ~t has also ~een found that this selection provides good performance not onl~ wit~ 21.5 K feet of #26 gauge cable, but als~ for all practical operation conditions. Accordingly~
summing junction 28, in conjunction with the equalization and output impedance control, in accordance with the present invention, provides optimum sidetone control for all practical operating conditions --including parallel operation with conventional telephone networks.
Referring now to the receive equali~ation attenuator 38 of Figure 6, it can be seen that attenuator 38 is also a two-pole filter attenuator similar in function and structure to attenuator 22 of ~igures 1 and 2.
Accordingly, the operation of attenuator 38 need no~ be described in great detail herein. However, it can be seen that both receive and sidetone signals flow through resistors R601 and R602. ~urther~ capacitor C601 in combination with diode D601, and capacitor C502 in combination with diode D602 provide the control of attenuation and frequency response characteristics, in response to the value of the derived operating potential Vs~
It should be noted however, with reference to attenuator 38 of Figure 6, that the dynamic resistance of a diode is most useful as a variable resistance when the applied AC signal level is kept at a relatively low value so as to prevent excessive distortion. As a practical matter, prevention of load distortion necessitates that the AC signal not ~ ' ;~
.. , . ; ~ , : . . --- ~7~26~3 exceed approximately lO or 12 milliYolts. For this reason, relatively small signal levels are applied to and derived from summing junction 28 and relatively large gains are provided by the receive amplifier 42 of Figure 1, in accordance with anot~er feature of the present invention.
~ hat has been taught, thenJ is an electronic telephone network facilitating, notably, automatic telephone line equalization, substan~ially reduced sensitivit~ of the associated sidetone signal, and operation at very low terminal voltage levels including parallel operation at low voltage operations~ The form of the invention illustrated and described herein is but a preferred embodiment of these teachings, in the form currently preferred for manufacture. It is shown as an illustration of the inventive concepts, however, rather than by way of limitation, and it is pointed out that various modifications, and alterations may be indulged in within the scope of the appended claims.
,: , , . - . . : :
Claims (13)
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:
1. An amplifier adapted for coupling telephone signals to a telephone line and for controlling the DC current flowing through said telephone line to a predetermined value at a given telephone line loop resistance, said ampli-fier comprising; at least one output transistor having first and second main electrodes coupled across said telephone line, and having a control electrode;
means for applying said telephone signals to said control electrode; means coupled between said control electrode and one of said main electrodes for deriving a first signal related to the DC current flowing through said main electrodes; means coupled across said telephone line for deriving a second signal related to the DC voltage appearing across said telephone line; means for comparing said first and second signals to provide a third signal indicative of the difference between said first and second signals;
and means coupled between said control electrode and the comparing means, and responsive to said third signal for adjusting the value of said DC
current flowing through said main electrodes to said predetermined value.
means for applying said telephone signals to said control electrode; means coupled between said control electrode and one of said main electrodes for deriving a first signal related to the DC current flowing through said main electrodes; means coupled across said telephone line for deriving a second signal related to the DC voltage appearing across said telephone line; means for comparing said first and second signals to provide a third signal indicative of the difference between said first and second signals;
and means coupled between said control electrode and the comparing means, and responsive to said third signal for adjusting the value of said DC
current flowing through said main electrodes to said predetermined value.
2. The amplifier according to claim 1, wherein said main electrodes comprise a collector and an emitter electrode, wherein said control electrode comprises a base electrode and wherein said means for deriving said first signal is coupled between said base and emitter electrodes.
3. The amplifier according to claim 2, wherein said first signal is proportional to the DC current flowing through said telephone line.
4. The amplifier according to claim 3, wherein said first signal is the base-emitter DC voltage of said transistor.
5. The amplifier according to claim 1, wherein a plurality of transistors, having their respective electrodes parallel connected, are provided.
6. The amplifier according to claim 5, wherein said transistors have substantially matched DC operating characteristics.
7. The amplifier according to claim 1, wherein said means for comparing said first and second signals comprises a balanced differential input amplifier including a first, second and third transistors, said first transistor providing an active constant current source for said second and third transistors, said second and third transistors having input electrodes respectively coupled to one of said first and second signals, and one of said first and second transistors having its output electrode coupled to said control electrode of said output transistor.
8. The amplifier according to claim 1, wherein said means for applying said telephone signals incorporates said means for comparing said first and second signals.
9. An amplifier adapted for coupling telephone signals to a telephone line while controlling the DC line current flowing through said telephone line to a predetermined value at a given telephone line loop resistance, said amplifier comprising: at least one output transistor having first and second main electrodes coupled across said telephone line, and having a control electrode; means for applying said telephone signals to said control electrode; first means coupled between said control electrode and one of said main electrodes for deriving a first signal related to the DC
line current flowing through said main electrodes; second means for providing a reference signal related to said given telephone line loop resistance;
and means coupled to said first and second means and to said control electrode, and responsive to said first signal and said reference signal for adjusting said DC line current to said predetermined value.
line current flowing through said main electrodes; second means for providing a reference signal related to said given telephone line loop resistance;
and means coupled to said first and second means and to said control electrode, and responsive to said first signal and said reference signal for adjusting said DC line current to said predetermined value.
10. The amplifier according to claim 9, wherein said first signal is derived from the base-emitter voltage of said output transistor.
11. The amplifier according to claim 10, wherein said reference signal is provided by a voltage divider coupled across said telephone line.
12. The amplifier according to claim 9, wherein said output transistor is modulated by said telephone signals over a range of conductivity values including its cut-off and saturated values.
13. The ampifier according to claim 9, wherein said output transistor includes a feedback loop between one of said main electrodes and said control electrode, said feedback loop including means responsive to said telephone signals for translating the AC output impedance of said amplifer between two levels wherein said output impedance is decreased when said amplifier is applying said telephone signals to said telephone line.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA324,419A CA1072698A (en) | 1975-07-08 | 1979-03-29 | Electronic telephone network |
Applications Claiming Priority (3)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US05/594,026 US4002852A (en) | 1975-07-08 | 1975-07-08 | Electronic telephone network |
| CA256,536A CA1061491A (en) | 1975-07-08 | 1976-07-07 | Electronic telephone network |
| CA324,419A CA1072698A (en) | 1975-07-08 | 1979-03-29 | Electronic telephone network |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1072698A true CA1072698A (en) | 1980-02-26 |
Family
ID=27164549
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA324,419A Expired CA1072698A (en) | 1975-07-08 | 1979-03-29 | Electronic telephone network |
Country Status (1)
| Country | Link |
|---|---|
| CA (1) | CA1072698A (en) |
-
1979
- 1979-03-29 CA CA324,419A patent/CA1072698A/en not_active Expired
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