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Search Results (401)

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Keywords = millimeter-wave design

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12 pages, 1744 KiB  
Article
DGRO: Doppler Velocity and Gyroscope-Aided Radar Odometry
by Chao Guo, Bangguo Wei, Bin Lan, Lunfei Liang and Houde Liu
Sensors 2024, 24(20), 6559; https://doi.org/10.3390/s24206559 - 11 Oct 2024
Abstract
A stable and robust odometry system is essential for autonomous robot navigation. The 4D millimeter-wave radar, known for its resilience in harsh weather conditions, has attracted considerable attention. As the latest generation of FMCW radar, 4D millimeter-wave radar provides point clouds with both [...] Read more.
A stable and robust odometry system is essential for autonomous robot navigation. The 4D millimeter-wave radar, known for its resilience in harsh weather conditions, has attracted considerable attention. As the latest generation of FMCW radar, 4D millimeter-wave radar provides point clouds with both position and Doppler velocity information. However, the increased uncertainty and noise in 4D radar point clouds pose challenges that prevent the direct application of LiDAR-based SLAM algorithms. To address this, we propose a SLAM framework that fuses 4D radar data with gyroscope readings using graph optimization techniques. Initially, Doppler velocity is employed to estimate the radar’s ego velocity, with dynamic points being removed accordingly. Building on this, we introduce a pre-integration factor that combines ego-velocity and gyroscope data. Additionally, leveraging the stable RCS characteristics of radar, we design a corresponding point selection method based on normal direction and propose a scan-to-submap point cloud registration technique weighted by RCS intensity. Finally, we validate the reliability and localization accuracy of our framework using both our own dataset and the NTU dataset. Experimental results show that the proposed DGRO system outperforms traditional 4D radar odometry methods, especially in environments with slow speeds and fewer dynamic objects. Full article
(This article belongs to the Section Radar Sensors)
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<p>Overview of the proposed DGRO system.</p>
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<p>Results of registering two radar point clouds using RCS-ICP. (<b>a</b>) Initial point clouds, (<b>b</b>) Results after registration.</p>
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<p>The ground robot and the sensors for tests. Note that the realsense camera and the RGBD camera are not used.</p>
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<p>Mapping results by DGRO on data of a large-scale industrial park.</p>
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<p>Comparison of accuracy and computational time for different registration algorithms.</p>
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19 pages, 17495 KiB  
Article
Study on the Design Method of High-Resolution Volume-Phase Holographic Gratings
by Shuo Wang, Lei Dai, Chao Lin, Long Wang, Zhenhua Ji, Yang Fu, Quyouyang Gao and Yuquan Zheng
Sensors 2024, 24(19), 6493; https://doi.org/10.3390/s24196493 - 9 Oct 2024
Abstract
Volume-phase holographic gratings are suitable for use in greenhouse gas detection imaging spectrometers, enabling the detection instruments to achieve high spectral resolution, high signal-to-noise ratios, and high operational efficiency. However, when utilized in the infrared wavelength band with high dispersion requirements, gratings struggle [...] Read more.
Volume-phase holographic gratings are suitable for use in greenhouse gas detection imaging spectrometers, enabling the detection instruments to achieve high spectral resolution, high signal-to-noise ratios, and high operational efficiency. However, when utilized in the infrared wavelength band with high dispersion requirements, gratings struggle to meet the demands for low polarization sensitivity due to changes in diffraction performance caused by phase delays in the incidence of light waves with distinct polarization states, and current methods for designing bulk-phase holographic gratings require a large number of calculations that complicate the balance of diffraction properties. To overcome this problem, a design method for transmissive bulk-phase holographic gratings is proposed in this study. The proposed method combines two diffraction theories (namely, Kogelnik coupled-wave theory and rigorous coupled-wave theory) and establishes a parameter optimization sequence based on the influence of design parameters on diffraction characteristics. Kogelnik coupled-wave theory is employed to establish the initial Bragg angle range, ensuring that the diffraction efficiency and phase delay of the grating thickness curve meet the requirements for incident light waves in various polarization states. Utilizing rigorous coupled-wave theory, we optimize grating settings based on criteria such as a center wavelength diffraction efficiency greater than 95%, polarization sensitivity less than 10%, maximum bandwidth, and spectral diffraction efficiency exceeding 80%. The ideal grating parameters are ultimately determined, and the manufacturing tolerances for various grating parameters are analyzed. The design results show that the grating stripe frequency is 1067 lines per millimeter, and the diffraction efficiencies of TE and TM waves are 96% and 99.89%, respectively. The diffraction efficiency of unpolarized light is more than 88% over the whole spectral range with an average efficiency of 94.49%, an effective bandwidth of 32 nm, and a polarization sensitivity of less than 7%. These characteristics meet the performance requirements for dispersive elements based on greenhouse gas detection, the spectral resolution of the detection instrument is up to 0.1 nm, and the signal-to-noise ratio and working efficiency are improved by increasing the transmittance of the instrument. Full article
(This article belongs to the Section Optical Sensors)
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Figure 1
<p>(<b>a</b>) Structure of the VPHG; (<b>b</b>) transmissive VPHG recording; (<b>c</b>) reflective VPHG recording.</p>
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<p>Bragg diffraction principle. (Blue arrows represent the incident and diffracted light, and black arrows represent the grating vector).</p>
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<p>K-vector circle: (<b>a</b>) grating recording; (<b>b</b>) grating Bragg diffraction and diffraction deviating from Bragg conditions; (<b>c</b>) wavelength-shifted reconstruction principle.</p>
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<p>(<b>a</b>) Diagram of the principle of diffraction theory calculation; (<b>b</b>) types of refractive index modulation within the grating.</p>
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<p>Comparison of the two theories for different grating periods.</p>
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<p>Comparison of the two theories for different grating thicknesses.</p>
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<p>Comparison of the two theories for different refractive index modulations.</p>
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<p>Comparison of the diffraction performance of the two types of gratings: (<b>a</b>) wavelength selectivity curves for different polarization states of the transmission-type VPHG; (<b>b</b>) angular selectivity curves for different polarization states of the transmission-type VPHG; (<b>c</b>) wavelength selectivity curves for different polarization states of the reflection-type VPHG; (<b>d</b>) angular selectivity curves for different polarization states of the reflection-type VPHG.</p>
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<p>(<b>a</b>) The relationship between diffraction efficiency and grating thickness for different polarization states when Δ<span class="html-italic">n</span> = 0.045. (<b>b</b>) The relationship between diffraction efficiency and grating thickness for different polarization states when Δ<span class="html-italic">n</span> = 0.05. (<b>c</b>) The relationship between diffraction efficiency and refractive index modulation for different polarization states when <span class="html-italic">d</span> = 25 μm. (<b>d</b>) The relationship between diffraction efficiency and refractive index modulation for different polarization states when <span class="html-italic">d</span> = 30 μm.</p>
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<p>(<b>a</b>) The effect of d and Δ<span class="html-italic">n</span> on diffraction efficiency for TE wave incidence. (<b>b</b>) The effect of d and Δ<span class="html-italic">n</span> on diffraction efficiency for TM wave incidence.</p>
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<p>(<b>a</b>) Wavelength selectivity curves for different values of Δ<span class="html-italic">n</span> and <span class="html-italic">d</span> when Δ<span class="html-italic">n·d</span> is constant. (<b>b</b>) Angular selectivity curves for different values of Δ<span class="html-italic">n</span> and <span class="html-italic">d</span> when Δ<span class="html-italic">n·d</span> is constant.</p>
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<p>Thickness and diffraction efficiency curves for different periods and Bragg angles: (<b>a</b>) Λ = 1.0757 μm, <span class="html-italic">θ<sub>B</sub></span> = 30°; (<b>b</b>) Λ = 1.0141 μm, <span class="html-italic">θ<sub>B</sub></span> = 32.03°; (<b>c</b>) Λ = 0.9377 μm, <span class="html-italic">θ<sub>B</sub></span> = 35°; (<b>d</b>) Λ = 0.8367 μm, <span class="html-italic">θ<sub>B</sub></span> = 40°.</p>
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<p>Four scenarios where the diffraction efficiency at the intersection points of the thickness and diffraction efficiency curves for different polarization states is equal to 0.95.</p>
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<p>Analysis of the designed VPHG diffraction’s performance: (<b>a</b>) wavelength selectivity curves for different polarization states; (<b>b</b>) angular selectivity curves for different polarization states; (<b>c</b>) polarization sensitivity; (The red line indicates the value of polarization sensitivity.) (<b>d</b>) wavelength selectivity curve for non-polarized light. (The red line indicates the value of diffraction efficiency of unpolarized light).</p>
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<p>Diffraction performance analysis of VPHG designed using Kogelnik theory: (<b>a</b>) wavelength selectivity curves for different polarization states; (<b>b</b>) angular selectivity curves for different polarization states; (<b>c</b>) polarization sensitivity; (The red line indicates the value of polarization sensitivity). (<b>d</b>) wavelength selectivity curve for non-polarized light. (The red line indicates the value of diffraction efficiency of unpolarized light).</p>
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<p>Diffraction performance analysis of VPHG designed using RCWA theory: (<b>a</b>) wavelength selectivity curves for different polarization states; (<b>b</b>) angular selectivity curves for different polarization states; (<b>c</b>) polarization sensitivity; (The red line indicates the value of polarization sensitivity). (<b>d</b>) wavelength selectivity curve for non-polarized light. (The red line indicates the value of diffraction efficiency of unpolarized light).</p>
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<p>Results of tolerance analysis for different parameters of VPHG: (<b>a</b>) Analysis results of grating period accuracy; (<b>b</b>) Bragg Angle accuracy analysis results; (<b>c</b>) Accuracy analysis results of grating thickness; (<b>d</b>) Accuracy analysis results of the index modulation system.</p>
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15 pages, 8083 KiB  
Article
Advanced Metamaterial-Integrated Dipole Array Antenna for Enhanced Gain in 5G Millimeter-Wave Bands
by Domin Choi, Md Abu Sufian, Jaemin Lee, Wahaj Abbas Awan, Young Choi and Nam Kim
Appl. Sci. 2024, 14(19), 9138; https://doi.org/10.3390/app14199138 - 9 Oct 2024
Abstract
A metamaterial-based non-uniform dipole array antenna is presented for high gain 5G millimeter-wave applications with a wideband characteristic. Initially, a non-uniform dipole array is designed on a 0.202 mm thick Rogers RO4003C substrate, offering a wide operating bandwidth ranging from 23.1 GHz to [...] Read more.
A metamaterial-based non-uniform dipole array antenna is presented for high gain 5G millimeter-wave applications with a wideband characteristic. Initially, a non-uniform dipole array is designed on a 0.202 mm thick Rogers RO4003C substrate, offering a wide operating bandwidth ranging from 23.1 GHz to 44.8 GHz. The dipole array antenna emits unidirectional end-fire radiation with a maximum gain of 8.1 dBi and an average gain of 6.7 dBi. Subsequently, to achieve high gain performance, a 5 × 7 metamaterial structure is designed in the direction of the antenna radiation. The implemented metamaterial structure is optimized for the operating frequency, enhancing the directivity of the antenna radiation and resulting in a gain increment of more than 3 dBi compared to the dipole array alone. The developed metamaterial-integrated dipole array antenna offers an operating bandwidth (S11 < −10 dB) of more than 21 GHz (63.92%), ranging from 23.1 GHz to 44.8 GHz, covering the most commonly used 5G millimeter-wave frequency bands (n257, n258, n259, n260, and n261). Furthermore, the presented antenna yields a stable high gain with a peak gain of 11.21 dBi and a good radiation efficiency of more than 64%. The proposed antenna is an excellent option for millimeter-wave 5G systems due to its overall properties, particularly its high gain and end-fire radiation characteristics, combined with a wide operating bandwidth. Full article
(This article belongs to the Section Electrical, Electronics and Communications Engineering)
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<p>Representation of mm wave device connection in 5G-NR environment.</p>
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<p>The dipole array antenna: (<b>a</b>) a view of the individual sides and (<b>b</b>) a partially transparent view of the substrate.</p>
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<p>The performance characteristics of the dipole array antenna: (<b>a</b>) the reflection coefficient, and (<b>b</b>) the gain and 3D directivity at 28 GHz and 38 GHz.</p>
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<p>The surface current distribution of the dipole array at different operating frequencies.</p>
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<p>Parametric analysis of the dipole array antenna for different parameters: (<b>a</b>) for <span class="html-italic">P</span><sub>1</sub> and (<b>b</b>) for <span class="html-italic">P</span><sub>4</sub>.</p>
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<p>The properties of the metamaterial unit cell: (<b>a</b>) a schematic diagram with the design parameters, (<b>b</b>) the gain enhancement mechanism, (<b>c</b>) the characteristic response, and (<b>d</b>) the material properties.</p>
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<p>The geometry and design parameters of the proposed metamaterial-integrated dipole array antenna.</p>
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<p>The gain response of the dipole antenna varies with different numbers of metamaterial cells (<b>a</b>) horizontally and (<b>b</b>) vertically.</p>
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<p>Photographs of the presented metamaterial-integrated dipole array antenna: (<b>a</b>) the fabricated prototype and (<b>b</b>) the radiation characterization measurement setup.</p>
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<p>Photographs of the presented metamaterial-integrated dipole array antenna: (<b>a</b>) the fabricated prototype and (<b>b</b>) the radiation characterization measurement setup.</p>
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<p>The reflection coefficient of the metamaterial-integrated dipole array antenna.</p>
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<p>The radiation pattern of the developed metamaterial-integrated dipole array antenna: (<b>a</b>) simulated 3D directivity at different frequencies and (<b>b</b>) 2D polar radiation pattern with gain scale.</p>
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<p>Simulated and measured gain, and radiation efficiency performance: (<b>a</b>) antenna without metamaterial structure and (<b>b</b>) antenna with metamaterial structure.</p>
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<p>The 2-port MIMO configuration of the proposed metamaterial-integrated non-uniform dipole array.</p>
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<p>The results of the MIMO configuration of the proposed antenna: (<b>a</b>) S-parameters and (<b>b</b>) the ECC and DG.</p>
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23 pages, 8425 KiB  
Article
Optical Design Study with Uniform Field of View Regardless of Sensor Size for Terahertz System Applications
by Jungjin Park, Jaemyung Ryu and Hojong Choi
Appl. Sci. 2024, 14(19), 9097; https://doi.org/10.3390/app14199097 - 8 Oct 2024
Abstract
The focal length in a typical optical system changes with the angle of view, according to the size of the sensor. This study proposed an optical terahertz (THz) system application where the focal length changed while the angle of view was fixed; thus, [...] Read more.
The focal length in a typical optical system changes with the angle of view, according to the size of the sensor. This study proposed an optical terahertz (THz) system application where the focal length changed while the angle of view was fixed; thus, the image height was variable and responded to various sensor sizes. Therefore, it is possible to respond to various sensors with one optical system when the inspection distance is fixed. The fundamental optical system was designed by arranging the refractive power, which was determined according to the sensor size using the Gaussian bracketing method. A zoom optical system that changed the image height by fixing the angle of view and changed the focal length by moving the internal lens group was designed. THz waves exhibit minimal change in the refractive index depending on the wavelength. Moreover, their long-wavelength characteristics facilitate the development of millimeter-level pixel sizes. Therefore, the root mean square size of the maximum spot was 0.329 mm, which corrected the aberration to less than 1 mm (smaller than the pixel size). Further, a lighting analysis at 3 and 6 m locations confirmed the expansion of the lighting area by the magnification of the sensor size. After turning off certain light sources, we checked the contrast ratio via lighting analysis and confirmed that the size of one pixel was clearly distinguishable. Consequently, this newly designed optical system performed appropriately as an optical inspection system for THz system applications. Full article
(This article belongs to the Collection Optical Design and Engineering)
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<p>Relationship between focal length and angle of view in (<b>a</b>) general optical system and (<b>b</b>) optical system for THz applications.</p>
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<p>Relationship between focal length and field of view.</p>
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<p>The angle of view is according to the focal length.</p>
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<p>Thick single-lens optical layout at finite point.</p>
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<p>The fundamental layout of the zoom optical system is based on the placement of refractive power.</p>
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<p>Spot diagrams at (<b>a</b>) wide, (<b>b</b>) middle, and (<b>c</b>) tele ends.</p>
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<p>Optical path diagram for height-variable zoom lens.</p>
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<p>Zoom locus of height-variable zoom lens.</p>
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<p>Distortion grid at object distance infinity at (<b>a</b>) wide, (<b>b</b>) middle, and (<b>c</b>) tele ends.</p>
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<p>Distortion grid at an object distance of 3 m at (<b>a</b>) wide, (<b>b</b>) middle, and (<b>c</b>) tele ends.</p>
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<p>Distortion grid at an object distance of 6 m at (<b>a</b>) wide, (<b>b</b>) middle, and (<b>c</b>) tele ends.</p>
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<p>(<b>a</b>) Spot diagrams and (<b>b</b>) diffraction MTF at infinite object distance for the wide end, (<b>c</b>) spot diagrams and (<b>d</b>) diffraction MTF at infinite object distance for the middle end, and (<b>e</b>) spot diagrams and (<b>f</b>) diffraction MTF at infinite object distance for the telephoto end.</p>
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<p>(<b>a</b>) Spot diagrams and (<b>b</b>) diffraction MTF at infinite object distance for the wide end, (<b>c</b>) spot diagrams and (<b>d</b>) diffraction MTF at infinite object distance for the middle end, and (<b>e</b>) spot diagrams and (<b>f</b>) diffraction MTF at infinite object distance for the telephoto end.</p>
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<p>(<b>a</b>) Spot diagrams and (<b>b</b>) diffraction MTF at an object distance of 3 m for the wide end, (<b>c</b>) spot diagrams and (<b>d</b>) diffraction MTF at an object distance of 3 m for the middle end, and (<b>e</b>) spot diagrams and (<b>f</b>) diffraction MTF at an object distance of 3 m for the telephoto end.</p>
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<p>(<b>a</b>) Spot diagrams and (<b>b</b>) diffraction MTF at an object distance of 3 m for the wide end, (<b>c</b>) spot diagrams and (<b>d</b>) diffraction MTF at an object distance of 3 m for the middle end, and (<b>e</b>) spot diagrams and (<b>f</b>) diffraction MTF at an object distance of 3 m for the telephoto end.</p>
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<p>(<b>a</b>) Spot diagrams and (<b>b</b>) diffraction MTF at an object distance of 6 m for the wide end, (<b>c</b>) spot diagrams and (<b>d</b>) diffraction MTF at an object distance of 6 m for the middle end, and (<b>e</b>) spot diagrams and (<b>f</b>) diffraction MTF at an object distance of 6 m for the telephoto end.</p>
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<p>Illuminance analysis system of height-variable zoom lens.</p>
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<p>Illuminance analysis results of receiver enlarged in the (<b>a</b>) center of the wide angle, (<b>b</b>) around the wide angle, (<b>c</b>) center of the middle end, (<b>d</b>) around the middle end, (<b>e</b>) center of the telescope, and (<b>f</b>) around the telescope when the object distance is 3 m.</p>
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<p>Illuminance analysis results of the receiver enlarged in the (<b>a</b>) center of the wide angle, (<b>b</b>) around the wide angle, (<b>c</b>) center of the middle end, (<b>d</b>) around the middle end, (<b>e</b>) center of the telescope, and (<b>f</b>) around the telescope when the object distance is 6 m.</p>
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7 pages, 3998 KiB  
Article
Development and Characterization of an Advanced Voltage-Controllable Capacitor Based on AlInGaN/GaN-Si (111) Epitaxy
by He Guan and Guiyu Shen
Coatings 2024, 14(10), 1254; https://doi.org/10.3390/coatings14101254 - 1 Oct 2024
Abstract
The AlInGaN/GaN heterojunction epitaxy material with high cutoff frequency and saturated power density has become a very popular candidate for millimeter wave applications in next-generation mobile communication. In this study, an advanced voltage-controllable capacitor based on the AlInGaN/GaN-Si (111) epitaxy was proposed by [...] Read more.
The AlInGaN/GaN heterojunction epitaxy material with high cutoff frequency and saturated power density has become a very popular candidate for millimeter wave applications in next-generation mobile communication. In this study, an advanced voltage-controllable capacitor based on the AlInGaN/GaN-Si (111) epitaxy was proposed by employing a bi-directional series MIS capacitor structure. The capacitor was fabricated by using a pad area of 40 μm × 40 μm, with a 1 μm distance between the positive and negative electrodes. The test results show that the capacitance is turned on with a saturation capacitance density and a maximum leakage current density of 0.30 fF/μm2 of 0.37 pA/μm2, respectively, for the control voltage from −6.5 V to 6 V. In particular, in the proposed design method, the saturation capacitance required for the practical application can be obtained by simply adjusting the capacitance area. The capacitor showcases characteristics of rapid turn-on and turn-off responses coupled with low loss, underscoring its promising prospects for deployment in RF switching applications. Full article
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<p>The InAlGaN/GaN-Si epitaxy schematic and test result; (<b>a</b>) is the schematic of the AlInGaN/GaN epitaxy structure, (<b>b</b>) is the TEM analysis of AlInGaN/GaN heterojunction, (<b>c</b>) is the TEM analysis of AlN/AlGaN buffer layer, and (<b>d</b>) is the EDS Scan result of Al.</p>
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<p>The InAlGaN/GaN-Si epitaxy schematic and test result; (<b>a</b>) is the schematic of the AlInGaN/GaN epitaxy structure, (<b>b</b>) is the TEM analysis of AlInGaN/GaN heterojunction, (<b>c</b>) is the TEM analysis of AlN/AlGaN buffer layer, and (<b>d</b>) is the EDS Scan result of Al.</p>
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<p>The schematic of the device structure and the equivalent circuit of the device.</p>
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<p>The optical microscopy.</p>
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<p>The C-V test curves of the capacitor.</p>
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<p>The I-V test curves of the capacitor.</p>
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22 pages, 4902 KiB  
Review
A Review of Microstrip Patch Antenna-Based Passive Sensors
by Zain Ul Islam, Amine Bermak and Bo Wang
Sensors 2024, 24(19), 6355; https://doi.org/10.3390/s24196355 - 30 Sep 2024
Abstract
This paper briefly overviews and discusses the existing techniques using antennas for passive sensing, starting from the antenna operating principle and antenna structural design to different antenna-based sensing mechanisms. The effects of different electrical properties of the material used to design an antenna, [...] Read more.
This paper briefly overviews and discusses the existing techniques using antennas for passive sensing, starting from the antenna operating principle and antenna structural design to different antenna-based sensing mechanisms. The effects of different electrical properties of the material used to design an antenna, such as conductivity, loss tangent, and resistivity, are discussed to illustrate the fundamental sensing mechanisms. Furthermore, the key parameters, such as operating frequency and antenna impedance, along with the factors affecting the sensing performance, are discussed. Overall, passive sensing using an antenna is mainly achieved by altering the reflected wave characteristics in terms of center frequency, return loss, phase, and received/reflected signal strength. The advantages and drawbacks of each technique are also discussed briefly. Given the increasing relevance, millimeter-wave antenna sensors and resonator sensors are also discussed with their applications and recent advancements. This paper primarily focuses on microstrip-based radiating structures and insights for further sensing performance improvement using passive antennas, which are outlined in this study. In addition, suggestions are made for the current scientific and technical challenges, and future directions are discussed. Full article
(This article belongs to the Special Issue Feature Review Papers in Physical Sensors)
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<p>Illustration of an antenna-based sensing system.</p>
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<p>Microstrip patch antenna: (<b>a</b>) perspective view, (<b>b</b>) top view, and (<b>c</b>) side view [<a href="#B45-sensors-24-06355" class="html-bibr">45</a>].</p>
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<p>Illustration of antenna-based sensing mechanism.</p>
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<p>(<b>a</b>) Sensing based on frequency shift with respect to salinity level and temperature, (<b>b</b>) fabricated prototype and experimental set up [<a href="#B17-sensors-24-06355" class="html-bibr">17</a>].</p>
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<p>(<b>a</b>) Measured sugar concentration sensing based on Return Loss variation, (<b>b</b>) experimental setup [<a href="#B19-sensors-24-06355" class="html-bibr">19</a>].</p>
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<p>Sensing based on phase variation [<a href="#B77-sensors-24-06355" class="html-bibr">77</a>].</p>
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<p>(<b>a</b>) Measured frequency shift and return loss variation response for glucose concentration sensing (<b>b</b>) experimental setup [<a href="#B14-sensors-24-06355" class="html-bibr">14</a>].</p>
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<p>RSSI-based sensing mechanism [<a href="#B86-sensors-24-06355" class="html-bibr">86</a>].</p>
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<p>RSSI-based sensing mechanism (<b>a</b>) measured frequency response (<b>b</b>) experimental setup [<a href="#B23-sensors-24-06355" class="html-bibr">23</a>].</p>
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26 pages, 8051 KiB  
Article
Artificial Intelligence for the Evaluation of Postures Using Radar Technology: A Case Study
by Davide De Vittorio, Antonio Barili, Giovanni Danese and Elisa Marenzi
Sensors 2024, 24(19), 6208; https://doi.org/10.3390/s24196208 - 25 Sep 2024
Abstract
In the last few decades, major progress has been made in the medical field; in particular, new treatments and advanced health technologies allow for considerable improvements in life expectancy and, more broadly, in quality of life. As a consequence, the number of elderly [...] Read more.
In the last few decades, major progress has been made in the medical field; in particular, new treatments and advanced health technologies allow for considerable improvements in life expectancy and, more broadly, in quality of life. As a consequence, the number of elderly people is expected to increase in the following years. This trend, along with the need to improve the independence of frail people, has led to the development of unobtrusive solutions to monitor daily activities and provide feedback in case of risky situations and falls. Monitoring devices based on radar sensors represent a possible approach to tackle postural analysis while preserving the person’s privacy and are especially useful in domestic environments. This work presents an innovative solution that combines millimeter-wave radar technology with artificial intelligence (AI) to detect different types of postures: a series of algorithms and neural network methodologies are evaluated using experimental acquisitions with healthy subjects. All methods produce very good results according to the main parameters evaluating performance; the long short-term memory (LSTM) and GRU show the most consistent results while, at the same time, maintaining reduced computational complexity, thus providing a very good candidate to be implemented in a dedicated embedded system designed to monitor postures. Full article
(This article belongs to the Section Radar Sensors)
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<p>Visual output: the red circle represents the sensing device and shows its positioning in the volume under monitoring; the yellow and orange circles are the spots indicating that two people are in the room, while the blue small circles form the point clouds. In this image, only the person on the right (orange spot) has a large and well-defined point cloud, while the person identified with the yellow spot has only a few points in the cloud.</p>
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<p>RNN principle of functioning and architecture of the LSTM cell.</p>
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<p>Pseudo-code for LSTM, Bi-LSTM, projected LSTM and GRU, in the case of the subdivision between the training and test sets (<b>a</b>) and for the leave-one-out approach (<b>b</b>). Line 6 differs in the two cases, since, in (<b>a</b>), there is the subdivision into training and test sets with all ratios previously mentioned; in (<b>b</b>), instead, it considers the single subject left out from the training, following the leave-one-out approach. Line 8, in addition, has been written in a generalized way since it depends on the DL approach considered (written in italics, i.e., LSTM).</p>
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<p>Room where the first experimental tests were performed, shown from different angles. The device can be seen in the top-right corner of the third image, on the right of the page (highlighted by the red circle).</p>
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<p>Second room, where all other experimental tests were performed, shown from different angles. The device can be seen in the top-right corner of the second image, on the right of the page, stuck to the wall over the door (highlighted by the red circle). Since, here, there was more room for movement, walking, sitting and falling tests were conducted.</p>
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<p>The figure shows a person randomly walking in the room. The graph shows the 3 spatial coordinates (x in red, y in green and z in blue), with their maximum (red circle), minimum (blue circle) and mean values (red cross). As can be seen, the z coordinate is reduced when the subject moves closer to the sensor, as shown by the other two coordinates, x and y, having smaller values as well.</p>
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<p>The image on the left displays a distortion in the point cloud and also a double spot, which could be mistaken as indicating two people in the room. The schematic body reconstruction clearly highlights that, without prior knowledge of the measurement context, the situation could be easily wrongly interpreted. The three-dimensional representation on the right shows that, even with no one in the room, metallic furniture produces reflections, resulting in an actual (albeit small) point cloud.</p>
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<p>The figures show the speed on the two axes, x and y, related to a person randomly walking in the room in (<b>a</b>,<b>b</b>), and falling in (<b>c</b>,<b>d</b>). The crosses always represent the mean value of the corresponding curve. In (<b>c</b>), the position along the three axes is reported, and, in (<b>d</b>), the speed of the fall is observed. In this case, the legend of colors and indicators is the same as in <a href="#sensors-24-06208-f006" class="html-fig">Figure 6</a>. Compared to walking, a fall presents a very rapid increase in speed, followed by a prolonged stop.</p>
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<p>(<b>a</b>,<b>b</b>) show the same test as presented in <a href="#sensors-24-06208-f008" class="html-fig">Figure 8</a>c,d, while (<b>c</b>) presents another experiment of a person falling. The crosses always represent the mean value of the corresponding curve. The movement graphs are associated with the corresponding spot and number of points in the cloud for each frame. The device works by collecting 10 frames/s. In both cases, the evolution is very similar, as can be derived in (<b>b</b>,<b>c</b>), respectively. Here, the person is identified by the system after a short transient period according to the blue line. The number of points in the cloud is given at any frame by the orange graph and clearly shows that, after the person is detected, the number decreases, and it is considerably reduced when the fall occurs, potentially causing problems in reconstructing the point cloud.</p>
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<p>(<b>a</b>) is the graphical representation of two classes of output, standing and falling, where the colored circles are those from the training set, while the others are the detected ones. (<b>b</b>) is the same representation with the addition of the sitting posture.</p>
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<p>Same representations as in <a href="#sensors-24-06208-f010" class="html-fig">Figure 10</a>. Here, the classes appear more separated compared to the previous method, but the results are very similar.</p>
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<p>Two-class detection was performed between falling and standing upright. As in the previous approaches, the behavior of the algorithm is good and it allows one to discriminate between postures.</p>
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<p>The images show two people in the same room that are in an upright position and periodically walk. On the left, the software correctly detects both of them, each with a single spot and corresponding point cloud. On the right, the image presents one subject with two associated spots, of which only the red one is correct, while the purple circle is an artifact.</p>
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<p>The image shows two people in a room, where the one on the left suffers from an artifact: the system loses the detection of the person for a few frames, and, when it recovers (image on the right), the reconstruction is altered towards the floor with the spot created at a very low level, which is incompatible with a person standing.</p>
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<p>A single person standing (<b>a</b>,<b>c</b>) and sitting (<b>b</b>,<b>d</b>): in the second case the point cloud is compacted to the most reflecting part of the body, the upper torso. This is the reason that the concentration of points is localized higher than the center of gravity of the person. (<b>c</b>,<b>d</b>) show also the corresponding confidence ellipses, with very different shapes and eccentricities, since, for the seated position, it resembles a circle.</p>
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<p>Output of the LSTM method considering the three postures. Sitting is shown in green, the fall is shown in blue and the upright position is shown in red. As above, the training sets are denoted by the fully colored circles, whereas the others denote the test sets.</p>
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<p>Confusion matrices for all AI methods considering all postures for the 50-50 ratio between the training and test sets: (<b>a</b>) LSTM; (<b>b</b>) Bi-LSTM; (<b>c</b>) projected LSTM; (<b>d</b>) GRU.</p>
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<p>Confusion matrices for all AI methods considering all postures for the 60-40 ratio between the training and test sets: (<b>a</b>) LSTM; (<b>b</b>) Bi-LSTM; (<b>c</b>) projected LSTM; (<b>d</b>) GRU.</p>
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<p>Confusion matrices for all AI methods considering all postures for the 70-30 ratio between the training and test sets: (<b>a</b>) LSTM; (<b>b</b>) Bi-LSTM; (<b>c</b>) projected LSTM; (<b>d</b>) GRU.</p>
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<p>Confusion matrices for all AI methods considering all postures for the 80-20 ratio between the training and test sets: (<b>a</b>) LSTM; (<b>b</b>) Bi-LSTM; (<b>c</b>) projected LSTM; (<b>d</b>) GRU.</p>
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<p>Confusion matrices for all AI methods considering all postures for the 90-10 ratio between the training and test sets: (<b>a</b>) LSTM; (<b>b</b>) Bi-LSTM; (<b>c</b>) projected LSTM; (<b>d</b>) GRU. As is clearly shown, the results are very promising, with slightly better performance in the cases of LSTM and Bi-LSTM.</p>
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<p>Confusion matrices for all AI methods considering only seated and upright postures: (<b>a</b>) LSTM; (<b>b</b>) Bi-LSTM; (<b>c</b>) projected LSTM; (<b>d</b>) GRU. In this case, the results are even more comparable than in <a href="#sensors-24-06208-f017" class="html-fig">Figure 17</a>, possibly because a person lying on the floor after a fall does not assume a precise posture, while sitting and standing are more stable positions.</p>
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18 pages, 5645 KiB  
Article
Assessing Vulnerabilities in Line Length Parameterization and the Per-Unit-Length Paradigm for Phase Modulation and Figure-of-Merit Evaluation in 60 GHz Liquid Crystal Phase Shifters
by Jinfeng Li and Haorong Li
Symmetry 2024, 16(10), 1261; https://doi.org/10.3390/sym16101261 - 25 Sep 2024
Abstract
The figure-of-merit (FoM) is a crucial metric in evaluating liquid crystal (LC) phase shifters, significantly influencing the selection of superior device candidates. This paper identifies, for the first time, a fundamental limitation in the widely-used High-Frequency Structure Simulator (HFSS), a closed-source commercial tool, [...] Read more.
The figure-of-merit (FoM) is a crucial metric in evaluating liquid crystal (LC) phase shifters, significantly influencing the selection of superior device candidates. This paper identifies, for the first time, a fundamental limitation in the widely-used High-Frequency Structure Simulator (HFSS), a closed-source commercial tool, when modeling reconfigurable delay line phase shifters (RDLPS) based on LC at millimeter-wave (mmW) frequencies for Beyond 5G (B5G) and Sixth-Generation (6G) applications. Specifically, the study reveals unreliable predictions of differential phase shifts (DPS) when using the line length parameterization (LLP) approach, with an accuracy of only 47.22%. These LLP-induced inaccuracies lead to misleading FoM calculations, potentially skewing comparative analyses against phase shifters implemented with different geometries or advanced technologies. Additionally, the per-unit-length (PUL) paradigm, commonly employed by microwave circuit engineers for evaluating and optimizing microwave transmission line designs, is also found to have limitations in the context of mmW RDLPS based on LC. The PUL methodology underestimates the FoM by 1.38206°/dB for an LC coaxial RDLPS at 60 GHz. These findings underscore a critical symmetry implication, where the assumed symmetry in phase shift response is violated, resulting in inconsistent performance assessments. To address these challenges, a remediation strategy based on a scenario-based “Length-for-π” (LFP) framework is proposed, offering more accurate performance characterization and enabling better-informed decision-making in mmW phase shifter design. Full article
(This article belongs to the Special Issue Feature Papers in Section "Engineering and Materials" 2024)
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<p>Illustration of the LC-enabled mmW phase-modulating mechanism via a coaxial delay line filled with tunable dielectrics (i.e., LC) from the isotropic state (0 V bias) to fully-aligned state (saturated bias), leading to the effective permittivity (<math display="inline"><mi>ℰ</mi></math><sub>eff</sub>) variation in line with the LC dielectric constant variation as derived by [<a href="#B25-symmetry-16-01261" class="html-bibr">25</a>,<a href="#B29-symmetry-16-01261" class="html-bibr">29</a>] for the mmW signal fully-occupied in the single-dielectric coaxial transmission line system.</p>
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<p>Illustrations of two distinct operation routines of running LLP (lengths varying from design 1 to design N) for LC-filled coaxial delay line’s differential phase shift prediction at 60 GHz.</p>
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<p>Photograph of our fabricated and assembled 60 GHz LC-filled phase shifters designed in two lengths for 0 to 180° (π) shifting and 0 to 360° (2π) shifting functionalities, respectively.</p>
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<p>DPS of coaxial lines of different lengths working at 60 GHz under the optimal cross-section size (fixed LC thickness <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">T</mi> </mrow> <mrow> <mi mathvariant="normal">L</mi> <mi mathvariant="normal">C</mi> </mrow> </msub> </mrow> </semantics></math> = 0.34876 mm). Length-varying-induced DPS evolution is compared among two computational operating approaches and benchmarked with theory [<a href="#B25-symmetry-16-01261" class="html-bibr">25</a>].</p>
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<p>Maximum insertion loss of LC-filled coaxial delay lines of various lengths at 60 GHz under an optimal cross-section dimension (fixed LC thickness <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">T</mi> </mrow> <mrow> <mi mathvariant="normal">L</mi> <mi mathvariant="normal">C</mi> </mrow> </msub> </mrow> </semantics></math> = 0.34876 mm).</p>
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<p>The 60 GHz FoM vs. line length of LC-filled coaxial phase shifter based on line length scanning run in the same model vs. run in separate models.</p>
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<p>Accuracy statistics of DPS (and FoM) by line length scanning (from 1 mm to 36 mm) run in a single HFSS parametric model of LC-filled coaxial phase shifter at 60 GHz.</p>
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<p>Elapsed time (in seconds) upon convergence for LC-filled coaxial delay lines of various lengths (from 1 mm to 36 mm) at 60 GHz under an optimal cross-section dimension (fixed LC thickness <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">T</mi> </mrow> <mrow> <mi mathvariant="normal">L</mi> <mi mathvariant="normal">C</mi> </mrow> </msub> </mrow> </semantics></math> = 0.34876 mm) and subject to two extreme biasing states of LC.</p>
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<p>Memory consumed (in MB) upon convergence for LC-filled coaxial delay lines of various lengths (from 1 mm to 36 mm) at 60 GHz under an optimal cross-section dimension (fixed LC thickness <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">T</mi> </mrow> <mrow> <mi mathvariant="normal">L</mi> <mi mathvariant="normal">C</mi> </mrow> </msub> </mrow> </semantics></math> = 0.34876 mm) and subject to two extreme biasing states of LC.</p>
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<p>The number of meshed elements solved upon convergence for LC-filled coaxial delay lines of various lengths (from 1 mm to 36 mm) at 60 GHz under an optimal cross-section dimension (fixed LC thickness <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">T</mi> </mrow> <mrow> <mi mathvariant="normal">L</mi> <mi mathvariant="normal">C</mi> </mrow> </msub> </mrow> </semantics></math> = 0.34876 mm) and subject to two extreme biasing states of LC.</p>
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<p>L = 1 mm: forward transmission (<math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">S</mi> </mrow> <mrow> <mn>21</mn> </mrow> </msub> </mrow> </semantics></math>) and forward reflection (<math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">S</mi> </mrow> <mrow> <mn>11</mn> </mrow> </msub> </mrow> </semantics></math>) coefficients for the coaxially tunable dielectric thickness of 0.41 mm design (50 Ω matched at <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ε</mi> </mrow> <mrow> <mi mathvariant="normal">L</mi> <mi mathvariant="normal">C</mi> </mrow> </msub> </mrow> </semantics></math> = 3.3). Simulated results were renormalized to 50 Ω and recorded for two extreme biasing states of LC.</p>
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<p>L = 1 mm: forward transmission (<math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">S</mi> </mrow> <mrow> <mn>21</mn> </mrow> </msub> </mrow> </semantics></math>) and forward reflection (<math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">S</mi> </mrow> <mrow> <mn>11</mn> </mrow> </msub> </mrow> </semantics></math>) coefficients for the coaxially tunable dielectric thickness of 0.34 mm design (50 Ω matched at <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ε</mi> </mrow> <mrow> <mi mathvariant="normal">L</mi> <mi mathvariant="normal">C</mi> </mrow> </msub> </mrow> </semantics></math> = 2.754). Simulated results were renormalized to 50 Ω and recorded for two extreme biasing states of LC.</p>
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<p>L = 15.92 mm for π shifting: forward transmission (<math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">S</mi> </mrow> <mrow> <mn>21</mn> </mrow> </msub> </mrow> </semantics></math>) and forward reflection (<math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">S</mi> </mrow> <mrow> <mn>11</mn> </mrow> </msub> </mrow> </semantics></math>) coefficients for the coaxially tunable dielectric thickness of <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="normal">T</mi> </mrow> <mrow> <mi mathvariant="normal">L</mi> <mi mathvariant="normal">C</mi> </mrow> </msub> </mrow> </semantics></math> = 0.34876 mm (50 Ω matched at <math display="inline"><semantics> <mrow> <msub> <mrow> <mi mathvariant="sans-serif">ε</mi> </mrow> <mrow> <mi mathvariant="normal">L</mi> <mi mathvariant="normal">C</mi> </mrow> </msub> </mrow> </semantics></math> = 2.8). Simulated results were renormalized to 50 Ω and recorded for two extreme biasing states of LC.</p>
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<p>Frequency response of FoM for three designs of a similar cross-section geometry but differing in line lengths: L = 1 mm (per-unit-length framework 1), 10 mm (per-unit-length framework 2), and 15.92 mm (length for π shifting).</p>
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18 pages, 3476 KiB  
Article
Study of Millimeter-Wave Fuze Echo Characteristics under Rainfall Conditions Using the Monte Carlo Method
by Bing Yang, Zhe Guo, Kaiwei Wu and Zhonghua Huang
Appl. Sci. 2024, 14(18), 8352; https://doi.org/10.3390/app14188352 - 17 Sep 2024
Abstract
Due to the similarity in wavelength between millimeter-wave (MMW) signals and raindrop diameters, rainfall induces significant attenuation and scattering effects that challenge the detection performance of MMW fuzes in rainy environments. To enhance the adaptability of frequency-modulated MMW fuzes in such conditions, the [...] Read more.
Due to the similarity in wavelength between millimeter-wave (MMW) signals and raindrop diameters, rainfall induces significant attenuation and scattering effects that challenge the detection performance of MMW fuzes in rainy environments. To enhance the adaptability of frequency-modulated MMW fuzes in such conditions, the effects of rain on MMW signal attenuation and scattering are investigated. A mathematical model for the multipath echo signals of the fuze was developed. The Monte Carlo method was employed to simulate echo signals considering multiple scattering, and experimental validations were conducted. The results from simulations and experiments revealed that rainfall increases the bottom noise of the echo signal, with rain backscatter noise predominantly affecting the lower end of the echo signal spectrum. However, rain conditions below torrential levels did not significantly impact the detection of strong reflection targets at the high end of the spectrum. The modeling approach and findings presented offer theoretical support for designing MMW fuzes with improved environmental adaptability. Full article
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<p>Attenuation coefficient.</p>
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<p>Rainfall scattering-phase function.</p>
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<p>Detection of MMW fuze under rainfall conditions, where the dots represent raindrops with different sizes, and the arrows indicate the propagation direction of MMW signals.</p>
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<p>Coordinate system of (<b>a</b>) particle launch, (<b>b</b>) target scattering, and (<b>c</b>) raindrop scattering. The letter A represents the actual scattering direction of the particles, B and C represent the projections of the actual scattering direction on the XOY and YOZ planes, respectively. The green arrow, red arrow, and the blue arrow represent the actual direction of particle incidence, the actual scattering direction and its projections of the particles, the specular reflection direction and its projections.</p>
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<p>Beat signal diagram at 9 m without rain: (<b>a</b>) time domain; (<b>b</b>) frequency domain.</p>
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<p>Beat signal spectrum at rainfall of 16 mm/h and operating frequency of 35 GHz: (<b>a</b>) 6 m, (<b>b</b>) 9 m, and (<b>c</b>) 12 m. The beat signal spectrum at rainfall of 50 mm/h and operating frequency of 35 GHz: (<b>d</b>) 6 m, (<b>e</b>) 9 m, and (<b>f</b>) 12 m. The beat signal spectrum at rainfall of 50 mm/h and operating frequency of 60 GHz: (<b>g</b>) 6 m, (<b>h</b>) 9 m, and (<b>i</b>) 12 m.</p>
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<p>Amplitude of (<b>a</b>) the target signal; (<b>b</b>) rain backscatter noise.</p>
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<p>Beat signal spectrum at distance of 9 m, rainfall rate of 50 mm/h, and operating frequency of 60 GHz for 3 dB beam widths of (<b>a</b>) 120°, (<b>b</b>) 60°, and (<b>c</b>) 30°.</p>
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<p>Detection scenarios: (<b>a</b>) 35 GHz and 60 GHz; (<b>b</b>) 24 GHz. The arrows in the figure connect the actual detection scene and the virtual schematic scene.</p>
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<p>24 GHz detector: (<b>a</b>) test results; (<b>b</b>) simulation result.</p>
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<p>Test results without rain and with torrential rain: (<b>a</b>) 35 GHz; (<b>b</b>) 60 GHz.</p>
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<p>Bottom noise test result (<b>a</b>) in [<a href="#B8-applsci-14-08352" class="html-bibr">8</a>]; (<b>b</b>) simulation.</p>
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40 pages, 4416 KiB  
Review
A Review on Millimeter-Wave Hybrid Beamforming for Wireless Intelligent Transport Systems
by Waleed Shahjehan, Rajkumar Singh Rathore, Syed Waqar Shah, Mohammad Aljaidi, Ali Safaa Sadiq and Omprakash Kaiwartya
Future Internet 2024, 16(9), 337; https://doi.org/10.3390/fi16090337 - 14 Sep 2024
Abstract
As the world braces for an era of ubiquitous and seamless connectivity, hybrid beamforming stands out as a beacon guiding the evolutionary path of wireless communication technologies. Several hybrid beamforming technologies are explored for millimeter-wave multiple-input multi-output (MIMO) communication. The aim is to [...] Read more.
As the world braces for an era of ubiquitous and seamless connectivity, hybrid beamforming stands out as a beacon guiding the evolutionary path of wireless communication technologies. Several hybrid beamforming technologies are explored for millimeter-wave multiple-input multi-output (MIMO) communication. The aim is to provide a roadmap for hybrid beamforming that enhances wireless fidelity. In this systematic review, a detailed literature review of algorithms/techniques used in hybrid beamforming along with performance metrics, characteristics, limitations, as well as performance evaluations are provided to enable communication compatible with modern Wireless Intelligent Transport Systems (WITSs). Further, an in-depth analysis of the mmWave hybrid beamforming landscape is provided based on user, link, band, scattering, structure, duplex, carrier, network, applications, codebook, and reflecting intelligent surfaces to optimize system design and performance across diversified user scenarios. Furthermore, the current research trends for hybrid beamforming are provided to enable the development of advanced wireless communication systems with optimized performance and efficiency. Finally, challenges, solutions, and future research directions are provided so that this systematic review can serve as a touchstone for academics and industry professionals alike. The systematic review aims to equip researchers with a deep understanding of the current state of the art and thereby enable the development of next-generation communication in WITSs that are not only adept at coping with contemporary demands but are also future-proofed to assimilate upcoming trends and innovations. Full article
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<p>Intelligent Transportation Systems (ITSs): the integration of satellite communication, millimeter-wave (mmWave) roadside units, and connected vehicles.</p>
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<p>MmWave-unmanned aerial vehicle (UAV) communications with scenarios such as aerial photography, surveillance, and remote sensing.</p>
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<p>Organization of the systematic review.</p>
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<p>In-depth analysis of Mm-wave hybrid beamforming landscape.</p>
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<p>Hybrid architecture with amplifiers [<a href="#B45-futureinternet-16-00337" class="html-bibr">45</a>].</p>
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<p>Subarray architecture [<a href="#B49-futureinternet-16-00337" class="html-bibr">49</a>,<a href="#B50-futureinternet-16-00337" class="html-bibr">50</a>].</p>
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<p>Muti-group hybrid beamforming design [<a href="#B60-futureinternet-16-00337" class="html-bibr">60</a>].</p>
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<p>Double analog combiner with low-resolution ADCs [<a href="#B62-futureinternet-16-00337" class="html-bibr">62</a>].</p>
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<p>Hybrid architecture [<a href="#B63-futureinternet-16-00337" class="html-bibr">63</a>].</p>
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<p>Subconnected architecture with switches [<a href="#B64-futureinternet-16-00337" class="html-bibr">64</a>].</p>
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<p>Fully connected architecture [<a href="#B64-futureinternet-16-00337" class="html-bibr">64</a>].</p>
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<p>Current trends in Mw-wave hybrid beamforming field.</p>
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<p>Challenges, solutions, and future research directions in Mw-wave hybrid beamforming field.</p>
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19 pages, 3745 KiB  
Article
A Three-Dimensional Fully Polarized Millimeter-Wave Hybrid Propagation Channel Model for Urban Microcellular Environments
by Chunzhi Hou, Qingliang Li, Jinpeng Zhang, Zhensen Wu, Yushi Zhang, Lixin Guo, Xiuqin Zhu and Pengbo Du
Electronics 2024, 13(18), 3629; https://doi.org/10.3390/electronics13183629 - 12 Sep 2024
Abstract
Millimeter-wave channel modeling is the basis of fifth-generation (5G) communication network design and applications. In urban microcellular environments, the roughness of wall surfaces can be comparable to the wavelengths of millimeter waves, resulting in walls that cannot be considered as smooth surfaces. Therefore, [...] Read more.
Millimeter-wave channel modeling is the basis of fifth-generation (5G) communication network design and applications. In urban microcellular environments, the roughness of wall surfaces can be comparable to the wavelengths of millimeter waves, resulting in walls that cannot be considered as smooth surfaces. Therefore, channel modeling methods based on only traditional three-dimensional ray tracing (RT) or the three-dimensional parabolic equation (PE) result in the limited computational accuracy of millimeter-wave channel models for urban environments. Based on the scattering theory of a rough surface and the typical scattering characteristics of a millimeter wave, the end field of the three-dimensional vector PE is regarded as the initial field of three-dimensional RT. Moreover, the number of scattered rays and scattering angles are introduced. Finally, a three-dimensional fully polarized millimeter-wave hybrid propagation channel model (3DFPHPCM) is proposed. The proposed model improves the computational accuracy of a single deterministic model. Millimeter-wave channel measurements in non-line-of-sight (NLOS) environments were carried out to verify and optimize the proposed 3DFPHPCM. The results show that the root mean square error (RMSE) and mean absolute error (MAE) of the proposed 3DFPHPCM are both minimized when compared to three-dimensional RT or the three-dimensional PE, which indicates that the proposed 3DFPHPCM has higher computational accuracy. Moreover, its runtime is the shortest among the methods. The results presented herein provide technical support for the layout of base stations. Full article
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<p>Sampling of scattered rays.</p>
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<p>The channel measurements in the NLOS scenario.</p>
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<p>Antenna pattern of the E-plane and H-plane horn antennas and normalized antenna pattern at 39 GHz, where (<b>a</b>) is two-dimensional horn antenna pattern and (<b>b</b>) is three-dimensional horn antenna pattern.</p>
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<p>Real-time MIMO channel detection system.</p>
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<p>Measurement in the urban microcellular environment.</p>
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<p>The trend of path loss with distance for different numbers of scattered rays when the scattering angle is 5°.</p>
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<p>The trend of path loss with distance for different numbers of rays when the number of the scattered rays is 9 at a scattering angle of 5°.</p>
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<p>The trend of path loss with distance for different scattering angles when the number of scattered rays is 9.</p>
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<p>The trend of path loss with distance around the scattering angle of 5° when the number of scattered rays is 9. (<b>a</b>) The scattering angle is from 4.1° to 5° and (<b>b</b>) the scattering angle is from 5° to 5.9°.</p>
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<p>The path loss varies with distance for scattering angles of 4.8° and 4.9° when the number of scattered rays is 9. (<b>a</b>) The scattering angle is from 4.8° and (<b>b</b>) the scattering angle is from 4.9°.</p>
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<p>The path loss varies with distance for scattering angles of 5° and 5.1° when the number of scattered rays is 9. (<b>a</b>) The scattering angle is from 5° and (<b>b</b>) the scattering angle is from 5.1°.</p>
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<p>The path loss varies with distance when using different methods.</p>
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19 pages, 11144 KiB  
Article
Millimeter-Wave Choke Ring Antenna with Broad HPBW and Low Cross-Polarization for 28 GHz Dosimetry Studies
by Philip Ayiku Dzagbletey and Jae-Young Chung
Electronics 2024, 13(17), 3531; https://doi.org/10.3390/electronics13173531 - 5 Sep 2024
Abstract
A choke ring horn antenna has been designed for use as an RF applicator in a compact range in vitro 28 GHz bioelectromagnetic exposure system. The 30 mm × 50 mm horn antenna was fabricated and measured to operate from 27.75 GHz to [...] Read more.
A choke ring horn antenna has been designed for use as an RF applicator in a compact range in vitro 28 GHz bioelectromagnetic exposure system. The 30 mm × 50 mm horn antenna was fabricated and measured to operate from 27.75 GHz to 34.5 GHz with a −20 dB measured S11 and a measured antenna gain of more than 10 dBi. A wide sectoral (flat top) and symmetric E- and H-plane pattern with a half-power beamwidth of more than 60 degrees was achieved with a cross-polarization discrimination of better than 28 dB. Electromagnetic slots were introduced in the antenna to suppress excess cavity mode radiation which inherently impacts the cross-polarization levels of choke ring antennas. The proposed antenna was successfully integrated into the compact measurement chamber in partnership with the Korea Telecommunication Research Institute (ETRI) and is currently in use for real-time 5G millimeter-wave dosimetry studies. Full article
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<p>Proposed mmWave in vitro exposure system (not drawn to scale).</p>
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<p>Power density profile example showing spot size.</p>
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<p>Power density curve on a plane.</p>
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<p>Spot size calculation.</p>
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<p>Remodeled horn antennas at 28 GHz with corresponding <b>E</b>-field and power density curves. (<b>a</b>) Conical Horn; (<b>b</b>) Pyramidal horn; (<b>c</b>) Choke ring horn; (<b>d</b>) contoured smooth-walled horn.</p>
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<p>Design concept of CRHA with conical flared opening.</p>
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<p>Simulated E-field and 3D radiation pattern of conical CRHA.</p>
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<p>Simulated gain of conical CRHA at 28 GHz.</p>
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<p>Parametric sweep for CRHA flare opening at 28 GHz.</p>
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<p>Simulated S11 curve of conical CRHA.</p>
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<p>Simulated gain curve of cylindrical CRHA design concept.</p>
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<p>CAD drawing of proposed choke ring horn antenna (CRHA).</p>
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<p>Top view of CRHA showing surface current in the choke. (<b>a</b>) CRHA without EM slots. (<b>b</b>) CRHA with EM slots.</p>
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<p>Simulated gain curve at 28 GHz of CRHA without EM slots.</p>
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<p>Simulated gain curve at 28 GHz of CRHA with EM slots.</p>
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<p>Photos of fabricated and assembled CRHA. (<b>a</b>) CRHA without EM slots. (<b>b</b>) CRHA with EM slots.</p>
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<p>Photos of fabricated and assembled CRHA. (<b>a</b>) Farfield measurement in anechoic chamber. (<b>b</b>) S-parameter measurement.</p>
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<p>Simulated and measured S11 curves of proposed CRHA.</p>
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<p>Simulated and measured gain of CRHA at 27.5 GHz.</p>
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<p>Simulated and measured gain of CRHA at 28 GHz.</p>
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<p>Simulated and measured gain of CRHA at 28.5 GHz.</p>
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<p>Photo showing CRHA and open-ended waveguide probe in anechoic chamber.</p>
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<p>Measured E-field power density profiles with varying exposure distances. (<b>a</b>) 50~100 mm distance, (<b>b</b>) 110~150 mm distance, (<b>c</b>) 160~200 mm distance, (<b>d</b>) 210~300 mm distance.</p>
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<p>Measured E-field power density profiles with varying exposure distances. (<b>a</b>) 50~100 mm distance, (<b>b</b>) 110~150 mm distance, (<b>c</b>) 160~200 mm distance, (<b>d</b>) 210~300 mm distance.</p>
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<p>Measured and post-processed field properties of the proposed CRHA at 28 GHz. (<b>a</b>) Spot size and E-field magnitude. (<b>b</b>) Impedance and power density. (<b>c</b>) Exposure efficiency.</p>
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34 pages, 5375 KiB  
Article
Advancing mmWave Altimetry for Unmanned Aerial Systems: A Signal Processing Framework for Optimized Waveform Design
by Maaz Ali Awan, Yaser Dalveren, Ali Kara and Mohammad Derawi
Drones 2024, 8(9), 440; https://doi.org/10.3390/drones8090440 - 28 Aug 2024
Viewed by 302
Abstract
This research advances millimeter-wave (mmWave) altimetry for unmanned aerial systems (UASs) by optimizing performance metrics within the constraints of inexpensive automotive radars. Leveraging the software-defined architecture, this study encompasses the intricacies of frequency modulated continuous waveform (FMCW) design for three distinct stages of [...] Read more.
This research advances millimeter-wave (mmWave) altimetry for unmanned aerial systems (UASs) by optimizing performance metrics within the constraints of inexpensive automotive radars. Leveraging the software-defined architecture, this study encompasses the intricacies of frequency modulated continuous waveform (FMCW) design for three distinct stages of UAS flight: cruise, landing approach, and touchdown within a signal processing framework. Angle of arrival (AoA) estimation, traditionally employed in terrain mapping applications, is largely unexplored for UAS radar altimeters (RAs). Time-division multiplexing multiple input–multiple output (TDM-MIMO) is an efficient method for enhancing angular resolution without compromising the size, weight, and power (SWaP) characteristics. Accordingly, this work argues the potential of AoA estimation using TDM-MIMO to augment situational awareness in challenging landing scenarios. To this end, two corner cases comprising landing a small-sized drone on a platform in the middle of a water body are included. Likewise, for the touchdown stage, an improvised rendition of zoom fast Fourier transform (ZFFT) is investigated to achieve millimeter (mm)-level range accuracy. Aptly, it is proposed that a mm-level accurate RA may be exploited as a software redundancy for the critical weight-on-wheels (WoW) system in fixed-wing commercial UASs. Each stage is simulated as a radar scenario using the specifications of automotive radar operating in the 77–81 GHz band to optimize waveform design, setting the stage for field verification. This article addresses challenges arising from radial velocity due to UAS descent rates and terrain variation through theoretical and mathematical approaches for characterization and mandatory compensation. While constant false alarm rate (CFAR) algorithms have been reported for ground detection, a comparison of their variants within the scope UAS altimetry is limited. This study appraises popular CFAR variants to achieve optimized ground detection performance. The authors advocate for dedicated minimum operational performance standards (MOPS) for UAS RAs. Lastly, this body of work identifies potential challenges, proposes solutions, and outlines future research directions. Full article
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<p>Uncertainty in altitude estimation due to wide HPBW of radar antenna [<a href="#B13-drones-08-00440" class="html-bibr">13</a>].</p>
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<p>Signal processing flow for point cloud generation in automotive FMCW radars [<a href="#B36-drones-08-00440" class="html-bibr">36</a>].</p>
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<p>Radar cube exhibiting slow-time, fast-time, and spatial dimensions [<a href="#B39-drones-08-00440" class="html-bibr">39</a>].</p>
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<p>Fundamentals of AoA estimation in an SIMO radar [<a href="#B43-drones-08-00440" class="html-bibr">43</a>].</p>
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<p>Standard deviation in terrain elevation for various land types [<a href="#B46-drones-08-00440" class="html-bibr">46</a>].</p>
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<p>Comparison of range profile: (<b>a</b>) Single chirp per frame; (<b>b</b>) 16 chirps per frame.</p>
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<p>Range profile and CFAR threshold.</p>
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<p>Range profile and CFAR threshold: (<b>a</b>) CFAR−CA; (<b>b</b>) CFAR−CASO.</p>
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<p>AoA estimation at high altitude.</p>
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<p>2Tx–4Rx virtual antenna array in an MIMO radar.</p>
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<p>1Tx–8Rx physical antenna array in an SIMO radar.</p>
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<p>Combined radiation patterns: (<b>a</b>) 1 × 8 SIMO; (<b>b</b>) 2 × 4 MIMO.</p>
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<p>TDM-MIMO chirp frame with Doppler induced due to radial velocity.</p>
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<p>VTOL UAS landing on a drone ship surrounded by water.</p>
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<p>Angular FFT showing peaks in respective bins: (<b>a</b>) Scenario 1; (<b>b</b>) Scenario 2.</p>
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<p>Zoom FFT implementation.</p>
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<p>Range profiles: (<b>a</b>) Coarse; (<b>b</b>) fine.</p>
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22 pages, 893 KiB  
Article
Unlicensed Spectrum Access and Performance Analysis for NR-U/WiGig Coexistence in UAV Communication Systems
by Zhenzhen Hu, Yong Xu, Yonghong Deng and Zhongpei Zhang
Drones 2024, 8(9), 439; https://doi.org/10.3390/drones8090439 - 28 Aug 2024
Viewed by 377
Abstract
Unmanned aerial vehicles (UAVs) are extensively employed in pursuit, rescue missions, and agricultural applications. These operations necessitate substantial data and video transmission, requiring significant spectral resources. The unlicensed millimeter wave (mmWave) spectrum, especially in the 60 GHz frequency band, offers promising potential for [...] Read more.
Unmanned aerial vehicles (UAVs) are extensively employed in pursuit, rescue missions, and agricultural applications. These operations necessitate substantial data and video transmission, requiring significant spectral resources. The unlicensed millimeter wave (mmWave) spectrum, especially in the 60 GHz frequency band, offers promising potential for UAV communications. However, WiGig users are the incumbent users of the 60 GHz unlicensed spectrum. Therefore, to ensure fair coexistence between UAV-based new radio-unlicensed (NR-U) users and WiGig users, unlicensed spectrum-sharing strategies need to be meticulously designed. Due to the beam directionality of the NR-U system, traditional listen-before-talk (LBT) spectrum sensing strategies are no longer effective in NR-U/WiGig systems. To address this, we propose a new cooperative unlicensed spectrum sensing strategy based on mmWave beamforming direction. In this strategy, UAV and WiGig users cooperatively sense the unlicensed spectrum and jointly decide on the access strategy. Our analysis shows that the proposed strategy effectively resolves the hidden and exposed node problems associated with traditional LBT strategies. Furthermore, we consider the sensitivity of mmWave to obstacles and analyze the effects of these obstacles on the spectrum-sharing sensing scheme. We examine the unlicensed spectrum access probability and network throughput under blockage scenarios. Simulation results indicate that although obstacles can attenuate the signal, they positively impact unlicensed spectrum sensing. The presence of obstacles can increase spectrum access probability by about 60% and improve system capacity by about 70%. Full article
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<p>Co-location of UE and WUE in a NR-U/WiGig coexistence system.</p>
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<p>False alarm and miss detection problems in a NR-U/WiGig coexistence system.</p>
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<p>The scenario of gNB, WiGig, UE, and WUE aligned in a straight line.</p>
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<p>The proposed spectrum sensing access protocol flowchart.</p>
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<p>The frame structures of gNB and WiGig for coexistence systems.</p>
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<p>The unlicensed spectrum coexistence systems with blockage.</p>
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<p>The Simulation System Model for Coexistence in Unlicensed Spectrum.</p>
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<p>Received Power Comparison with Different Spectrum Access Strategies.</p>
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<p>Sum Rate Comparison with Different Interference Threshold under the Proposed Spectrum Access Strategy.</p>
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<p>Access Probability Comparison with Different Interference Threshold.</p>
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<p>Access Probability Comparison with and without Blockage.</p>
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<p>Sum Rate Comparison with and without Blockage.</p>
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<p>Sum Rate Versus Access Probability.</p>
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<p>Sum Rate Versus mainbeamwidth of gNB, <math display="inline"><semantics> <msub> <mi>θ</mi> <mrow> <mi>B</mi> <mo>,</mo> <mi>M</mi> </mrow> </msub> </semantics></math>.</p>
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9 pages, 4326 KiB  
Communication
A Highly Integrated Millimeter-Wave Circularly Polarized Wide-Angle Scanning Antenna Unit
by Guishan Yuan, Sai Guo, Kan Wang and Jiawen Xu
Electronics 2024, 13(16), 3328; https://doi.org/10.3390/electronics13163328 - 22 Aug 2024
Viewed by 359
Abstract
This paper introduces a novel, small-sized, highly integrated, circularly polarized wide-angle scanning antenna using substrate-integrated waveguide (SIW) technology at millimeter-wave frequencies. The antenna unit addresses requirements for high data transmission rates, wide spatial coverage, and strong interference resistance in communication systems. By integrating [...] Read more.
This paper introduces a novel, small-sized, highly integrated, circularly polarized wide-angle scanning antenna using substrate-integrated waveguide (SIW) technology at millimeter-wave frequencies. The antenna unit addresses requirements for high data transmission rates, wide spatial coverage, and strong interference resistance in communication systems. By integrating radiating square waveguides, circular polarizers, filters, and matching loads, the antenna enhances out-of-band suppression, eliminates cross-polarization, and reduces manufacturing complexity and costs. Utilizing this antenna unit as a component, a 4 × 4 phased array antenna with a two-dimensional ±60° scanning capability is designed and simulated. The simulation and measurement results confirm that the phased array antenna achieves the desired scan range with a gain reduction of less than 3.9 dB. Full article
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<p>Schematic diagram of the SIW square waveguide.</p>
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<p>The structure of the radiating square waveguide incorporating the circular polarizer.</p>
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<p>(<b>a</b>) Model of the SIW filter structure. (<b>b</b>) The field distribution of the SIW filter structure.</p>
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<p>Model of the matching load structure.</p>
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<p>Model of the antenna unit structure from different perspectives: (<b>a</b>) Perspective; (<b>b</b>) Front view; (<b>c</b>) Side view.</p>
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<p>Electrical performance of the filter: (<b>a</b>) VSWR; (<b>b</b>) insertion loss.</p>
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<p>The VSWR of the matching load.</p>
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<p>The active VSWR for θ = 0°, 20°, 40°, and 60°: (<b>a</b>) φ = 0°; (<b>b</b>) φ = 90°.</p>
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<p>The illustration of the array layout.</p>
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<p>The simulation of the active radiation pattern at frequency <span class="html-italic">f</span><sub>0</sub> GHz.</p>
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<p>The fabricated antenna array.</p>
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<p>The comparison of simulated and measured results of the scanning pattern at the center frequency.</p>
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